Research on Single-Phase PWM Converter with Reverse Conducting IGBT Based on Loss Threshold Desaturation Control

energies

Article Research on Single-Phase PWM Converter with Reverse Conducting IGBT Based on Loss Threshold Desaturation Control

Xianjin Huang * ID , Dengwei Chang, Chao Ling and Trillion Q. Zheng

School of Electrical Engineering, Beijing Jiaotong University, No. 3 Shangyuancun, Beijing 100044, China; [email protected] (D.C.); [email protected] (C.L.); [email protected] (T.Q.Z.) * Correspondence: [email protected]; Tel.: +86-136-8358-7910

Received: 15 September 2017; Accepted: 5 November 2017; Published: 12 November 2017

Abstract: In the application of vehicle power supply and distributed power generation, there are strict requirements for the pulse width modulation (PWM) converter regarding power density and reliability. When compared with the conventional insulated gate bipolar transistor (IGBT) module, the Reverse Conducting-Insulated Gate Bipolar Transistor (RC-IGBT) with the same package has a lower thermal resistance and higher current tolerance. By applying the gate desaturation control, the reverse recovery loss of the RC-IGBT diode may be reduced. In this paper, a loss threshold desaturation control method is studied to improve the output characteristics of the single-phase PWM converter with a low switching frequency. The gate desaturation control characteristics of the RC-IGBT’s diode are studied. A proper current limit is set to avoid the ineffective inﬂiction of the desaturation pulse, while the bridge arm current crosses zero. The expectation of optimized loss decrease is obtained, and the better performance for the RC-IGBTs of the single-phase PWM converter is achieved through the optimized desaturation pulse distribution. Finally, the improved predictive current control algorithm that is applied to the PWM converter with RC-IGBTs is simulated, and is operated and tested on the scaled reduced power platform. The results prove that the gate desaturation control with the improved predictive current algorithm may effectively improve the RC-IGBT’s characteristics, and realize the stable output of the PWM converter.

Keywords: RC-IGBT; low switching frequency; PWM converter; diode desaturation; loss threshold control

1. Introduction The main feature of the pulse width modulation (PWM) converter is ensuring the high power factor and AC side sine wave current control on the target of the output DC voltage, as well as on the bidirectional power transmission. With the development of power semiconductor devices, the improvement of modeling and control strategy and sensing technology, the working performance of PWM converters is constantly improving. Electric rail transportation, uninterruptible power system (UPS), and a distributed power generation that is based on wind and solar energy storage have higher requirements on the power density, working temperature, cycle life, and reliability. The current research on PWM converters mainly concentrates on three aspects: new circuit topology, universality of the converter mathematical model, and control strategy [1–5]. Ref. [6] adopted a hybrid clamped five-level topology to successfully solve the non-uniformity of the DC capacitor voltage of the three-phase PWM converter. Ref. [7] gave a mathematical model of a single-phase PWM converter based on the unipolar modulation in the stationary coordinate and d-q coordinate. Refs. [8,9] proposed the predictive current control to achieve zero steady-state error of the grid current, and greatly reduce its harmonic content. In addition, due to the fact that power devices are the cornerstone of the converter system, there have also been researches

Energies 2017, 10, 1845; doi:10.3390/en10111845 www.mdpi.com/journal/energies Energies 2017, 10, 1845 2 of 17 performed on electrical stress, temperature loss, and reliability of PWM converters. Ref. [10] explained that the Reverse Conducting-Insulated Gate Bipolar Transistor (RC-IGBT) has a uniform current density and temperature distribution uniformity in insulated gate bipolar transistor (IGBT) or diode mode, and greatly improves the reliability of the equipment. Ref. [11] effectively realized the two-way flow of the point current, and improved the efficiency of the inverter by introducing reverse blocking IGBT (RB-IGBT) to the T-type three-level inverter. Due to their strong electrical characteristics and gate control capability, IGBT devices have now fully occupied 500 Hz–100 kHz high voltage and high power applications. In the past 30 years, IGBT devices have been constantly updated in terms of structure results, leading to the continuous improvement in performance. The structures of the devices have undergone a technical revolution of non-punch through (NPT), field stop (FS), and so on, in which, from the earliest punch through (PT) type, the surface structure developed from a plane grid to a groove gate structure, and various new structures and derivative devices emerged in an endless stream [12]. As a result, the resistance junction temperature has become increasingly higher, the device loss is reducing, the switching speed is becoming faster, and the voltage level and current capacity are improving significantly, gradually approaching the material performance limit. Commercial IGBT single modules have reached the 6500 V/1000 A and 4500 V/3000 A levels [13]. Reverse conducting IGBT allows for IGBT chips and freewheeling diode chips to integrate in the same cellular chip within the internal structure and function design, which improves the thermal efficiency of the chip and reduces the parasitic parameters of chip connection. It has a higher current tolerance than that of the traditional IGBT with the same package. The RC-IGBT integrated diode is controlled by desaturation pulse, which can significantly optimize the diode reverse recovery characteristic. The reverse conducting IGBT device is applied to the locomotive traction converter, which has a positive effect on reducing the number of devices and improving the performance of the temperature rise and heat dissipation [14,15]. In this paper, the driving and loss characteristics of RC-IGBT are introduced, and the improved predictive control method of RC-IGBT for single-phase PWM rectifier is studied. By setting the desaturation pulse control sequence of four switch devices in full bridge circuit, the diode desaturation control is realized to reduce reverse recovery loss. Because the inductor current may be zero crossing repeatedly and the small current detection may be error in the practical operation of single-phase PWM rectifier, it can cause the desaturation pulse repeated application, the time overlap, logic confusion, and other problems at the case of the criterion of the direction of the inductor current. A loss reduction threshold desaturation control method for RC-IGBT single-phase PWM rectifier is proposed. The threshold value of the current is determined by the threshold of loss variation in order to judge the application of the desaturation pulse. The loss of threshold desaturation control of RC-IGBT PWM rectifier is simulated by simulation, and the system loss is verified by analyzing and comparing the system loss with different threshold values. Finally, the experiment is carried out on a low power experimental platform, and the results also verify the correctness of the scheme.

2. Control Characteristics of RC-IGBT PWM Converter

2.1. Driving and Loss Characteristics of RC-IGBT As shown in Figure1, RC-IGBT allows the free-wheeling diode (FWD) and IGBT to be successfully integrated into silicon, by partly replacing the P-Collector (p-doped collector) with N-Collector (n-doped collector) [16,17]. EnergiesEnergies 20172017,, 1010,, 18451845 3 of 1718 Energies 2017, 10, 1845 3 of 18

(a) (b) (a) (b) Figure 1.The diagram of Reverse Conducting-Insulated Gate Bipolar Transistor (RC-IGBT): (a) The Figure 1. The diagram of Reverse Conducting-Insulated Gate Bipolar Transistor (RC-IGBT): (a) The internal structure; (b) the electrical symbol. internalFigure 1. structure;The diagram (b) the of electricalReverse Conduc symbol.ting-Insulated Gate Bipolar Transistor (RC-IGBT): (a) The internal structure; (b) the electrical symbol. The operating characteristics of the RC-IGBT integrated diode are affected by the gate voltage, Theand the operating application characteristics of the gate voltage of the can RC-IGBT change integratedthe flowing diodepath and are the affected recombination by the gate process voltage, The operating characteristics of the RC-IGBT integrated diode are affected by the gate voltage, and theof the application carriers. The of gate the gatepulse voltagevoltage can affect change the theforward ﬂowing voltag pathe drop and of the the recombinationintegrated diode process in and the application of the gate voltage can change the flowing path and the recombination process of theRC-IGBT. carriers. As The shown gate in pulse Figure voltage 2, the V-I can characterist affect theic forward curves of voltage the integrated drop of diode the integratedat 25 °C and diode of the carriers. The gate pulse voltage can affect the forward voltage drop of the integrated diode in in RC-IGBT.125 °C working As shown temperatures in Figure are2, thegiven V-I for characteristic different gate voltages. curves of The the arrow integrated in the figure diode shows at 25 that◦C and RC-IGBT.the gate As voltages shown arein −Figure15 V, 0 2,V, theand V-I 15 V,characterist respectively.ic Thecurves greater of the the integratedgate voltage, diode the higher at 25 the °C and 125 ◦C working temperatures are given for different gate voltages. The arrow in the ﬁgure shows that 125 °Cforward working voltage temperatures drop across are the given same for current different level gatewill be.voltages. In the twoThe levelarrow PWM in the control, figure driving shows that the gate voltages are −15 V, 0 V, and 15 V, respectively. The greater the gate voltage, the higher the the gatesignal voltages of the conventional are −15 V, 0IGBT V, and was 15not V, considering respectively. the working The greater state ofthe the gate diode; voltage, otherwise, the higherin the the forward voltage drop across the same current level will be. In the two level PWM control, driving forwardapplication voltage of drop driving across RC-IGBT, the same the diode current conducti level onwill state be. should In the betwo detected level PWMto impose control, the low driving signal of the conventional IGBT was not considering the working state of the diode; otherwise, in the signallevel of the gate conventional voltage on the IGBT RC-IGBT was for not the considering aim of reducing the working the diode state conduction of the diode;loss [18,19]. otherwise, in the application of driving RC-IGBT, the diode conduction state should be detected to impose the low level application of driving RC-IGBT,60 the diode conduction state should be detected to impose the low gate voltage on the RC-IGBTVGE for=－15V the aim of reducing the diode conduction loss [18,19]. level gate voltage on the RC-IGBT for the aim of reducing the diode conduction loss [18,19]. 50 60 VGE=＋15V

C － 40VGE= 15V -I t

n VGE=0V 50e r r 30 u VGE=＋15V /A C 40 -I ode c ode i t 20 d n VGE=0V e

r 25℃ r 30 u 10 125℃

ode c ode 0 i 20 d 024681218 10 14 16 ℃ diode voltage-VCE/V 25 10 125℃ Figure 2. Static characteristic of RC-IGBT in the diode mode [20]. 0 024681218 10 14 16 The low level gate voltage applied to the diode mode of the RC-IGBT could maintain the low diode voltage-VCE/V diode conduction voltage drop, while the high level gate voltage could provide additional compound carriers for the diodeFigure reverse 2. Static recovery characteristic to reduce of RC-IGBTRC-I the GBTreverse in the recovery diode mode current. [[20].20 ].Before the reverse recovery process of the diode, a high level pulse is applied to desaturating the saturated diode temporarily. This pulse setting method is called the RC-IGBT diode desaturation control (RCDC). The lowlow levellevel gate gate voltage voltage applied applied to theto the diode diode mode mode of the of RC-IGBTthe RC-IGBT could could maintain maintain the low the diode low The desaturation pulse control is based on several key factors, such as the imposed pulse time, the diode conduction voltage drop, while the high level gate voltage could provide additional compound conductionamplitude voltage of the drop, pulse while voltage, the th highe pulse level duration, gate voltage the pulse could turn-off provide time, additional and lock compoundtime. Ref. [20] carriers carriers for the diode reverse recovery to reduce the reverse recovery current. Before the reverse for thestudied diode the reverse relationship recovery between to reduce the single the RC-IGBT reverse desaturation recovery current. pulse control Before and the the reverse device recoveryloss processrecovery(including of process the diode,IGBT of loss, the a high diodediode, level loss, a pulsehighsteady levelis state applied pulsloss, edynamic tois desaturatingapplied loss, andto desaturating total thesaturated loss). Our the research diode saturated temporarily. in this diode Thistemporarily. pulse setting This methodpulse setting is called method the RC-IGBT is called diode the RC-IGBT desaturation diode control desaturation (RCDC). Thecontrol desaturation (RCDC). pulseThe desaturation control is based pulse on control several keyis based factors, on suchseveral as thekey imposed factors, pulsesuch time,as the the imposed amplitude pulse of thetime, pulse the voltage,amplitude the of pulse the pulse duration, voltage, the pulsethe pulse turn-off duration, time, andthe pulse lock time. turn-off Ref. time, [20] studiedand lock the time. relationship Ref. [20] studied the relationship between the single RC-IGBT desaturation pulse control and the device loss (including IGBT loss, diode loss, steady state loss, dynamic loss, and total loss). Our research in this

Energies 2017, 10, 1845 4 of 17 between the single RC-IGBT desaturation pulse control and the device loss (including IGBT loss, diodeEnergies loss, 2017 steady, 10, 1845 state loss, dynamic loss, and total loss). Our research in this paper is based4 of 18 on the fact that the proper desaturation pulse control has been selected on the single RC-IGBT. In a switching paper is based on the fact that the proper desaturation pulse control has been selected on the single processRC-IGBT. of RC-IGBT, In a switching the desaturation process of RC-IGBT, pulse control the desaturation can achieve pulse better control loss suppression.can achieve better loss 0 suppression.Figure3b shows the sequential logic of the desaturation pulse U ge-T1. The T1 turns off at the time t0, theFigure diode 3b shows starts the to sequential be free-wheeling, logic of the the desaturation order of the pulse T2 Uturning′ge-T1. The on T1 isturns execute off at atthe the time time t1, the desaturationt0, the diode starts pulse to startsbe free-wheeling, to be applied the order to the of T 1theat T the2 turning time ton2, itis stopsexecute being at the applied time t1, tothe the T1 at thedesaturation time t3, and pulse the starts T2 turns to be applied on actually to the at T1 the at the time timet4 .t2 Because, it stops being the time applied of turning to the T1 on at ofthe the T2 time t3, and the T2 turns on actually at the time t4. Because the time of turning on of the T2 cannot be cannot be predicted in advance, the gate level pulse signal of T2 is delayed, and the delay time is tdelay. predicted in advance, the gate level pulse signal of T2 is delayed, and the delay time is tdelay. The tfree is The tfree is the system processing time, the tdesat is the desaturation pulse width, and the tlock is the dead the system processing time, the tdesat is the desaturation pulse width, and the tlock is the dead time. time. Similarly, when the load current IL is reverse (outﬂow node), it is necessary to detect the forward Similarly, when the load current IL is reverse (outflow node), it is necessary to detect the forward conduction time of the T1, and to also apply the desaturation pulse to the T2 [21,22]. conduction time of the T1, and to also apply the desaturation pulse to the T2 [21,22]. TheThe values values of of the the above above time time havehave the following relationships: relationships:

tdelay= tfree+ tdesat+ tlock (1) tdelay = t free + t desat + t lock (1) tdelaytdelayis determinedis determined by by the the three three latterlatter values. values.

Figure 3. Desaturation pulse control of RC-IGBT in half bridge: (a) The structure of RC-IGBT in half bridge;Figure (b) 3. The Desaturation logic of desaturation pulse control pulse of RC-IGBT control. in half bridge: (a) The structure of RC-IGBT in half bridge; (b) The logic of desaturation pulse control.

TheThe loss loss of IGBTof IGBT converter converter is is mostlymostly obtained by by integrating integrating the the voltage voltage and andcurrent current of the of actual the actual circuit,circuit, and and then then obtained obtained according according to theto the V-I V-I curve curve of of the the device device parameter paramete manualr manual [ 23[23].]. Figure Figure4 4shows the currentshows the and current switch and status switch on status the rectify on the operation rectify operation of single-phase of single-phase PWM PWM rectifier. rectifier. The totalThe total loss of all the deviceslossEnergies of all 2017 in the the, 10 devices, halfx FOR modulation PEER in the REVIEW half modulation period could period be calculated could be calculated based on based the working on the working status. status.5 of 18 The green region corresponding to one IGBT conduction loss and one diode conduction loss, the red region corresponding to the conduction loss of two diodes, and the yellow region is is corresponding to the conduction loss of two IGBTs. The dashed area corresponding one IGBT conduct intog the loss of the IGBT turning off, and the solid line region is corresponding to the IGBTone d ioturn-onde conductin gloss and diode two diodes conducting reverse recovery loss. two IGBTs conducting IGBT turning off IGBT turning on diode reverse recovery 0 t 2Eon_IGBT Eon, Erec Eoff 2Eon_diode Eon_IGBT Eon_diode

Figure 4. Loss analysis of the single phase PWM rectiﬁer during a modulation period. Figure 4. Loss analysis of the single phase PWM rectifier during a modulation period.

m+1

Econ_ IGBT_ half = ∑{VCE[IC(t2i-1)]∙IC(t2i-1)∙TT_ 2i-1} + 2VCE[IC(t2)]∙IC(t2)∙TT_ 2 (2) i=1

Eon_ half = ∑{Eon[IC_ on(t2i)]} + Eon[IC_ on(t1)] (3) i=2 m

Eoff_ half = ∑{Eoff[IC_ off(t2i-1)]} + Eoff[IC_ off(t2)] (4) i=2

m+1 m

Eon_ diode_ half = ∑{VF[IF1(t2i-1)]∙IF(t2i-1)∙TD_ 2i-1} + 2 ∑{VF[IF1(t2i)]∙IF(t2i)∙TD_ 2i} (5) i=1 i=2

Erec_ half = ∑{Erec[IC_ on(t2i)]} + Erec[IC_ on(t1)] (6) i=2

The total loss during a modulation cycle: (7) Etot = 2(Econ_ IGBT_ half + Eon_ half + Eoff_ half + Eon_ Diode_ half + Erec_ half)

where IC (ti) and TT_i are the average current and the conduction time of the IGBT conduction of the i- th working state; IC_on(ti) and IC_off(ti) are the collector currents of IGBT turn-on and turn-off time of the i-th working state, IF(ti) and TD_i is the average current and the conduction time of the diode conduction of the i-th working state, and m is the ratio of the carrier frequency to the modulation wave frequency. The RC-IGBT loss calculation is similar to that of the conventional IGBT, including IGBT switching loss and conduction loss, diode conduction loss, and reverse recovery loss [24]. The calculation of the IGBT switching loss and conduction loss for RC-IGBT is as the same as that for conventional IGBT, without considering if the RC-IGBT worked under RCDC or not. The calculation

Energies 2017, 10, 1845 5 of 17

The green region corresponding to one IGBT conduction loss and one diode conduction loss, the red region corresponding to the conduction loss of two diodes, and the yellow region is corresponding to the conduction loss of two IGBTs. The dashed area corresponding to the loss of the IGBT turning off, and the solid line region is corresponding to the IGBT turn-on loss and diode reverse recovery loss. In the half modulation cycle, IGBT conduction loss, IGBT turn-on loss, IGBT turn-off loss, diode conduction loss, and diode reverse recovery loss could be written as follows: m+1 Econ_ IGBT_ half = ∑ {VCE[IC(t2i−1)]·IC(t2i−1)·TT_ 2i−1} + 2VCE[IC(t2)]·IC(t2)·TT_ 2 (2) i=1 m Eon_ half = ∑{Eon[IC_ on(t2i)]} + Eon[IC_ on(t1)] (3) i=2 m n h io h i Eoff _ half = ∑ Eoff IC_ off (t2i−1) + Eoff IC_ off (t2) (4) i=2 m+1 m Eon_ diode_ half = ∑ {VF[IF1(t2i−1)]·IF(t2i−1)·TD_ 2i−1} + 2 ∑{VF[IF1(t2i)]·IF(t2i)·TD_ 2i} (5) i=1 i=2 m Erec_ half = ∑{Erec[IC_ on(t2i)]} + Erec[IC_ on(t1)] (6) i=2 The total loss during a modulation cycle: Etot = 2 Econ_ IGBT_ half + Eon_ half + Eoff _ half + Eon_ Diode_ half + Erec_ half (7) where IC (ti) and TT_i are the average current and the conduction time of the IGBT conduction of the i-th working state; IC_on(ti) and IC_off(ti) are the collector currents of IGBT turn-on and turn-off time of the i-th working state, IF(ti) and TD_i is the average current and the conduction time of the diode conduction of the i-th working state, and m is the ratio of the carrier frequency to the modulation wave frequency. The RC-IGBT loss calculation is similar to that of the conventional IGBT, including IGBT switching loss and conduction loss, diode conduction loss, and reverse recovery loss [24]. The calculation of the IGBT switching loss and conduction loss for RC-IGBT is as the same as that for conventional IGBT, without considering if the RC-IGBT worked under RCDC or not. The calculation of the diode loss of RC-IGBT under saturation control is consistent with that of the conventional IGBT diode. Under the diode desaturation control, the calculation method is similar, but the selected parameters are different according to the desaturation pulse. The loss difference between the RC-IGBT under the desaturation control and the RC-IGBT under the saturation control (the conventional IGBT) is analyzed as following. During a switching period, the difference of the IGBT turn-on loss is defined as ∆Eon, the conducting loss and the turn-off loss are the same (constant, if the conventional IGBT), the difference of the diode conduction loss is defined as ∆Eon_diode, and the difference of the diode reverse recovery loss is defined as ∆Erec. Under the two control conditions, the total loss variation of RC-IGBT, ∆Etot can be expressed as:

∆Etot = ∆Eon + ∆Eon_diode + ∆Erec + const (8)

For the convenience of calculation, the duty cycle is set 0.5, the theoretical turn-on time of the diode is half of the switching period. IGBT turn-on loss actually includes the loss caused by normal turn-on current and the loss that is caused by the superposed reverse recovery current of the corresponding diode. Under the two control conditions, the turn-on loss variation of RC-IGBT, ∆Etot can be expressed as: 2 1 1 trr 1 2 trrtA ∆Eon = βVCE·IRM·trr − βVCE·IRM· − 2β − β VCE·IRM· (9) 2 6 ton 6 ton where ton is the turn-on time of RC-IGBT, the reverse recovery time is trr = tA + tB, tA, and tB are corresponding to the rise time and the fall recovery time, IRM is the peak current of the diode reverse Energies 2017, 10, 1845 6 of 18

defined as ΔEon, the conducting loss and the turn-off loss are the same (constant, if the conventional IGBT), the difference of the diode conduction loss is defined as ΔEon_diode, and the difference of the diode reverse recovery loss is defined as ΔErec. Under the two control conditions, the total loss variation of RC-IGBT, ΔEtot can be expressed as: (8) ∆Etot = ∆Eon + ∆Eon_diode + ∆Erec + const For the convenience of calculation, the duty cycle is set 0.5, the theoretical turn-on time of the diode is half of the switching period. IGBT turn-on loss actually includes the loss caused by normal turn-on current and the loss that is caused by the superposed reverse recovery current of the corresponding diode. Under the two control conditions, the turn-on loss variation of RC-IGBT, ΔEtot can be expressed as: 2 1 1 trr 1 2 trrtA (9) Energies 2017, 10, 1845 ∆Eon = βVCE·IRM·trr － βVCE·IRM· － 2β － β VCE·IRM· 6 of 17 2 6 ton 6 ton

where ton is the turn-on time of RC-IGBT, the reverse recovery time is trr = tA + tB, tA, and tB are corresponding to the rise time and the fall recovery time, IRM is the peak current of the diode reverse recovery, IC is the collector current, VCE is the DC voltage, and β is the inﬂuence coefﬁcient with the desaturationrecovery, pulse, IC is the which collector is only current, related VCE is to the the DC peak voltage, current and ofβ is the thediode influence reverse coefficient recovery, with the rise desaturation pulse, which is only related to the peak current of the diode reverse recovery, the rise time and the fall recovery time. time and the fall recovery time. Under the two control conditions, the diode reverse recovery loss variation of RC-IGBT, ∆Erec can Under the two control conditions, the diode reverse recovery loss variation of RC-IGBT, ΔErec be expressedcan be expressed as: as: 1 1 2 ∆Erec = 1 βVCE·IRM·trr 1+ 2β − β VCE·IRM·tA (10) (10) ∆E = 6 βV ·I ·t ＋ 26β－ β2V ·I ·t rec 6 CE RM rr 6 CE RM A The diode conduction loss under the desaturation control is less than that of the saturation The diode conduction loss under the desaturation control is less than that of the saturation control. Under the two control conditions, the diode conduction loss variation of RC-IGBT, ∆Erec can control. Under the two control conditions, the diode conduction loss variation of RC-IGBT, ΔErec can be expressedbe expressed as: as: ∆Erec = ∆VF·IC·(tdiode_ on − tdes) (11) ∆Erec = ∆VF·IC·tdiode_ on－ tdes (11) where ∆VF is the difference of the diode conduction voltages under the two control conditions with where ΔVF is the difference of the diode conduction voltages under the two control conditions with t t differentdifferent gate gate voltages, voltages,des tisdes theis the desaturation desaturation pulsepulse width, and and tdiode_ondiode_on is theis diode the diode conduction conduction time. time.

2.2. Pulse2.2. Pulse Timing Timing of PWM of PWM Converter Converter with with RC-IGBT RC-IGBT

FigureFigure5 is the 5 is main the main circuit circuit structure structure of the of single-phasethe single-phase voltage-source voltage-source PWM PWM converter. converter. TT1~T~T4 4 are full controlledare full controlled devices, devices,Lm and Lm Randm are, Rm are, respectively, respectively, an an inductance inductance and and equivalent equivalent resistance resistance of the of the AC side,AC side, and Candd is C ad is capacitor a capacitor of of the the DC DC side. side. In In addition,addition, uus sandand is iares are the the grid grid voltage voltage and andcurrent, current, and uab and udc are the AC side voltage and the DC output voltage, respectively [25]. and uab and udc are the AC side voltage and the DC output voltage, respectively [25].

+ T1 T3

is Lm Rm a Cd RL

us uab udc b

T2 T4

Figure 5. Main circuit structure of single-phase voltage-source pulse width modulation (PWM) Figure 5. Main circuit structure of single-phase voltage-source pulse width modulation (PWM) converter with RC-IGBT. converter with RC-IGBT.

In the implementation of energy exchange, the direction of the current of the DC voltage source ﬂowing into the DC circuit and the direction of the DC voltage connecting to the AC side, namely the PWM converter in the converter or inverter working conditions, depending on the states of four switches in the main circuit [26]. With the traction condition as an example, Table1 gives eight working states of the desaturation pulse logic of the single-phase PWM converter. “Conventional pulse” represents four PWM pulses of a conventional PWM converter, and “desaturation pulse” represents four pulses of an RC-IGBT PWM converter. For example, “0110” means that the gate voltages of switches 2 and 3 are high, and the gate voltages of switches 1 and 4 are low. The desaturation pulse logic of RC-IGBT saturation at each state switching point is also given in the table. For example, “3 to 1” means switching from state 3 to state 1, and the T2 turns on and the T1 is reverse recovery. At the same time, the T1 is applied with a desaturation pulse. According to the working mode of the PWM converter with RC-IGBT gate desaturation control, Figure6 shows the desaturation control process of the single-phase PWM converter in the traction condition. The solid line boxes represent the alternating work processes of the rectiﬁcation working mode and the power supply short circuit mode, while the dashed boxes represent the alternating work processes of the inverter working mode and the power short circuit mode. The letters on the arrow represent the devices, which need to be desaturated in the corresponding working mode. At this point, Energies 2017, 10, 1845 7 of 17 a desaturation pulse is applied depending on the direction of the inductor current and the state of the switch on the opposite side. The letters around the digits correspond to devices in the diode mode that need to be applied to the low drive voltage. The operating states 4 to 5 and the working states 8 to 1 will lead to an inductance current zero crossing. In other words, the full devices will switch from diode mode to IGBT mode. In addition, the desaturation control process of the PWM converter in the regenerative condition is similar to that in the traction condition.

Table 1. Timing of desaturation pulse for RC-IGBT in different working states.

us is State Conventional Pulse Desaturation Pulse Conducting Switches

1 0110 0110 T2-IGBT,T3-IGBT 2 0101 0100 T2-IGBT,T4-DIODE 3 1010 0010 T1-DIODE,T3-IGBT 4 1001 0000 T ,T >0 1-DIODE 4-DIODE 3→1 T2 turned on T1 desaturated 4→2 T1 reverse recovery

2→1 T3 turned on T4 desaturated 4→3 T4 reverse recovery

5 1001 1001 T1-IGBT,T4-IGBT 6 1010 1000 T1-IGBT,T3-DIODE 7 0101 0001 T2-DIODE,T4-IGBT 8 0110 0000 T ,T <0 2-DIODE 3-DIODE 7→5 T1 turned on T2 desaturated 8→6 T2 reverse recovery

6→5 T4 turned on T3 desaturated Energies 2017, 10, 1845 8→7 T3 reverse recovery8 of 18

Figure 6. Process of the desaturation control of single-phase PWM converter in traction. Figure 6. Process of the desaturation control of single-phase PWM converter in traction. 3. Loss Threshold Desaturation Control of RC-IGBT PWM Converter 3. Loss Threshold Desaturation Control of RC-IGBT PWM Converter 3.1. Improved Predictive Current Control 3.1. ImprovedBased Predictive on the predictive Current Control current control of conventional PWM converter and the criterion of Basedinductance on the current, predictive an improved current predictive control current of conventional control for the PWMPWM converter, converter which and is thesuitable criterion of for the RC-IGBT is obtained [27–31]. The conventional single-phase PWM converter is controlled by inductance current, an improved predictive current control for the PWM converter, which is suitable the predictive current method. In a PWM period, the inductance current satisfies the following for theequation: RC-IGBT is obtained [27–31]. The conventional single-phase PWM converter is controlled by the predictive current method. In a PWM period, the inductance current satisfies the following equation: i(t+= T) i* (t) (12) sk s sk∗ is(tk + Ts) = i s(tk) (12) According to the analysis of the circuit topology, the following equation is established: * − * =− −L(ims i) s uuRiab s m s (13) Ts

**=− * − IKUusp()+1/() dc dc TUudt i dc dc (14)

=−Ls * − u(t)u(tab s)[ I s sinωtit s ()] (15) Ts

The four channels’ voltage generated by the PWM generator is processed by the logic of desaturation. The relations of the gate voltages are described as Formulas (16) and (17). uge_des1(t)~ uge_des4(t) are the four gate voltages after desaturation. The ε(t) is the step function. While the current is positive, ε[is(t)] = 1, and the current is reverse, ε[is(t)] = 0. fdelay and fdes are the delay function and the desaturation processing function, respectively. () () () uge _ des111 t ut ge ut ge =−{}1()ε i t ∗+f ε it ()∗ f  () sdelaysdes()  () (16) utge _ des444  ut ge  u ge t

()  () () utge _des22 ut ge ut ge 2  = ε i ()tf∗+− {} 1ε it () ∗ f  () sdelay()  s des () (17) utge _des33   u getut  ge 3 According to the above equations, the following control charts in Figure 7 are obtained:

Energies 2017, 10, 1845 8 of 17

According to the analysis of the circuit topology, the following equation is established: ∗ ∗ Lm(i s − is) u ab = us − Rmis − (13) Ts Z ∗ ∗ ∗ I s = Kp(U dc − udc) + 1/Ti (U dc − udc)dt (14)

Ls ∗ uab(t) = us(t) − [I ssinωt − is(t)] (15) Ts The four channels’ voltage generated by the PWM generator is processed by the logic of desaturation. The relations of the gate voltages are described as Formulas (16) and (17). uge_des1(t)~uge_des4(t) are the four gate voltages after desaturation. The ε(t) is the step function. While the current is positive, ε[is(t)] = 1, and the current is reverse, ε[is(t)] = 0. f delay and f des are the delay function and the desaturation processing function, respectively. " # " # " # uge_des1(t) uge1(t) uge1(t) = {1 − ε[is(t)]} ∗ fdelay + ε[is(t)] ∗ fdes (16) uge_des4(t) uge4(t) uge4(t) " # " # " # u (t) u (t) u (t) ge_des2 = ε[i (t)] ∗ f ge2 + {1 − ε[i (t)]} ∗ f ge2 (17) u (t) s delay u (t) s des u (t) Energies 2017, 10ge, 1845_des 3 ge3 ge3 9 of 18 AccordingEnergies 2017 to, 10, the 1845above equations, the following control charts in Figure7 are obtained:9 of 18 U*dc uab(t) I*s + - PWM U*dc PI Sin Lm/TS uab(t) I+*s -+ - GeneratorPWM PI Sin - Lm/TS + udc + Generator - + us dc - ε u (t) is(t) PLL ε[is(t)] s ε u (t) is(t) PLL ε[is(t)]

+ Uge_des1(t) 1-ε[is(t)] fdelay ge1 + UUge_des1ge_des4(t)(t) u (t) 1-ε[is(t)] fdelay + uge4uge1(t()t) + Uge_des4(t) uge4(t) ε[is(t)] fdes ε[is(t)] fdes + Uge_des2(t) ε[is(t)] delay f + Uge_des2(t) uge2(t) ε[is(t)] fdelay + Uge_des3(t) uge2(t) Uge_des3(t) uge3(t) + uge3(t) 1-ε[is(t)] fdes 1-ε[is(t)] fdes

Figure 7. Control block diagram of the improved predictive current control. FigureFigure 7. Control 7. Control block block diagram diagram of the improv improveded predictive predictive current current control. control.

FromFrom the the analysis analysis of of every every state state of of thethe PWMPWM rectifier inin Section Section 2.2, 2.2, provided provided that that the the direction direction ofFrom theof the inductor the inductor analysis current current of is every is determined, determined, state of the the working working PWM rectiﬁer mode ofof each ineach Section bridge bridge arm2.2 arm, can provided can be be accurately accurately that thejudged. judged. direction of the inductorFigureFigure 8 shows current8 shows the the is logic determined, logic diagram diagram of theof the the working desaturationdesaturation mode pulsepulse of each controlcontrol bridge of of the the armPWM PWM can converter converter be accurately with with RC- RC- judged. FigureIGBT.8IGBT. shows the logic diagram of the desaturation pulse control of the PWM converter with RC-IGBT.

Figure 8. Logic diagram of the desaturation pulse control of the PWM converter with RC-IGBT. FigureFigure 8. Logic 8. Logic diagram diagram of of the the desaturation desaturation pulse co controlntrol of of the the PWM PWM converter converter with with RC-IGBT. RC-IGBT. The inputs T1, T2, T3, and T4 in Figure 8 are four conventional input signals with a dead time of 5 μThes, and inputs the inductor T1, T2, T 3current,, and T 4i Lin, is Figure also an 8 input. are four The conventional four output signals input signalsare outputs with T a1, deadT2, T3 ,time and of 5 μTs,4. andThe thedirection inductor of thecurrent, input iLcurrent, is also i Lan is input.judged The firstly. four When output its signals direction are is outputs forward, T 1the, T2 ,pulse T3, and T4. generationsThe direction of theof theT1 and input T2 arecurrent analyzed iL is together.judged firstly. Because When the T its1 has direction been in isdiode forward, mode thein this pulse 1 generationsprocess, the of drivingthe T1 andvoltage T2 areof the analyzed T should together. be low. BecauseIn addition the to T meeting1 has been the in needs diode of modethe gate in of this the RC-IGBT in diode mode that is applied to the low voltage, the desaturation pulse must also be process, the driving voltage of the T1 should be low. In addition to meeting the needs of the gate of theimposed RC-IGBT to inreduce diode the mode diode that reverse is applied recovery to lo thess. Importantly,low voltage, the the rising desaturation edge of input pulse T 2must should also be be detected, the output T1 is obtained by the desaturation pulse procedure of input T2. At the same time, imposed to reduce the diode reverse recovery loss. Importantly, the rising edge of input T2 should be the output T2 can also be obtained by delaying the input T2 for the time tdelay. Similarly, the pulse detected, the output T1 is obtained by the desaturation pulse procedure of input T2. At the same time, generations of T3 and T4 are based on the desaturation pulse procedure and the delay of input T4. The the output T2 can also be obtained by delaying the input T2 for the time tdelay. Similarly, the pulse process of the reverse input current is treated similarly. generations of T3 and T4 are based on the desaturation pulse procedure and the delay of input T4. The process3.2. Desaturation of the reverse Control input of Losscurrent Decrement is treated Threshold similarly.

3.2. DesaturationIn the improved Control predictive of Loss Decrement current controlThreshold strate gy with the desaturation pulse for the single- phase PWM converter, the value and direction of the inductor current are the key to RCDC control, In the improved predictive current control strategy with the desaturation pulse for the single- phase PWM converter, the value and direction of the inductor current are the key to RCDC control,

Energies 2017, 10, 1845 9 of 17

The inputs T1,T2,T3, and T4 in Figure8 are four conventional input signals with a dead time of 5 µs, and the inductor current, iL, is also an input. The four output signals are outputs T1,T2,T3, and T4. The direction of the input current iL is judged ﬁrstly. When its direction is forward, the pulse generations of the T1 and T2 are analyzed together. Because the T1 has been in diode mode in this process, the driving voltage of the T1 should be low. In addition to meeting the needs of the gate of the RC-IGBT in diode mode that is applied to the low voltage, the desaturation pulse must also be imposed to reduce the diode reverse recovery loss. Importantly, the rising edge of input T2 should be detected, the output T1 is obtained by the desaturation pulse procedure of input T2. At the same time, the output T2 can also be obtained by delaying the input T2 for the time tdelay. Similarly, the pulse generations of T3 and T4 are based on the desaturation pulse procedure and the delay of input T4. The process of the reverse input current is treated similarly.

3.2. Desaturation Control of Loss Decrement Threshold In the improved predictive current control strategy with the desaturation pulse for the Energies 2017, 10, 1845 10 of 18 single-phase PWM converter, the value and direction of the inductor current are the key to RCDC control,and and directly directly affect affect the working the working and loss characteristics and loss characteristics of RC-IGBT. When of RC-IGBT. compared to When the operation compared to the operationat saturation at saturation mode, the loss mode, of RC-IGBT the loss in the of RCDC RC-IGBT control in decreases. the RCDC In a control given modulation decreases. period, In a given the performance of the RCDC control is closely related to the loss variation. When considering the modulation period, the performance of the RCDC control is closely related to the loss variation. shock of the inductor current error in zero crossing, as well as the error of the sensor detection, the When consideringRCDC control the in shockthe small of thecurrent inductor condition current will le errorad to ina mutation zero crossing, in the time as well sequence as the of error the of the sensor detection,desaturation the pulse, RCDC which control in turn, in may the cause small the current bridge arm condition to short willcircuit, lead and to pulse a mutation blockade inand the time sequencegreater of the harmonic. desaturation In addition, pulse, the which loss variation in turn, of may the causeRCDC thecontrol bridge is not arm obvious to short under circuit, the small and pulse blockadecurrent and condition. greater harmonic. In order to reduce In addition, the fault therisk lossthat is variation caused by ofthe the triggering RCDC of control the desaturation is not obvious under thepulse, small and currentto utilize condition. the loss suppression In order of to large reduce current the conditions, fault risk thata desaturation is caused control by the scheme triggering of based on the loss decrement threshold per-unit value is proposed. According to the scheme, as the desaturation pulse, and to utilize the loss suppression of large current conditions, a desaturation compared to the RC-IGBT saturation control, the loss decrement of RC-IGBT in the RCDC control is controlchanged scheme little based near on the the zero loss crossing decrement of the threshold current. The per-unit loss decrement value is proposed.changes largely According in the to the scheme,moment, as compared thus the to working the RC-IGBT current saturationis great. By control,setting the the threshold loss decrement of the loss ofdecrement RC-IGBT per-unit in the RCDC controlvalue is changed corresponding little near a small the current zero crossing value, the of RCDC the current. control is The executed lossdecrement when the actual changes per-unit largely in the moment,value of thus the loss the decrement working currentis below isthe great. threshold, By setting and is not the executed threshold when of it the exceeds loss the decrement threshold. per-unit value correspondingWe set the threshold a small currentof the loss value, decrement the RCDC per-unit control value equal is executed to ΔEth*, when the the actual actual per- per-unit unit value of the loss decrement is less than or equal to ΔEth*, the inductor current is corresponding value of the loss decrement is below the threshold, and is not executed when it exceeds the threshold. to [−ILmax, −ILth] and [ILth, ILmax], and the RCDC control is performed. When the actual per-unit value of Wethe set loss the decrement threshold is ofgreater the loss than decrement ΔEth*, then the per-unit inductor value current equal is corresponding to ∆Eth*, when to [− theILth, actualILth] and per-unit value ofthe the RCDC loss control decrement is not isperformed. less than ILmax or equalis the peak to ∆ Evalueth*, theof the inductor inductor current current, isand corresponding ILth is the to [−ILmaxcurrent, −ILth threshold] and [ILth corresponding,ILmax], and to the the RCDCloss threshol controld. Figure is performed. 9 shows the When logic block the actual diagram per-unit of the value desaturation control for the threshold of the loss decrement per-unit value corresponding to the of the loss decrement is greater than ∆Eth*, then the inductor current is corresponding to [−ILth,ILth] inductance current zero-crossing threshold value. and the RCDC control is not performed. I is the peak value of the inductor current, and I is For the selection of the loss thresholdLmax decrement per-unit value, one should refer to the specific Lth the currentenvironment threshold in which corresponding the device is to applied the loss and threshold. the working Figure current9 shows flowing the through logic it. block When diagram the of the desaturationdevice is operating control within for the the threshold rated current of the range, loss the decrement 10% peak per-unitcurrent is valueselected corresponding as the current to the inductancethreshold current ILth. zero-crossing threshold value.

Figure 9.FigureLogic 9. Logic block block diagram diagram of ofthe the desaturation contro controll for the for threshold the threshold of the loss of decrement the loss per- decrement per-unitunit value. value.

4. Simulation and Experiment For the selection of the loss threshold decrement per-unit value, one should refer to the speciﬁc environment4.1. Simulation in which Analysis the device is applied and the working current ﬂowing through it. When the In order to verify the feasibility of the desaturation pulse control in the PWM converter with conventional IGBT, the simulation model was built in Matlab Simulink 8.6, R2015B (MathWorks, Natick, MA, USA). The controller adopted a double closed loop approach that was based on the improved predictive current control. The Table 2 shows the simulation parameters of the main circuit originated from the design parameters of the locomotive four quadrant converter of China Railway High-speed 2 (CRH2):

Energies 2017, 10, 1845 10 of 17 device is operating within the rated current range, the 10% peak current is selected as the current threshold ILth.

4. Simulation and Experiment

4.1. Simulation Analysis In order to verify the feasibility of the desaturation pulse control in the PWM converter with conventional IGBT, the simulation model was built in Matlab Simulink 8.6, R2015B (MathWorks, Natick, MA, USA). The controller adopted a double closed loop approach that was based on the improved predictive current control. The Table2 shows the simulation parameters of the main circuit originated from the design parameters of the locomotive four quadrant converter of China Railway High-speed 2 (CRH2):

Table 2. Design parameters of locomotive four quadrant converter of China Railway High-speed 2 (CRH2). Energies 2017, 10, 1845 11 of 18

Table 2.Parameter Design parameters Name of locomotive four quadrant Signal converter of China Railway Value High-speed 2

(CRH2).AC side rated voltage UN 1950 V AC sideParameter rated frequency Name SignalfN Value50 Hz DC side output voltage U 3500 V AC side rated voltage UN d 1950 V OperatingAC side rated frequency frequency fN fs 50 Hz450 Hz ACDC side side inductance output voltage UdL m 35002.195 V mH ACOperating side resistor frequency fsR m 450 Hz0.01 Ω DC side supportAC side inductance capacitance LmC d 2.195 mH5 mF FilterAC inductance side resistor RmL 2 0.011.27 Ω mH DCFilter side capacitance support capacitance CdC 2 5 mF 2 mF LoadFilter resistor inductance L2 R 1.27 mH8.75 Ω Filter capacitance C2 2 mF Load resistor R 8.75 Ω Figure 10 shows the models of the predictive current control of the conventional IGBT PWM rectiﬁer andFigure the improved 10 shows the predictive models of current the predictive control current with desaturationcontrol of the conventional pulses of the IGBT RC-IGBT PWM PWM rectifier and the improved predictive current control with desaturation pulses of the RC-IGBT PWM rectiﬁer under the same simulation parameters. rectifier under the same simulation parameters.

Figure 10.FigureThe 10. simulation The simulation circuit circuit of of the the single-phase single-phase PWM PWM rectifier rectiﬁer under under saturation saturation control control and and desaturationdesaturation control. control.

The simulation results are shown in Figure 11. The output voltage udc, the grid voltage us and the inductor current is are shown under different operations: (a) the conventional IGBT PWM rectifier; (b) the RC-IGBT PWM rectifier under the desaturation control of the current threshold 10%; and, (c) the RC-IGBT PWM rectifier under the all period desaturation control. It can be seen from the diagram that the three control methods can all ensure the DC output stable at 3500 V. The voltage and current of the grid keep synchronous, using fast Fourier transformation (FFT) analysis, the grid current total harmonic distortion (THDs) are 2.02% under saturation control, 2.01% under desaturation control of the current threshold 15%, and 2.01% under all period desaturation control. The system works stably in the rectifier state.

Energies 2017, 10, 1845 11 of 17

The simulation results are shown in Figure 11. The output voltage udc, the grid voltage us and the inductor current is are shown under different operations: (a) the conventional IGBT PWM rectifier; (b) the RC-IGBT PWM rectifier under the desaturation control of the current threshold 10%; and, (c) the RC-IGBT PWM rectifier under the all period desaturation control. It can be seen from the diagram that the three control methods can all ensure the DC output stable at 3500 V. The voltage and current of the grid keep synchronous, using fast Fourier transformation (FFT) analysis, the grid current total harmonic distortion (THDs) are 2.02% under saturation control, 2.01% under desaturation control of the current threshold 15%, and 2.01% under all period desaturation control. The system works stably in the rectifierEnergies 2017 state., 10, 1845 12 of 18

(a)

us/V is/A udc/V 4000 3000 2000 1000 0 －1000 －2000 －3000 0.4 0.42 0.44 0.46 0.48 0.5 t/s

(b) (c)

FigureFigure 11. Steady-state 11. Steady-state waveforms: waveforms: output output voltagevoltage uudcdc, grid, grid voltage voltage us andus andthe inductor the inductor current currentis is under (undera) conventional (a) conventional IGBT; IGBT; (b) ( allb) all period period desaturation desaturation control; control; (c) (desaturationc) desaturation control control of the current of the current thresholdthreshold 10%. 10%.

AccordingAccording to the to simulation,the simulation, the the average average current value value of ofthe the device device in a switching in a switching cycle could cycle could be calculated. Referring to the device parameter manual, the total loss of a modulation cycle under be calculated. Referring to the device parameter manual, the total loss of a modulation cycle under rated conditions can be calculated. The following Table 3 shows the loss data of the four conditions: rated conditionsconventional can IGBT, be calculated.RC-IGBT saturation The following control, RC-IGBT Table3 shows current the threshold loss data desaturation of the four control, conditions: conventionaland RC-IGBT IGBT, full RC-IGBT cycle desaturation saturation control. control, Due RC-IGBT to the different current characteristics threshold desaturationof the conventional control, and RC-IGBTIGBT, full all cycle of the desaturation partial loss and control. the total Due loss toof thethe desaturation different characteristics controlled RC-IGBT of the are conventional decreased. IGBT, all of theThe partial IGBT lossturn-on and loss, the the total diode loss conduction of the desaturation loss and reverse controlled recovery RC-IGBT loss of the are RC-IGBT decreased. under The IGBT turn-onsaturation loss, the control diode are conduction increased. lossAdditionally, and reverse there recoveryis no difference loss ofin thethe RC-IGBTtotal loss between under the saturation desaturation control of the current threshold 10% and the whole cycle desaturation control. control are increased. Additionally, there is no difference in the total loss between the desaturation control of theTable current 3. Datasheet threshold of loss 10%for conventional and the whole insulate cycled gate desaturationbipolar transistor control. (IGBT) and RC-IGBT As shownPWM inconverters. Figure 12, the partial loss and the total loss of the system are compared under different loads.PWM When Converte comparedr toEcon_IGBT the conventional/mJ Eon/mJ IGBT,Eoff/mJ the totalEon_diode loss/mJ of RC-IGBTErec/mJ single-phaseEtot/mJ PWM rectifier underCon-IGBT the saturation *1 control 3520 increases 24,160 by about 15,680 5.2% under11,950 three different13,040 loads,68,350 respectively. 2 The total lossRC-IGBT(SAT) of RC-IGBT * single-phase 3420 PWM 26,880 rectifier 15,620 under all12,510 of the period13,480 desaturation 71,910 control RC-IGBT(RCDC) *3 3420 17,340 15,620 11,510 12,430 60,320 decreases byRC-IGBT(THDC) 11.7% under *4 the rated 3420 load, 10.8% 17,500 under 15,620 the half load,11,590 and 12.5%12,590 under60,520 the 1.5 times load. In otherNote: words,*1 expresses under conventional the condition IGBT; *2 expresses of a RC-IGBT large load saturation and control high (SAT); load * current,3 expresses RC- the loss of the RC-IGBT underIGBT all the period desaturation desaturation control control; decreased*4 expresses RC-IGBT more significantly. desaturation control When of thethe current threshold is different,threshold the loss (THDC) suppression 10%. is different. So it is necessary to adjust the setting according to the actual workingAs shown conditions. in Figure 12, the partial loss and the total loss of the system are compared under different loads. When compared to the conventional IGBT, the total loss of RC-IGBT single-phase PWM rectifier under the saturation control increases by about 5.2% under three different loads, respectively. The total loss of RC-IGBT single-phase PWM rectifier under all of the period desaturation control decreases by 11.7% under the rated load, 10.8% under the half load, and 12.5%

Energies 2017, 10, 1845 12 of 17

Table 3. Datasheet of loss for conventional insulated gate bipolar transistor (IGBT) and RC-IGBT PWM converters.

PWM Converter Econ_IGBT/mJ Eon/mJ Eoff/mJ Eon_diode/mJ Erec/mJ Etot/mJ Con-IGBT *1 3520 24,160 15,680 11,950 13,040 68,350 EnergiesRC-IGBT(SAT) 2017, 10, 1845 *2 3420 26,880 15,620 12,510 13,480 71,91013 of 18 RC-IGBT(RCDC) *3 3420 17,340 15,620 11,510 12,430 60,320 underRC-IGBT(THDC) the 1.5 times *4 load. In3420 other words, 17,500 under the condition 15,620 of a 11,590 large load and 12,590 high load current, 60,520 the loss of the RC-IGBT under the desaturation control decreased more significantly. When the Note: *1 expresses conventional IGBT; *2 expresses RC-IGBT saturation control (SAT); *3 expresses RC-IGBT all periodcurrent desaturation threshold control; is different, *4 expresses the lo RC-IGBTss suppression desaturation is different. control So of it the is currentnecessary threshold to adjust (THDC) the setting 10%. according to the actual working conditions. 120,000 Erec 105.3% Eon_diode 100% 87.7% 100,000 Eoff 87.5% Eon 80,000 Econ_IGBT 105.2% 100% 88.5% 88.3% 60,000 105.1% 40,000 100% 89.5% 89.2%

20,000

0 *1 *2 *3 *4 *1 *2 *3 *4 *1 *2 *3 *4 0.7MW 1.4MW 2.1MW

Figure 12.FigureTotal 12.loss Total comparison loss comparison between between conventional conventional IGBT IGBT and and RC-IGBT RC-IGBT atat differentdifferent loads. (Note: (Note: PWM ConvertersPWM under Converters four typeunder conditions four type conditions *1,*2,*3 and*1, *2,* *43 inand Figure *4 in Figure 12 are 12 corresponding are corresponding to to those those of of Table 3). Table 3). By changing the current threshold, the single-phase PWM converter is simulated under different load currents,By changing and the the average current threshold, loss in the the positive single-phase half PWM cycle converter is calculated. is simulated Table under4 shows different that the load currents, and the average loss in the positive half cycle is calculated. Table 4 shows that the load load current IL was, respectively, 0.2, 0.5, 0.8, and 1.5 IN, the percentage of the current threshold current IL was, respectively, 0.2, 0.5, 0.8, and 1.5 IN, the percentage of the current threshold IPU (the IPU (theratio ratio of ofthe thecurrent current threshold threshold to the peak to the current) peak was current) equal wasto zero equal (complete to zero judgment (complete on zero- judgment on zero-crossing),crossing), 0.1 0.1to 0.8, to 1 0.8, (saturation 1 (saturation control), control), the average the loss average decrement loss per-unit decrement value in per-unit the half valuecycle in the half cycleis Δ isE* ∆(theE* ratio (the of ratio the difference of the difference of the average of the loss averagebetween the loss saturation between control the saturationand desaturation control and control to saturation control). The IN is 500 A. desaturation control to saturation control). The IN is 500 A.

Table 4. Average loss decrement per-unit value in half cycle of single-phase PWM converter under Table 4. Average loss decrement per-unit value in half cycle of single-phase PWM converter under different current thresholds and different load currents. different current thresholds and different load currents. IL\IPU 0 10% 20% 30% 40% 50% 60% 70% 80% 1 0.2 IN 0.1400 0.1317 0.1080 0.0894 0.0867 0.0834 0.0795 0.0603 0.0516 0 IL\IPU 0 10% 20% 30% 40% 50% 60% 70% 80% 1 0.5 IN 0.1513 0.1437 0.1319 0.1205 0.1177 0.1119 0.1040 0.0764 0.0608 0 0.2 IN 0.8 IN0.1400 0.15450.1317 0.1506 0.1080 0.1459 0.0894 0.1343 0.0867 0.1211 0.1212 0.0834 0.1155 0.0795 0.0820 0.0603 0.0625 0.0516 0 0 0.5 IN IN 0.1513 0.16060.1437 0.1544 0.1319 0.1493 0.1205 0.1384 0.1177 0.1332 0.1238 0.1119 0.1207 0.1040 0.0868 0.0764 0.0681 0.0608 0 0 0.8 IN 1.5 IN0.1545 0.16900.1506 0.1619 0.1459 0.1515 0.1343 0.1396 0.1211 0.1338 0.1261 0.1212 0.1233 0.1155 0.0859 0.0820 0.0669 0.0625 0 0 IN 0.1606 0.1544 0.1493 0.1384 0.1332 0.1238 0.1207 0.0868 0.0681 0 1.5 IN By0.1690 using the 0.1619 MATLAB 0.1515 three spline 0.1396 interpolat 0.1338ion to set 0.1261 up the binary 0.1233 relation 0.0859 among 0.0669 IL, IPU and 0 ΔE*, the three-diensional (3D) graphs of which are shown in Figure 13. When the IPU is less than 0.1, the ΔE* does not change significantly; when the IPU is higher than 0.1, with the decrease of IPU, the By using the MATLAB three spline interpolation to set up the binary relation among IL,IPU and ΔE* increases significantly. Taking the rated load as an example, when the IPU is 0.1, the maximum ∆E*, thevalue three-diensional 0.16 of ΔE* is (3D) obtained. graphs When of which considering are shown repeatability in Figure and 13. Whensmall current the IPU lowis less loss than 0.1, the ∆E*characteristics does not change near significantly;the zero current when of the the load IPU current,is higher the optimal than 0.1, graph with of thethe decreaseloss decrement of IPU of, the ∆E* increasesthe significantly. PWM converter Taking is drawn, the namely rated load IPU 10%, as an the example, curve of Δ whenE* and theIL. In IPU addition,is 0.1, theFigure maximum 13 also value 0.16 of ∆showsE* is obtained.the curve of When ΔE* and considering IPU when IL repeatability is IN. The intersection and small point current of the lowtwo losscurves characteristics is the best near the zeroworking current point of the of RC-IGBT load current, single-phase the optimal PWM rectifier graph under of the rated loss conditions. decrement of the PWM converter is drawn, namely IPU 10%, the curve of ∆E* and IL. In addition, Figure 13 also shows the curve of ∆E* and IPU when IL is IN. The intersection point of the two curves is the best working point of RC-IGBT single-phase PWM rectifier under rated conditions.

Energies 2017, 10, 1845 13 of 17 Energies 2017, 10, 1845 14 of 18

Energies 2017, 10, 1845 14 of 18

Figure 13.FigureSurface 13. Surface graph graph of the of the variety variety of of average average loss de decrementcrement per per unit unit value value with load with current loadcurrent and and current threshold percentage. current threshold percentage. 4.2. ExperimentalFigure 13. Surface Results graph of the variety of average loss decrement per unit value with load current and 4.2. Experimental Results Incurrent order threshold to further percentage. verify the design method above, the experimental study is carried out. The grid voltage is reduced by the voltage regulator to 195 V from 220 V. The experimental parameters In order4.2. Experimental to further Results verify the design method above, the experimental study is carried out. The grid are Us = 195 V, fN = 50 Hz, Ud* = 350 V, and switching frequency fS = 1 kHz. The PWM converter model voltage is reduced by the voltage regulator to 195 V from 220 V. The experimental parameters are is madeIn order up of to a halffurther bridge verify switching the design devices method for Infineonabove, the FZ1000R65KR3, experimental studyand another is carried half out. bridge The Us = 195 V, f = 50 Hz, U * = 350 V, and switching frequency f = 1 kHz. The PWM converter model is forgrid Infineon voltageN FZ750R65KE3 is reducedd by [32].the voltage The inductance regulator L mto is 195 12.8 V mH from andS 220 the V. capa Thecitors experimental are two parallel parameters and made uptwoare of U series as = half 195 capacitors, V, bridge fN = 50 switchingHz, namely Ud* = 470350 devices μV,F/450 and switchingV. for The Inﬁneon digital frequency signal FZ1000R65KR3, f Sprocessing = 1 kHz. The (DSP) andPWM in another converterthe controller half model bridge is for InﬁneonTexasis FZ750R65KE3 made Instruments up of a half [(TI)32 bridge]. TMS320LF28335 The switching inductance devices (TexasLm isforInstru 12.8 Infineonments mH andFZ1000R65KR3,(TI), theDallas, capacitors TX, USA),and another are to mainly two half parallel achieve bridge and two series capacitors,thefor Infineonimprovednamely FZ750R65KE3 current 470 predictiveµ [32].F/450 The control V. inductance The algori digital Lthm,m is signal 12.8and mH the processing andcomplex the capa programmable (DSP)citors are in thetwo controllerlogicparallel device andis Texas Instruments(CPLD)two series (TI) is capacitors, TMS320LF28335ALTERA’s namely EPM7128AETC100-7 470 (Texas μF/450 Instruments V. The (Altera, digital (TI),Silicon signal Dallas, processingValley, TX, CA, (DSP) USA), USA) in to theto mainly controllerrealize achievethe is the desaturationTexas Instruments control (TI) algorithm. TMS320LF28335 Figure 14 (Texas shows Instru the testments platform. (TI), Dallas, TX, USA), to mainly achieve improved current predictive control algorithm, and the complex programmable logic device (CPLD) is the improved current predictive control algorithm, and the complex programmable logic device ALTERA’s(CPLD) EPM7128AETC100-7 is ALTERA’s EPM7128AETC100-7 (Altera, Silicon (Altera, Valley, Silicon CA, USA) Valley, to realizeCA, USA) the desaturationto realize the control algorithm.desaturation Figure 14 control shows algorithm. the test Figu platform.re 14 shows the test platform.

Figure 14. Test platform for RC-IGBT PWM convertor.

The steady state experimental waveform PWM converter is under saturation control and all period desaturation control.Figure As show 14. Testn in platform Figure for15, RC-IGBTthe output PWM voltage convertor. udc under conventional control Figure 14. Test platform for RC-IGBT PWM convertor. is about 350 V, which is 1.79 times the grid voltage, and the ripple voltage is small. Although the switchingThe steady frequency state is experimental low, the AC sidewaveform current PWM waveform converter is still is close under to saturationthe sinusoidal control waveform, and all The steady state experimental waveform PWM converter is under saturation control and all withperiod a powerdesaturation factor control.equal to As 1. showAt then insame Figure time 15,, under the output saturation voltage control, udc under the conventional output voltage control and period desaturationoutputis about current 350 V, control. waveformwhich is As 1.79 of shown thetimes PWM the inFigure gridconverter voltage, 15, thear eand almost output the ripplethe voltage same voltage uasdc those isunder small. under conventional Although saturation the control is aboutcontrol.switching 350 V, Through which frequency isthe 1.79 isgrid low, timesside the current AC the side grid waveform current voltage, waveform data and are analyzed theis still ripple close by to voltagethe the MATLAB sinusoidal is small. FFT waveform, function, Although the switching with frequency a power factor is low, equal the to AC1. At side the currentsame time waveform, under saturation is still control, close to the the output sinusoidal voltage waveform,and with a poweroutput factorcurrent equalwaveform to 1. of At the the PWM same converter time, underare almost saturation the same control, as those the under output saturation voltage and control. Through the grid side current waveform data are analyzed by the MATLAB FFT function, output current waveform of the PWM converter are almost the same as those under saturation control.

Through the grid side current waveform data are analyzed by the MATLAB FFT function, the THD of the gird side current is 4.76% under saturation control. Those of desaturation pulse width 50 µs are Energies 2017,, 10,, 18451845 15 of 18 EnergiesEnergies 20172017,, 1010,, 18451845 1415 of 1718 the THD of the gird side current is 4.76% under saturation control. Those of desaturation pulse width thethe THDTHD ofof thethe girdgird sideside currentcurrent isis 4.76%4.76% underunder saturationsaturation control.control. ThoseThose ofof desaturationdesaturation pulsepulse widthwidth 50 μs are 4.60% under all period desaturation control. The results prove that the desaturation control 504.60%50 μμss areare under 4.60%4.60% all under periodunder allall desaturation periodperiod desaturationdesaturation control. The control.control. results TheThe prove resultsresults that proveprove the desaturation thatthat thethe desaturationdesaturation control is effective controlcontrol is effective and feasible. isandis effectiveeffective feasible. andand feasible.feasible.

(a) (b) ((aa)) ((bb))

FigureFigure 15.15. Steady-stateSteady-state experimental experimental waveforms: waveforms: the the grid grid voltage voltageus, output uss,, outputoutput voltage voltagevoltageu and u griddc andand current gridgrid Figure 15. Steady-state experimental waveforms: the grid voltage us, output voltagedc udc and grid currentis,(a) saturation iiss,, ((a) saturation control, control, THD = THD 4.76%; = 4.76%; (b) all ( periodb) all period desaturation desaturation control, control,tdesat = ttdesat 50 ==µ s,5050t μlocks, =tlocklock 3 ==µ 33s, current is, (a) saturation control, THD = 4.76%; (b) all period desaturation control, tdesat = 50 μs, tlock = 3 μTHDs, THD = 4.60%. = 4.60%. μμs,s, THDTHD == 4.60%.4.60%.

FigureFigure 16 compares the gate voltage waveforms of T1 andand T T2 underunder saturationsaturation controlcontrol andand allall Figure 16 compares the gate voltage waveforms of T1 and T2 under saturation control and all period desaturation control.control. ItIt cancan bebe seen seen from from Figure Figure 16 16bb that that T 1Tis1 isis desaturated desaturateddesaturated and andand the thethe gate gategate pulse pulsepulse of period desaturation control. It can be seen from Figure 16b that T1 is desaturated and the gate pulse ofT2 Tis2 unchangedisis unchangedunchanged under underunder desaturation desaturationdesaturation control controlcontrol when whenwhen the grid thethe current gridgrid currentcurrent is positive. isis positive.positive. When the WhenWhen grid currentthethe gridgrid is of T2 is unchanged under desaturation control when the grid current is positive. When the grid currentreverse, is T 2reverse,is desaturated T2 isis desaturateddesaturated and the gate andand pulse thethe gategate of T pulsepulse1 is unchanged. ofof TT1 isis unchanged.unchanged. current is reverse, T2 is desaturated and the gate pulse of T1 is unchanged. u uge-T1 uge-T1

T1 T1 T1 ge- ge- u (10V/div) ge- u (10V/div) u (10V/div) T2 T2

T2 i ge- is ge- s u

(10V/div) i ge- s u (10V/div) u (10V/div) s s i i s i (5A/div) (5A/div) (5A/div) u uge-T2 uge-T2 tt(4ms/div)(4ms/div) t(4ms/div) (a) (b) ((aa)) ((bb)) Figure 16. Steady-state experimental waveforms: the gate voltages uge of the T1 and T2 and the inductor Figure 16. Steady-stateSteady-state experimentalexperimental waveforms:waveforms: thethe gategate voltagesvoltages uge ofof thethe TT1 andand TT2 andand thethe inductorinductor Figure 16. Steady-state experimental waveforms: the gate voltages uge of the T1 and T2 and the inductor current, (a) saturation control; (b) all period desaturation control, tdesat = 50 µs, tlock = 3 µs. current, (a) saturation control; (b) all period desaturation control, tdesat == 5050 μs, ttlocklock == 33 μs. current, (a) saturation control; (b) all period desaturation control, tdesat = 50 μs, tlock = 3 μs.

Figure 1717 isis an an enlargement enlargement of of th thee positive positive half half period period of the of thecurrent current in Figure in Figure 17b. 17Theb. gate The level gate FigureFigure 1717 isis anan enlargementenlargement ofof ththee positivepositive halfhalf periodperiod ofof thethe currentcurrent inin FigureFigure 17b.17b. TheThe gategate levellevel oflevel the ofT1 the hashas T aa1 shortshorthas a pulsepulse short before pulsebeforebefore gategate voltagevoltage gate voltage ofof thethe ofTT2 theisis highhigh T2 is everyevery high time,time, every i.e.,i.e., time, thethe i.e., desaturationdesaturation the desaturation pulse.pulse. of the T1 has a short pulse before gate voltage of the T2 is high every time, i.e., the desaturation pulse. Figurepulse. Figure17a shows 17a showsthat the that desaturation the desaturation pulse width pulse widthof 50 μ ofs, 50andµs, Figure and Figure 17b shows 17b shows the interlock the interlock time FigureFigure 17a17a showsshows thatthat thethe desaturationdesaturation pulsepulse widthwidth ofof 5050 μμs,s, andand FigureFigure 17b17b showsshows thethe interlockinterlock timetime oftime 3 μ ofs. Therefore, 3 µs. Therefore, the experimental the experimental parameters parameters are comp areletely completely consistent consistent with those with in the those theoretical in the ofof 33 μμs.s. Therefore,Therefore, thethe experimentalexperimental parametersparameters areare compcompletelyletely consistentconsistent withwith thosethose inin thethe theoreticaltheoretical design.theoretical design. design.design.

(a) (b) ((aa)) ((bb)) Figure 17. Parameters under desaturation control: (a) desaturation pulse width 50 μs; (b) lock time FigureFigure 17. 17. ParametersParameters under underunder desaturation desaturationdesaturation control: control:control: (a )((aa desaturation)) desaturationdesaturation pulse pulsepulse width widthwidth 50 µ 5050s; (μbμs;)s; lock ((bb)) lock timelock time 3timeµs. 3 μs. 33 μμs.s.

Energies 2017, 10, 1845 15 of 17 Energies 2017, 10, 1845 16 of 18

Figure 18 shows the gate voltage waveforms of T1 and T2 under the condition of (a) the current Figure 18 shows the gate voltage waveforms of T1 and T2 under the condition of (a) the current threshold = 2 2 A A corresponding to to a a loss decrement th thresholdreshold per-unit value of 0.13, (b) the current threshold == 1010 A A corresponding corresponding to to a loss a loss decrement decrement threshold threshold per-unit per-unit value value of 0.08, of and 0.08, (c) and the current(c) the currentthreshold threshold = 13 A corresponding = 13 A corresponding to a loss decrementto a loss decrement threshold per-unitthreshold value per-unit of 0.05. value The of greater 0.05. The the greater the current threshold is, then the shorter the desaturation times of T1 and T2 will be. The PWM current threshold is, then the shorter the desaturation times of T1 and T2 will be. The PWM converters converterscan all work can steadily all work under steadily the different under current the differ thresholds.ent current As thethresholds. threshold As current the threshold increases, current the loss increases,decrement the threshold loss decrement per-unit value threshold gradually per-unit decreases. value Figure gradually 18a is thedecreases. waveform Figure of the 18a condition is the waveformthat has the of best the losscondition characteristics. that has the best loss characteristics.

(a) (b)

(c) Figure 18. Steady-state experimental waveforms under different loss thresholds: (a)I = 2 A, Figure 18. Steady-state experimental waveforms under different loss thresholds: (a) Ith = 2 thA, ΔE* = ∆E* = 0.13; (b)I = 10 A, ∆E* = 0.08; (c)I = 13 A, ∆E* = 0.05. 0.13; (b) Ith = 10 A,th ΔE* = 0.08; (c) Ith = 13 A,th ΔE* = 0.05.

5. Conclusions Due to the fact that it has its own low low thermal thermal resistance resistance and and high high current current density, density, the RC-IGBT satisfiedsatisﬁed thethe highhigh capacitycapacity single-phase single-phase PWM PWM converter converter requirements, requirements, namely namely high high power power density density and andreliability. reliability. In this In paper, this paper, the gate the desaturation gate desaturation control control of the diodeof the integrateddiode integrated in the RC-IGBT in the RC-IGBT is studied is studiedand the desaturationand the desaturation control of control switching of switching devices of devices the full of bridge the full circuit bridge is realized. circuit is The realized. improved The predictiveimproved currentpredictive control current of the control PWM of converter the PWM is co studied.nverter The is studied. loss decrement The loss threshold decrement desaturation threshold desaturationcontrol method control applied method to the applied RC-IGBT to single-phasethe RC-IGBT PWM single-phase converter PWM is proposed. converter In is order proposed. to avoid In orderthe desaturation to avoid the pulse desaturation chaos in pu thelse current chaos in zero the crossing, current zero by setting crossing, the by loss setting decrement the loss threshold decrement to thresholdlimit the current to limit threshold the current corresponding threshold corresponding to the desaturation to the pulsedesaturation applied pulse to RC-IGBT, applied the to RC-IGBT, operating performancethe operating of performance the PWM converter of the PWM in low converter switching in frequency low switching is improved. frequency Finally, is improved. the modeling Finally, and thesimulation modeling of theand PWM simulation converter of ofthe the PWM RC-IGBT converter are completed, of the RC-IGBT and the simulation are completed, results and conﬁrm the simulationthe effectiveness results of confirm the desaturation the effectiveness control. Experimentsof the desaturation are carried control. out onExperiments a small power are experimentcarried out platformon a small to power verify experiment the effectiveness platform of the to controlverify the strategy. effectiveness of the control strategy.

Acknowledgments: Acknowledgments: In thisthis paper,paper, the the samples samples and and test test data data are are recorded recorded with with the the strong strong support support of Toshiba of Toshiba and Inﬁneon Beijing Co. We wish to express our heartfelt thanks to them. The research is also supported by “the andFundamental Infineon Beijing Research Co. Funds We wish for to the express Central our University heartfelt (2017JBM061)”.thanks to them. The research is also supported by “the Fundamental Research Funds for the Central University (2017JBM061)”.

Author Contributions: Xianjin HUANG and Trillion Q. Zheng conceived and designed the experiments; Dengwei Chang and Chao Ling performed the experiments; Xianjin HUANG analyzed the data; Trillion Q. Zheng contributed materials and analysis tools; Xianjin HUANG wrote the paper.

Energies 2017, 10, 1845 16 of 17

Author Contributions: Xianjin Huang and Trillion Q. Zheng conceived and designed the experiments; Dengwei Chang and Chao Ling performed the experiments; Xianjin Huang analyzed the data; Trillion Q. Zheng contributed materials and analysis tools; Xianjin Huang wrote the paper. Conﬂicts of Interest: The authors declare no conﬂict of interest.

References

1. Wen, J.P. Research on the Control Strategies of Single-Phase Three-Level Pwm Rectifier; Beijing Jiaotong University: Beijing, China, 2012; pp. 1–96. 2. Song, K.J. Research and Implementation of Three Phase Current Source Rectifier; Nanjing University of Aeronautics and Astronautics: Nanjing, China, 2014; pp. 1–99. 3. Liu, B. Constant-Frequency-Hysteresis Control Strategy for Single Phase PWM Rectifiers; Southwest Jiaotong University: Chengdu, China, 2016; pp. 1–75. 4. Zhang, Y.C.; Peng, Y.B.; Qu, C.Q. Predictive current control of PWM rectifier under unbalanced grid voltage condition. Trans. China Electrotech. Soc. 2016, 31, 88–94. 5. Song, W.S.; Deng, Z.X. Model predictive power control scheme for single-phase PWM rectifiers with constant switching frequency. Electr. Mach. Control 2016, 20, 93–100. 6. Yin, K.; Ma, M.Y.; Cao, M.H.; He, Y.M.; Deng, Y.; Hen, X.N. Research on three-phase hybrid-clamped five-level PWM rectifier. Proc. CSEE 2012, 32, 59–66. 7. Huang, H. The Research of Single-Phase PWM Rectifier and Control Strategy; Huazhong University of Science & Technology: Wuhan, China, 2012; pp. 1–66. 8. Yang, L.Y.; Yang, S.; Zhang, W.P.; Chen, Z.G.; Long, X.U. An improved deadbeat predictive current control method for single-phase PWM rectifiers. Proc. CSEE 2015, 35, 5842–5850. 9. Deng, Z.X.; Song, W.S.; Cao, M.H. A model predictive current control scheme for single-phase PWM rectifiers. Proc. CSEE 2016, 36, 2992–3004. 10. Wei-Zhong, C.; Ze-Hong, L.; Bo, Z.; Min, R.; Jin-Ping, Z.; Yong, L.; Zhao-Ji, L. A snapback suppressed reverse-conducting IGBT with uniform temperature distribution. Chin. Phys. B 2014, 23, 513–518. 11. Qiu, H. T-Type Three-Level Inverter with Reverse Blocking IGBT; Beijing Jiaotong University: Beijing, China, 2015; pp. 1–80. 12. Li, X.P.; Liu, J.; Cao, G.; Gao, M.C.; Wang, Y.H.; Jin, R.; Wen, J.L. Development and application of reverse conducting-IGBT in smart grid. Smart Grid 2017, 5, 1–8. 13. Mitsubishi Datasheet: CM1000HG-130XA IGBT Module Data Sheet; Mitsubishi Electric Corporation: Tokyo, Japan, 2017. 14. Huang, X.J.; Ling, C.; Sun, H.; You, X.J.; Xiao, C.J. Working characteristics of Reverse-Conducting IGBT applied in vehicle converters. Semiconduct. Technol. 2017, 5, 363–370. 15. Fock-Sui-Too, J.L.; Chauchat, B.; Nicolau, S.; Madrid, F.; Austin, P.; Tounsi, P.; Mermet-Guyennet, M. Experimental study of power IGBT technologies at large range temperature. In Proceedings of the International Conference on Mixed Design of Integrated Circuits and Systems IET, Poznan, Poland, 19–21 June 2008; pp. 519–524. 16. Chen, W.Z. Theoretical Model and Novel Structures for RC-IGBT; University of Electronic Science and Technology of China: Chengdu, China, 2014; pp. 1–94. 17. Zhu, L.H. Study on Novel Reverse-Conducting IGBT; University of Electronic Science and Technology of China: Chengdu, China, 2014; pp. 1–101. 18. Werber, D.; Pfirsch, F.; Gutt, T.; Komarnitskyy,V.; Schaeffer, C.; Hunger, T.; Domes, D. 6.5 kV RCDC: For increased power density in IGBT-modules. In Proceedings of the IEEE International Symposium on Power Semiconductor Devices & IC’s, Waikoloa, HI, USA, 15–19 June 2014; pp. 35–38. 19. Werber, D.; Hunger, T.; Wissen, M.; Schuetze, T. A 1000 A 6.5 kV power module enabled by reverse-conducting trench-IGBT-technology. In Proceedings of the International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy and Energy Management (PCIM), Shanghai, China, 28–30 June 2016; pp. 351–358. 20. Huang, X.J.; Ling, C.; Chang, D.W.; You, X.J.; Zheng, T.Q. Loss characteristics of 6.5kV RC-IGBT applied to a traction converter. Energies 2017, 10, 891. [CrossRef] Energies 2017, 10, 1845 17 of 17

21. Eckel, H.; Wigger, D. Comparison of the output power of RC-IGBT and IGBT/diode inverters. In Proceedings of the Preliminary Conference Program PCIM Europe 2011, Nuremberg, Germany, 17–19 May 2011; pp. 38–43. 22. Hermann, R.; Krafft, E.U.; Marz, A. Reverse-conducting-IGBTs—A new IGBT technology setting new benchmarks in traction converters. In Proceedings of the IEEE European Conference on Power Electronics and Applications, Lille, France, 2–6 September 2013; pp. 1–8. 23. Huang, X.J.; You, X.J.; Zheng, Q.L. Traction converter with desaturated diode control for reverse conducting IGBT. Trans. China Electrotech. Soc. 2016, 31 (Suppl. 2), 58–66. 24. Zhang, Z.R.; Xu, Z.; Xue, Y.L. Valve losses evaluation based on piecewise analytical method for MMC–HVDC links. IEEE Trans. Power Deliv. 2014, 29, 1354–1362. [CrossRef] 25. Liu, Z.G. Power Electronics; Tsinghua University Press; Beijing University Press: Beijing, China, 2004; pp. 182–191. 26. Du, X.H. Modeling and Characteristic Research of High Power IGBTs Applied in Traction; Beijing Jiaotong University: Beijing, China, 2016; pp. 1–75. 27. Zhang, Y.; Liu, J. An improved model-free predictive current control of PWM rectifiers. In Proceedings of the 2017 20th International Conference on Electrical Machines and Systems (ICEMS), Sydney, Australia, 11–14 August 2017; pp. 1–5. 28. Kwak, S.; Kim, S.E.; Park, J.C. Predictive current control methods with reduced current errors and ripples for single-phase voltage source inverters. IEEE Trans. Ind. Inform. 2015, 5, 1006–1016. [CrossRef] 29. Luo, Y.; Liu, C.; Yu, F. Predictive current control of a new three-phase voltage source inverter with phase shift compensation. IET Electr. Power Appl. 2017, 5, 740–748. [CrossRef] 30. Garcia, C.F.; Rivera, M.E.; Rodríguez, J.R.; Wheeler, P.W.; Peña, R.S. Predictive current control with instantaneous reactive power minimization for a four-Leg indirect matrix converter. IEEE Trans. Ind. Electron. 2017, 2, 922–929. [CrossRef] 31. Liu, Y.C.; Ge, X.; Tang, Q.; Deng, Z.; Gou, B. Model predictive current control for four-switch three-phase rectifiers in balanced grids. Electron. Lett. 2017, 1, 44–46. [CrossRef] 32. Dütemeyer, T.; Schütze, T. Mitsubishi Datasheet: FZ750R65KE3 IGBT Module Data Sheet; Infineon Corporation: Munich, Germany, 2009.

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