PHENOMENON IN WAFER PROBING ENVIRONMENTS

EDWARD M. GODSHALK Cascade Microtech, Inc. Beaverton, OR

ABSTRACT

Investigation of and millimeter wave propagation in dielectric slabs and along copla- nar transmission lines on dielectric slabs, reveal effects that may be explained by phenomenon. These surface can be transmitted and received with wafer probes and influ- ence the transmission characteristics of coplanar transmission lines.

This paper presents measured data showing the presence of surface waves and how they interact with wafer probes and coplanar transmission lines. Methods for minimizing these in- teractions are explored and quantified. A discussion of surface wave effects on wafer calibrations is included

I INTRODUCTION

By their nature practically all wafer probing measurements involve some sort of dielectric slab (i.e. Alumina or GaAs), both in the calibration and actual measurement itself. These dielectric slabs may be viewed as open-boundary which support modes of propagation called surface waves (1). The two lowest order Transverse Electric (TE) and Transverse Magnetic (TM) modes are shown in Figure 1 for a dielectric sheet of thickness 2T. They are the TE0, TM0, TE1, and TM1 modes. The first two modes are classified as even modes due to their symmetry about the y axis, and similarly the latter two are odd modes. In all cases the strength of the mode falls off exponentially outside the dielectric (i.e. |x|>|T|) hence the term surface waves.

Note that if a metal layer is placed in the z-y plane where. x=0 the TE0 and TM1 modes will not be supported with these boundary conditions. This condition models the case of a dielectric slab of height T on a ground plane, as is common in many wafer probing situations. Mode charts for a .010" (0.254 mm) thick alumina (A12O3) slab are shown in Figure 2. Notice that the TE0 and TM0 modes have no cutoff , where as the TE1 and TM0 modes do.

Surface waves explain much of the phenomena discussed in this paper. In Section II two types of wafer probes are used to launch surface wave modes. Results are shown that indicate the pres- ence of these modes in a alumina slab. In Section III the interaction of Coplanar Waveguide

1 (CPW) with surface waves is explored, and methods for minimizing these interactions are pre- sented. Conclusions are presented in Section IV.

2 II WAFER PROBE INTERACTION WITH SURFACE WAVES

The two most common styles of wafer probe used today in microwave and millimeter wave wafer probing are the Ground-Signal (GS) and Ground-Signal-Ground (GSG) probe. Signals typ- ically originate in a network analyzer and reach the probe via either a coax or waveguide (2). The GS or GSG probe converts this signal into either a slot line or a coplanar field , re- spectively.

The probe tips of the GS and GSG probes are shown in Figure 3. The GS probe slot line field pat- tern effectively creates an electric dipole. The dipole will generate a transverse pat- tern in a dielectric slab when the probe tip is brought into contact with the slab surface. This will launch TE modes that are allowed in the dielectric. TM modes are also possible if the electric field can penetrate deep enough into the slab to couple into these modes. The GSG probe has two opposed dipoles that tend to cancel the transverse electric field of each other, but can gener- ate a net transverse if the electric fields are bent into the dielectric slab as shown. This will couple to the TM modes.

Experimental verification of the above hypothesis was achieved by first measuring the crosstalk (S21) between a GS and SG probe and then between a pair of GSG probes. An ideal pair of wafer probes would have no crosstalk, since this eliminates the possibility of one probe perturb ing the other. The GS/SG probes were 100 um pitch (separation between the two nickel fingers) and the GSG probes were 150 um pitch. Typical pitches range from 100 to 250 um for probes used above 26.5 GHz. The probes were calibrated; the GS/SG probes with SOLT (Short-Open- Load-Thru) and the GSG probes with LRM (Line-Reflect-Match).

In each case three tests were performed: (i) the two probe tips in air 0.15” (0.38 cm) above a piece of RF absorber, (ii) both tips on a .010” (.254mm) alumina dielectric slab (dielectric con- stant of 9.9) suspended in air 0.15” (0.38 cm) above RF absorber, and (iii) both tips on the alu- mina slab with the bottom surface of the slab placed on a aluminum plate. In all tests the probe tips were separated by .004” (0.10 mm). These three tests are designed to generate (i) no surface waves, (ii) the TE0 and TM0 surface wave modes, and (iii) only the TM0 mode. The results are shown in Figures 4 and 5. The data for the GS/SG pair is only valid to 40 GHz, since the coax connector on this probe overmodes above this frequency. The increased noise above 45 GHz in the GSG data is from the system noise floor of the network analyzer and should not be interpre- ted at surface wave phenomenon.

3 In test (i) the GS/SG probe. pair has a crosstalk of -42 to -39 dB from 10 to 40 GHz compared to - 53 to -49 dB for the GSG pair. Presumably the dipole cancelation in the GSG probe results in this additional 10 dB of isolation compared to the GS/SG probes. In test (ii) the GS/SG probes show an increase of about 4 dB in crosstalk due to coupling to the surface wave modes. When the ground plane is added in test (iii) the crosstalk increased only 0 to 3 dB supporting the hy- pothesis that the dipole couples strongest to the TE0 mode, which is now suppressed.

In the GSG case crosstalk increased 7 to 8 dB for tests (ii) and (iii) relative to test (i) versus only 4 dB for the GS/SG probes, although it should be noted that the absolute crosstalk level is still lower for the GSG case. The increase is nearly identical for tests (ii) and (iii) which suggests the TM0 mode is dominant for a GSG probe since only it propagates in both tests. This conclusion is supported by field overlap integral calculations which predict greater coupling to the TM0 mode relative to the TE0 mode for coplanar waveguide on a dielectric substrate (3). The GSG probes seem to couple tighter to the TM0 mode than the GS/SG probes do to the TE0 mode, based on the relative increases in crosstalk noted

For completeness, the suspended dielectric slab measurements should also be done on .020” thick alumina slabs, since a .020” suspended dielectric slab has the same vs. frequency re- sponse for the TM0 and TE1 modes as a .010" grounded slab. These measurements were per- formed and resulted in similar data to the .010" suspended slab experiment with the exception of 38-46 GHz for the GS/SG case. In this range, there was a pronounced null in S21 reaching -50 dB at approximately 42 GHz. A plausible explanation is that the .020” alumina slab is thick at 45 GHz, where is the in alumina. Around this frequency, the open boundary condition at the slab bottom may be transformed into a virtual short circuit at the slab top sur- face. This would effectively place a short circuit at the probe tips, explaining why resulting cross- talk is even lower than when no surface waves are present in test (i).

4 This data has application to the question of whether it is better to have the probe tips in air or on a substrate when it is desired to terminate the probe in a zero length open reference termination, such as during calibration. It has been found that the coefficient is high enough to use as an open in either case. An ideal open would have no perturbation from external objects, in- cluding the other probe. Thus, performing the open calibration with the probe tips in air will be more ideal, since this will minimize crosstalk and result in better isolation.

III COPLANAR WAVEGUIDE INTERACTION WITH SURFACE WAVES

A cross section of a Coplanar Waveguide (CPW) on a dielectric slab is shown in Figure 6. It has a ground plane of width A, gaps of width G, and a signal line of width W. and radia- tion loss are minimized for the CPW line when the ground-to-ground spacing (W+2G) is small compared to the dielectric wavelength and substrate thickness (3). Dispersion occurs when the CPW mode couples to the surface waves via radiated energy (4,5,6). The (W+2G) separation be- tween the ground planes may be thought of as an aperture, hence reducing this quantity reduces the radiated energy available for coupling to surface waves.

Reducing the substrate height, T, increases the cutoff frequency of higher order surface wave modes. By picking T small enough these modes are pushed above the frequency range of inter- est. The TE0 and TM0 have no cutoff frequency, hence they must be dealt with in an other man- ner. The TE0 mode can be eliminated by adding a ground plane to the substrate bottom, but this may result in microstrip modes (4). Some insight as to how the CPW interacts with surface waves may be gained by studying Figure 2. If the CPW ground plane widths, A, were infinite, and the gaps had no effect, the mode chart in Figure 2 (b) would represent this situation. No TE0 mode would propagate due to the ground plane. Notice as the frequency increases eventually the propagation phase constant of the CPW mode becomes equal to that of the TM0 mode. At this “critical” frequency, coupling may occur between the two modes possibly resulting in attenua- tion of the CPW modes. For this example T=.010" resulting in a critical frequency of 120 GHz.

Of course in the real world, ground planes are finite widths and the coupling is more complex (5.6). At very low CPW does not interact with other modes, hence only the dominant mode exists. It has been proposed that a “surface-wave-like mode” may exist at frequencies below the critical TE0 and TM0 frequencies, although the coupling is expected to be minor since the fields of this mode and the CPW mode are not very similar (5). As the frequency increases further a TM0 mode will begin to propagate beneath the CPW ground planes, since the boundary conditions are similar to those shown in Figure 2 (b). The critical frequency will not be exactly as shown due to the finite ground plane width.

Beyond the edges of the ground planes there is no metalization, so the boundary conditions and modes shown in Figure 2 (a) exist. The CPW mode can couple to these modes at the critical fre- quencies shown: 122 GHz for the TE0 mode, and 240 GHz for the TM0 mode. Typically this coupling is referred to as leakage, and can cause noticeable attenuation of CPW mode if the cou- pling is strong enough. As mentioned in Section II overlap integral theory predicts tight coupling between the CPW and TM0 modes.

5 To experimentally study the effect of coupling between the CPW mode and surface wave modes, a CPW line was made with a large aperture (W+2G = .035” (0.88mm) to promote radiation. The substrate height, T, was then increased from .010” (0.25mm) to .080” (2.03mm) resulting in the data shown in Figure 7. Increasing the substrate height lowers the critical frequencies where the CPW mode intersects the various modes shown in Figure 2. Table 1 lists the predicted critical frequencies where coupling is allowed.

Table 1. Predicted critical frequencies where the CPW mode has the same propagation phase con- stant as the various modes listed.

6 Strong coupling between the CPW mode and a surface wave mode is observed in Figure 7 as a null. At T=.040” a null begins to appear and decreases in frequency as the substrate height is in- creased to .080”. Comparing this data with Table 1 suggests that this null is a result of energy leakage from the CPW mode to the TM0 mode in the suspended dielectric slab existing beyond the ground plane edges. The predicted values are about 3-4 GHz lower than observed. A second null begins to appear at 50 GHz for T=.080”. If the TE1 mode line is extrapolated for the sus- pended dielectric case it intersects the CPW mode at about 50 GHz which leads us to assign that mode to this null.

The TE0 mode for the suspended substrate and TM0 mode for the grounded substrate (the region under the CPW ground planes) were not observed in the experimental data. Briefly, this implies that the coupling is weak between these modes and the CPW mode, but this should be investi- gated further.

To minimize coupling between the CPW mode and surface waves, for a given substrate height T, it seems logical that reducing the aperture (W+2G) should help. To determine what value of (W+2G) /T is acceptable for CPW lines three different cases were studied, and are listed in Table 1. For each case three different boundary conditions were presented at the substrate bottom: (i) the substrate was suspended in air over an RF absorber, (ii) the substrate was placed on a .050” (1.27mm) thick glass slide suspended in air over RF absorber, and (iii) the substrate was placed on an aluminum plate. The measurements were taken with a pair of 150 mm pitch GSG wafer probes.

Table 2. Characteristics of three CPW lines evaluated All dimensions in um.

These different boundary conditions were designed to alter the surface wave modes significantly and hence cause perturbation the CPW mode if mode coupling was present This was studied by measuring the insertion loss of the CPW line for each boundary condition. Note that the lines are all close to the same length, which is important since the degree of interaction is length depen- dent. The results are shown for cases 1 and 3 in Figures 8 and 9. The results for case 2 were sim- ilar to case 3, where the worst perturbation was less than 0.1 dB, versus 0.3 dB for case 1.

7 The data indicates that for calibration techniques which use CPW lines on this order of length or longer, such as Thru-Reflect-Line (TRL); the quantity (W+2G)/T should be. kept less than about 0.3 if interaction with surface waves is to be minimized. For thin millimeter wave GaAs sub- strates, (typically only 50 to 100 um thick to allow good heat transfer and prevent higher order modes) using (W+2G)/T = 0.3 and T=l00mm requires that W+2G = 30 um. This is too small to probe with currently available probes. If G is increased to 50 um, a realizable probe pitch, the becomes essentially microstip in nature since the ground plane now dominates the electric field around the signal line. This implies that for CPW TRL standards some sort of taper will be needed to transition from the very small gap and signal line dimensions of the trans- mission line to pads large enough for wafer probe tips.

Other options for realizing on wafer calibration standards on thin GaAs substrates are microstrip TRL standards or lumped element standards of small enough size to minimize unwanted interac- tion with the ground plane. If lumped element standards are used the parasitics of the individual elements must be accurately characterized

IV CONCLUSIONS

Surface wave modes have been shown to have a noticeable effect on wafer probe measurements. The crosstalk increases between two probes when their tips are placed on a dielectric slab due to transmission of energy via these modes.

Coplanar Waveguide transmission lines can be perturbed by interaction with surface wave modes if the ground-to-ground spacing is to large relative to the dielectric waveguide and sub-

8 strate height. This may prove to be a concern when using long CPW lines (i.e. greater than 1000 um) on thin substrates (i.e. 50 and 100 um GaAs) unless (W+2G)/T is not allowed to exceed 0.3.

It appears that for GaAs MMICs with thin substrates on wafer calibration standards could be TRL microstrip lines, CPW lines with tapers, or lumped elements of small enough dimensions to minimize interaction with surface wave modes. The microsnip approach will most likely imply a standardization of via holes if a standard calibration layout is to be developed. Similarly the CPW approach would imply that an identical taper was used for both the standards and device measurement ports.

Another solution is to use off-wafer calibration standards. Examples are commercially available Impedance Standard Substrates (ISS) and specially fabricated GaAs wafers.

The ISSs tend to be on either sapphire or alumina and are typically populated with lumped ele-, ment standards whose parasitics are well characterized and are dimensionally small enough to minimize interaction with surface waves. These lumped elements are commonly used for Short- Open-Load-Thru (SOLT), Line-Reflect-Match (LRM), and Line-Reflect-Reflect-Match (LRRM) calibrations. The later two have been developed to a high degree (7). Transmission line standards are usually included for TRL calibrations. It has been found that CPW lines on alumina have slightly lower insertion loss than identical lines on sapphire. Alumina also has no axial depen- dance, since it is homogeneous versus sapphire which is crystalline.

A GaAs wafer Impedance Standard. has been fabricated to support a sophisticated TRL method, used for calibration verification, developed by the National Institute of Standards and Technol- ogy (NIST) (8). This wafer is relatively immune to surface wave interaction by having (W+2G)/T = 0.34

These two off-wafer calibration solutions have good agreement when well understood calibra- tion methods are used. When the alumina ISS was used for a LRRM calibration, in conjunction with automatic determination of load inductance, and then compared with the NIST TRL method, less than 2.2% disagreement was found for the characteristic impedance to 50 GHz. This is an important observation, since the ISS used for the LRRM calibration was on alumina and the TRL standards were on GaAs. The implication is that high accuracy calibrations are transferable from an alumina ISS to a GaAs wafer without serious degradation.

REFERENCES

1. R. E. Collin, Field Theory of Guided Waves, New York: McGraw-Hill, 1960.

2. E. M. Godshalk, “A V-Band Wafer Probe Using Ridge-Trough Waveguide,” IEEE Trans. Mi- crowave Theory Tech., vol. MTT-39, pp. 2218-2228, Dec. 1991.

9 3. M. Riaziat, R. Majidi-Ahy, and I. J. Feng, “Propagation Modes and Dispersion Characteristics of Coplanar Waveguides,” IEEE Trans. Microwave Theory Tech., vol. MTT-38, pp. 245-251, March 1990.

4. R.W. Jackson, “Mode Conversion Due to Discontinuities in Modified Grounded Coplanar Waveguide,” 1988 IEEE MIT-S International Symposium, pp. 203-206, May 1988.

5. H. Shigesawa, M. Tsuji, and A. Oliner, “A New Mode-Coupling Effect on Coplanar Waveguides of Finite Width,” 1990 IEEE MIT-S International Symposium, pp. 1063-1066, May 1990.

6. M. Tsuji, H. Shigesawa, and A. Oliner, “New Interesting Leakage Behavior on Coplanar Waveguides of Finite and Infinite Thickness,” 1991 IEEE MTT-S International Microwave Sym- posium, pp. 563-566, June 1991.

7. A. Davidson, K. Jones, and E. Strid, “LRM and LRRM with Automatic Determination of Load Inductance,” 36th ARFTG Conf. Dig. (Monterey, CA), pp. 57-62, Nov. 1990.

8. D. Williams, R. Marks, and A. Davidson, "Comparison of On-Wafer Calibrations," 38th ARFTG Conf. Dig. (San Diego, CA), pp. 68-81.

ACKNOWLEDGMENTS

The author wishes to thank Jeff Williams and the Thin Film Lab staff for all their assistance in fabricating the coplanar substrates used in this research. Valuable suggestions were provided by Eric Strid and Dan d’Almeida. Dave Walker and Dylan Williams of NIST supplied the GaAs CPW structures, and Dr. Riaziat of Varian was helpful.

Copyright © 1992 Cascade Microtech, Inc. AR130

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