Overvoltage and Surge Protection in Variable Frequency Drives

A Dissertation Presented by Dawood Talebi Khanmiri

to The Department of Electrical and Computer Engineering

in partial fulfillment of the requirements for the degree of

Doctor of Philosophy in the field of Electrical Engineering

Northeastern University Boston, Massachusetts

October 2020

Acknowledgment

I would like to thank my advisor, Professor Brad Lehman, for his continuous support of my PhD research, for his patience and guidance, without whom this thesis could not be completed.

Besides my professor, I would like to thank Mersen who partially funded this research and generously provided the lab for high power experiments, especially Roy Ball, Jerry Mosesian, and late Craig

McKenzie, who helped me during my research.

I would, also, like to thank my thesis committee members, for their time and their insightful comments in my proposal review. I would like to thank all of my labmates and fellow grad students, too, for all the good memories that we have had together.

Last, but not least, I would like to thank my family who supported me in my studies, especially my wife, Dr. Nasibeh Nasiri, without whom I would not have been able to complete my PhD.

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Abstract

Overvoltage and surge protection of electric devices is an indispensable part of the system protection.

Metal Oxide Varistors (MOVs) are the most common components used for surge protection. However,

MOVs are known to degrade over time when they experience severe discharges. This degradation may, particularly, lessen protection capabilities for fast switching PWM applications, such as inverters and drives. For instance, Variable Frequency Drives (VFDs) are becoming more and more widespread for motor speed control and energy saving purposes. However, when connecting long cables between the inverter and the motor, high frequency overvoltages appear on motor terminals due to the voltage reflection phenomenon. These overvoltages can damage the motor and the cable insulation, and at the same time, cause installed MOVs to conduct high currents and fail. In this research, we investigate the MOV’s behavior and degradation in such applications. The VFD systems and characteristics of the overvoltages are first modeled. MOVs behavior is investigated through experimental tests and simulations. Finally, new methods of applying MOVs in PWM motor drives are proposed to mitigate the overvoltages.

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Table of Contents

1. Introduction ...... 1 1.1. Motivation ...... 1 1.2. Background ...... 3 1.2.1. Surges and overvoltages ...... 3 1.2.2. Standard Surge waveforms ...... 7 1.2.3. Metal Oxide Varistors ...... 9 1.2.4. MOV models ...... 12 1.2.5. MOVs’ failure modes ...... 12 1.2.5.1. Thermal runaway ...... 13 1.2.5.2. Puncture ...... 14 1.2.5.3. Cracking ...... 17 1.2.5.4. Flashover ...... 17 1.2.6. Effective Factors in MOVs’ Failure Modes ...... 18 1.3. Problem Statement...... 23 1.4. Dissertation Organization ...... 25 2. Degradation and Health Monitoring of Metal Oxide Varistors ...... 27 2.1. Energy Absorption Capability ...... 27 2.2. Degradation of MOVs ...... 33 2.3. Degraded MOV’s V-I curve ...... 39 2.4. Health Monitoring of MOVs ...... 40 2.4.1. Effects of temperature and voltage on health monitoring and lifetime estimation ...... 44 2.4.2. Proposed Health Monitoring Algorithm ...... 46 2.5. Summary and Conclusions ...... 48 3. Overvoltages in VFDs ...... 49 3.1. Introduction ...... 49 3.2. Effects of High Frequency Overvoltages...... 53 3.2.1. Effects on Motor ...... 53 3.2.2. Effects on Drive ...... 55 3.2.3. Other Effects ...... 57 3.3. Causes and Effective Factors ...... 58 3.3.1. Voltage Rise Time ...... 60 3.3.2. Cable Length ...... 62 3.4. Voltage waveforms in a PWM VFD system ...... 63 3.5. Simulation of a VFD system for overvoltages ...... 70 3.5.1. Cable model ...... 70

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3.5.2. Measurement of cable parameters ...... 72 3.5.3. Motor Models ...... 75 3.5.4. Measurement of motor parameters ...... 77 3.5.5. Experimental setup and Measurements...... 79 3.5.5.1. Cable parameters ...... 81 3.5.5.2. Motor parameters ...... 84 3.5.6. Simulation results ...... 86 3.6. Summary and conclusion...... 92 4. Overvoltage Mitigation and Protection in VFDs ...... 93 4.1. Surge Protection in VFDs ...... 93 4.2. Overvoltage Mitigation in VFDs ...... 97 4.2.1. Passive Filters ...... 98 a. Filters on Motor Terminals ...... 99 b. Filters on Inverter Output ...... 101 c. Filters on Both Motor Terminals and Inverter Output ...... 104 4.2.2. Active Filters ...... 105 4.2.3. PWM Control ...... 108 4.2.4. DC Cable between Rectifier and Inverter ...... 112 4.2.5. Energy Varistor ...... 113 5. MOVs in Variable Frequency Drive Systems ...... 116 5.1. MOV installation on VFD systems...... 117 5.2. Behavior of MOV under Repetitive Pulses ...... 117 5.2.1. MOV’s Response to slow Pulses from a power supply ...... 118 5.3. Behavior of MOV under PWM Pulses ...... 130 5.4. High Frequency Model of MOV ...... 135 5.4.1. Simulation of MOV in VFD system ...... 139 5.5. MOV Application Solutions for VFD systems ...... 143 5.5.1. MOV in series with GDT...... 143 5.5.1.1. Gas Discharge Tube (GDT) ...... 144 5.5.1.2. Simulation Results ...... 148 5.5.2. TSPD-Switched MOVs (TSMOV) ...... 150 5.5.2.1. Thyristor Surge Protective Device (TSPD) ...... 152 5.5.2.2. Simulation Results ...... 153 5.5.3. Reconfigurable Surge Protective Device (RSPD)...... 156 5.5.3.1. Simulation Results with Ideal Switches ...... 158 5.5.3.2. Switches in RSPD ...... 160 5.6. Summary and Conclusions ...... 164 6. Conclusions and Future Work ...... 165 6.1. Summary of Results ...... 166

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6.2. Suggestions for Future research ...... 168 Bibliography ...... 173

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Chapter 1: Introduction 1

Chapter 1:

1. Introduction

This chapter provides motivation, definitions and background for this research. A literature review on failure modes of MOVs is presented. Objectives and outcomes of the research are presented.

1.1. Motivation

The continued trend of miniaturization and use of lower voltages in electronics have made electronic circuits and devices more susceptible to overvoltage transients and surges [1], [2]. Smaller components with lower energy capacities, closer traces and dense circuits have triggered attention to overvoltage transients that might not have been seen in the past. Additionally, proliferation of electronic devices in everyday life and mission critical applications, such as industrial automation, communication, monitoring and control systems, has made any failure costly due to the repair and service interruption costs. Critical systems, such as personal computers and communication devices, are being damaged by surges [3]. This can inflict large financial losses due to downtime, loss of services, and customer dissatisfaction. For example, a Congressional Research Service study in 2012 estimates the cost of weather-related power outages at $20 to $55 billion annually, for American businesses and industries [4]. Furthermore, with the expansion of smart grids and wide collection of data and system information, electronic devices need to be installed on the power distribution lines and communication towers, where they are exposed to more frequent and more severe transients and surges [5]. When the increased susceptibility of the modern

Chapter 1: Introduction 2

computerized systems and their high cost of repair are combined with the fact that the exposure of the devices to surges has increased, a dire need of a reliable surge protection becomes obvious.

Overvoltage transients and surges are among major failure causes of power electronic converters and power supplies [6]. Overvoltage spikes damage the electronic devices and electrical appliances in two primary ways: 1) they may absorb more than the Energy Absorption Capability (EAC) or voltage tolerance limit of the components. This normally damages the electronic devices instantly and permanently. 2) they cause extra power loss and heating in the device and degrade it over time. In order to protect against surges and transient overvoltages Surge Protection Devices (SPDs) are used in electrical appliances and power systems.

Metal Oxide Varistors (MOVs) are protective devices that are widely used in SPDs and surge arresters.

MOVs are low-cost sintered ceramic blocks with a highly non-linear V-I characteristics. They show high resistance in normal system voltages and low resistance when an overvoltage incident occurs. An MOV is placed in parallel with the load and diverts the surge energy away by providing a low impedance path to the ground for the surge current and maintains the voltage within acceptable limits.

Generally, MOVs are expected to protect the load against infrequent surges and short transient overvoltages. However, in the power electronic systems, overvoltage transients might be generated within the system due to the interaction of converters and loads. For example, fast switching of IGBT switches in

Variable Frequency Drive (VFD) systems causes high frequency transient overvoltages that appear on the terminals of the motor [7], [8]. These repeated stresses can lead to the premature failure of the dielectric of the motor. Furthermore, these consistent repeating overvoltages pose new challenges to the application of the MOVs for the surge protection purpose. Frequent high current discharge generates extra heat in the

MOVs that may lead to their thermal runaway and failure. Understanding the surges, transient environment and the MOVs’ behavior is essential in surge protection of electronic converters.

Chapter 1: Introduction 3

1.2. Background

1.2.1. Surges and overvoltages

Surges are overvoltage spikes, usually having the lengths of several tens of microseconds. Based on

IEEE definition [9], they do not exceed one-half period of the fundamental frequency in duration.

Although, generally, SPDs are not expected to protect against temporary overvoltages (TOVs), consideration of TOVs is necessary to ensure the compatibility of SPDs to TOV-related stresses. A simplified curve of transient environment is shown in Figure 1.1 [9]. It should be noted that the boundary between No Effect, Upset, and Damage regions are not exact and vary with the withstand characteristics of the equipment exposed to the surges. The sensitivity of equipment is a key component in evaluating the effect of voltage sags and interruptions at a facility. However, it is not common to easily obtain the sensitivity data for components in their datasheets. A curve, developed by CBEMA (Computer and

Business Equipment Manufacturers Association) and revised by ITIC (The Information Technology

Industry Council) [10], describes an AC input voltage envelope which typically can be tolerated (with no interruption in function) by most information technology equipment. Figure 1.2 shows the ITIC curve.

Figure 1-1. Voltage and duration of transients and their effects on equipment [9]

Chapter 1: Introduction 4

Figure 1-2. ITIC (CBEMA) curve: Ride through capability of IT Equipment [10]

Mainly, surges are generated by two major sources: lightning and switching. Lightning surges originate from a direct flash to the power system, or an inductive coupling of a nearby flash into the power line.

Lightning flash density varies from 0 to 14 flashes/sq. km/year in the United States, while in tropical areas this number can reach 70 flashes/sq. km/year [11]. A cloud-to-ground (CG) flash is typically composed of a sequence of individual cloud-to-ground return strokes which transfer significant charge from the cloud to ground, each stroke exhibiting peak currents in the range of 5 kA to 300 kA. These strokes have a nominal duration of 20-50 microseconds, and are typically separated in time by 20 to 100 msec. A flash will typically be comprised of 2-3 strokes, but may contain as few as one and as many as twenty strokes [11].

Flash density map of the United States and stroke density map of the world are given in the Figures 1.3 and

1.4. A lightning flash can discharge a huge current, with a typical value of 20 kA, into network, which causes a severe transient overvoltage. Inductive coupling of lightning can cause severe overvoltages, too. A one kilometer distant lightning strike can induce about 200 volts in 1 meter of wire [12].

Switching surges are related to the release of the trapped energy in the inductance or capacitance of the system. They are caused by any switching operations in the power system such as: Load switching, renewable source switching, or fault clearing. The sudden change in the system can initiate damped oscillations with high frequencies that depend on resonant frequencies of the network. Magnitude and duration of the switching overvoltages depends on many parameters such as type of the load and the switching device [9].

Chapter 1: Introduction 5

Figure 1-3. US flash density map from National Lightning Detection Network [11] (Reprinted with permission)

Figure 1-4. World stroke density map from National Lightning Detection Network [11] (Reprinted with permission)

Overvoltages generated by the switching in power electronic converters are similar to the switching overvoltages in nature, with the difference that they occur continuously and with high frequencies. Figure

1.5 shows a simple circuit where a simple series RLC circuit is being switched on by an ideal switch at t =

0.005 s. The frequency of the switching transient is determined by values of inductance, capacitance, and resistance. The peak voltage of the switching transient is dependent on the values of circuit elements and on the voltage of the power system at the time of occurrence.

Chapter 1: Introduction 6

Switch turns on

Figure 1-5. Switching transient in a series RLC circuit with ac source

Switch turns on

Figure 1-6. Switching transient in a series RLC circuit with dc source

In most cases, the maximum overvoltage is in the order of twice the peak amplitude of the system voltage, but higher values can occur, especially when switching on or off inductive loads (motors, transformers) or capacitive loads. Also, interruption of short-circuit currents can cause high overvoltages

[9]. Figure 1.6 shows the transient response of the same circuit with a dc source. As it is seen from the figure, the frequency of the oscillations does not change as it is dependent on the values of resistance, inductance and capacitance of the circuit.

In addition to switching, fault clearing can cause high overvoltages. During a short circuit, a large current flows from faulty line to ground or to the other lines. Sudden interruption of this large current creates an overvoltage on lines inductance. Figure 1.7 shows an example of an overvoltage created due to clearing a fault.

Chapter 1: Introduction 7

Switch turns on

Figure 1-7. Overvoltage created due to fault clearing

TOVs are defined as power frequency overvoltages that occur in the power system and should be distinguished from switching overvoltages. For surges, the maximum duration is one half-cycle of the applicable power frequency [9]. TOVs, on the other hand, occur at the power system frequency. Swells, overvoltages longer in duration than a surge, but lasting only a few seconds, are considered to be a subset of TOVs. Typically, existing protective relays and circuit breakers clear the fault within a short time. The main parameters that influence the amplitude and the duration of TOVs are the type and configuration of the grounding system and the method of fault clearing in the medium voltage network [9]. The equipment is designed to tolerate the TOVs in the power system. Generally, SPDs do not have enough energy handling capability to limit TOVs, but they are exposed to these overvoltages and are expected to ride through these events with no failure and minimal degradation.

1.2.2. Standard Surge waveforms

IEEE C62.41.2-2002 defines two standard surge-testing waveforms: The Combination Wave, and The

100 kHz Ring Wave [13]. The combination wave is defined by two waveforms: a 1.2/50 µs open-circuit voltage waveform and an 8/20 µs short-circuit current waveform. The first and second numbers show the front time and the duration of the waveform, respectively. Figure 1.8 shows the combination wave and its parameters. The peak value of voltage and current is selected from the standard based on the location category and specification of the device under test.

Chapter 1: Introduction 8

(a) (b)

Figure 1-8. (a) Combination Wave 1.2/50 µs open-circuit voltage, (b) Combination Wave 8/20 µs short-circuit current [13]

IEEE standard defines the front time for voltage waveforms as 1.67 × (t90 - t30), where t90 and t30 are the times of the 90% and 30% amplitudes on the leading edge of the waveform. The duration is defined as the time between virtual origin and the 50% amplitude point on the tail. The virtual origin is the point where a straight line between the 30% and 90% points on the leading edge of the waveform intersects the V = 0 line. For the current waveform, the front time is defined as 1.25 × (t90 - t10), where t90 and t10 are the times of the 90% and 10% points on the leading edge of the waveform. Similar to the above, the virtual origin is the time that a straight line between the 10% and 90% amplitude points on the leading edge of the waveform intersects the I = 0 line. For the Combination Wave, the effective source impedance, the ratio of

Vpeak / Ipeak, is 2.0 Ω.

The Ring Wave is defined as a voltage waveform and no current waveform is specified. It is shown in

Figure 1-9. 100 kHz Ring Wave [13]. The amplitude will decay so that the ratio of adjacent peaks of opposite polarity is as follows:

- The ratio of the second peak to the first peak is between 40% and 90%.

- The ratio of the third peak to the second peak and the ratio of the fourth peak to the third peak are

between 40% and 80%.

- There is no requirement set on the amplitude of the Ring Wave beyond the fourth peak.

Chapter 1: Introduction 9

Figure 1-9. 100 kHz Ring Wave [13]

1.2.3. Metal Oxide Varistors

Metal Oxide Varistors (MOVs) are highly nonlinear components that are used to protect the sensitive circuits against surges and transient overvoltages. An MOV is placed in parallel with the load and diverts the surge energy away by providing a low-impedance path to the ground for the surge current and maintains the voltage within acceptable limits. They show very high resistance when voltage is low, around working voltages, and very low resistance when over-voltages occur. Figure 1.10 shows the basic principle of MOV application and its V-I characteristic.

Load MOV

Figure 1-10. MOV Application and Voltage-Current Characteristics

This curve can be divided into three regions. In voltages below the Maximum Continuous Operating

Voltage (MCOV) a small leakage current flows through the MOV and the resistance is high. This region

Chapter 1: Introduction 10

which sometimes is referred to as linear region shows the operating point of an MOV in normal system voltage and is called leakage current region. When the system voltage increases above MCOV, the MOV conducts a current larger than its leakage current and the voltage remains almost constant. This region is referred to as discharge region or normal operation region. In currents above the nominal discharge current, the voltage is enough to break all barriers and the only current limiting factor is the grains’ resistance. This region in which the voltage almost linearly increases with current is called up-turn region.

The voltage clamping characteristic of MOVs results from the complex microstructure of varistor ceramics. They are polycrystalline materials containing metal oxide grains of random size and shape. Zinc

Oxide is the most common material used in varistors. It is known that back-to-back Schottky barriers form at the grain boundaries. Switching occurs when the applied voltage exceeds the breakdown voltage, with the result that the conductivity increases by several orders of magnitude. The non-linear characteristics are quite dependent on the varistor formulation and the details of the fabrication process.

Figure 1.11 shows an MOV’s simplified microstructure. ZnO grains are highly conductive. At grain boundaries where ZnO grains are separated by a thin amorphous Bi rich film, back to back Schottky barriers form. These microvaristors are comparable to the symmetrical back-to-back Zener diodes with an average breakdown voltage of about 3.5 V. Their series and parallel connections determine the non-linear behavior of the varistor block.

Figure 1-11. Microstructure of ZnO varistor

Definitions of important specifications of an MOV in this work are as follow.

Maximum Continuous Operating Voltage (MCOV): The maximum designated root mean square (rms) value of the power frequency voltage that may be continuously applied to an SPD.

Chapter 1: Introduction 11

Nominal System Voltage: A nominal value assigned to designate a system of a given voltage class.

Clamping Voltage (Also called Let-through Voltage or Limiting Voltage): The maximum magnitude of voltage, measured at the leads, terminals, or receptacle contacts after the application of an impulse of specified wave shape and amplitude.

Nominal Varistor Voltage (V1mA): The voltage across the varistor at 1mA, DC.

Voltage Protection Rating (VPR) – A rating selected from a list of preferred values as given in the standard and assigned to each mode of protection. The value of VPR is determined by the average measured limiting voltage when applying combination wave of 3kA.

Leakage Current1: The current flowing through the MOV when applying the rated voltage.

Nominal Discharge Current (In): Peak value of the current, selected by the manufacturer, through the

SPD having a current wave shape of 8/20 µs where the SPD remains functional after 15 surges.

Maximum Surge Current: Peak value of the maximum non-repetitive surge current, given by the manufacturer.

Energy Absorption Capability (EAC)2: The energy that MOV can absorb in one incident of impulse or

TOV, before it fails.

Non-linearity coefficient (α): defined as the slope of the V-I characteristic in normal operation region and often is calculated as below:

푳풐품 푰 −푳풐품 푰 휶 = ퟐ ퟏ Equation 1-1 푳풐품 푽ퟐ−푳풐품 푽ퟏ

Usually for MOVs with leakage currents in microamperes, the points 1 and 2 are corresponding to currents

I1=0.1 mA and I2=1 mA on V-I curve.

Time to Failure (TtF)3: time between the application of the overvoltage or surge, until MOV’s destruction.

1 There is no agreed-upon definition of Leakage Current for low voltage MOVs. 2 There is no standard method to measure EAC. EAC in this work is measured by applying a given voltage until MOV’s destruction. 3 Not to be mistaken with Mean Time Between Failures (MBTF)

Chapter 1: Introduction 12

1.2.4. MOV models

A simple equivalent circuit for varistor modeling is shown in Figure 1-12. The main element of the model is a non-linear resistance, shown as V=f(I). Series resistance and inductance is mainly due to varistor leads and terminals. Typically, Rs is negligible and Ls is considered only when a surge with a fast rise time is considered [14]. Values of 푅푠 = 100 휇훺 and 퐿푠 = 10 푛퐻 can be assumed for these parameters [15].

Capacitance, CP, can be measured in different frequencies depending on the application.

(a) (b)

Figure 1-12. Varistor Model (a) Simple model (b) IEEE frequency dependent model

The simplest model to represent the non-linear V-I characteristic of an MOV is expressed as 퐼 = 푘. 푉훼, where α is the non-linearity coefficient. Based on this assumption, a voltage dependent current source or a current dependent voltage source can be used to model a varistor. More accurate expressions are also proposed [16]:

-log (I) log (I) log V = b1 + b2 log (I) + b3 e + b4 e Equation 1-2

1.2.5. MOVs’ failure modes

Although MOVs generally are durable devices, certain mechanical and electrical conditions can lead to their failure. Most of the times this failure involves high temperatures, and this can lead to fire or damage to the nearby equipment if no thermal or over current protection is devised. MOVs fail in four different ways: thermal runaway, puncture, cracking, and flashover [17]–[21].

Chapter 1: Introduction 13

1.2.5.1. Thermal runaway

Thermal runaway occurs when the generated heat by an increased leakage current rises beyond the heat dissipation capability of the MOV. This excess of heat would increase the MOV temperature and consequently, its leakage current. Increased leakage current would generate more heat and would rise the temperature again. This positive feedback could lead to rapid increase in the MOV’s temperature and a thermal runaway.

The generated heat, PLoss, due to the Watt loss in the varistor is related to its temperature as

퐰 퐏 ∝ 퐞퐱 퐩 (− 퐜) Equation 1-3 퐋퐨퐬퐬 퐤퐓

where wc is the activation energy for conduction, k is the Boltzmann’s constant, and T is the varistor’s average surface temperature. The dissipated power into the environment is almost a linear function of varistor’s temperature and can be expressed as

퐏퐝퐢퐬 = 훂 퐒 (퐓 − 퐓퐚) Equation 1-4

where α is the convective exchange coefficient, S is the total heat dissipation surface, and Ta is the ambient temperature [22]. As shown in Fig. 1.13, PLoss and Pdis may have one or two intersection points. In normal operating conditions the heat generated in the varistor is equal to the heat dissipated to the environment, so a balance is kept in point P1. When a varistor absorbs the surge energy, its temperature rises suddenly and increases power loss generated by the increased leakage current. At such a condition if the varistor temperature exceeds the stability limit P2 and enters the overheating region, its temperature cannot be stabilized by heat dissipation and a thermal runaway occurs [21].

Figure 1-13. Thermal stability diagram of an MOV

Chapter 1: Introduction 14

1.2.5.2. Puncture

Puncture and cracking occur because of imperfect microstructure of the MOV. Non-uniformity of grain sizes and breakdown voltages of potential barriers at grain boundaries create heterogeneous current densities within the MOV. This current concentrates into a few narrow paths and heats up those areas more than adjacent areas and increases the conductivity of those hot spots and paths. This leads to even a higher current density and increased locally generated heat. The generated heat creates a temperature gradient that causes thermal stresses between hot spots and neighbor portions, causing MOV block to crack, or increases the temperature of the hot spots up to the melting point of the ceramic, leading to a puncture failure.

Usually, puncture occurs in low density currents, where there is enough time for the hot spots to form. On the other hand, cracking failure generally occurs when the current density is high, and stresses become more than stability strength of the ceramic and crack the MOV before reaching to the melting point. Figure

1.14 shows some MOVs that failed in puncture mode.

Figure 1-14. Puncture failures in MOVs

Varistors are sintered from metal oxide powders which are mixed as well as possible to make the varistor disks uniform in thermal and electrical characteristics. But always imperfection exists in varistor’s sintering process and grain growth which makes the threshold voltage to vary in different sections of the disk’s surface. A measurement by small spot electrodes (3 mm in diameter) in [16] shows this threshold voltage can vary up to more than 10%. Figure 1-15 shows threshold voltage distribution in 2 different varistors with 114 mm diameter and 10 mm thickness.

Chapter 1: Introduction 15

Figure 1-15. Non-uniform threshold voltage distribution in ZnO varistors [23]

This difference comes from the non-uniformity in grain size and threshold voltage of the grain boundaries. The size of grains depends on the formulation of materials and the sintering process. For the varistors mentioned above average grain size was 18 μm, and the grain sizes were distributed in the range of 5-50 μm. The behavior of individual grain boundaries is dependent on varistor’s microstructure. Grain interfaces containing a thin amorphous Bi-rich film exhibit symmetrical current-voltage characteristic with a breakdown of 3.6 V in average, where higher threshold voltages are observed, too. Junctions between

ZnO and intergranular Bi2O3 exhibit asymmetrical current-voltage characteristics. Electrons traveling from a ZnO grain into the intergranular region experience a breakdown voltage at 3.2 V, while breakdown voltages in the opposite direction are 0.4 and 0.9 volts. Interfaces between ZnO and pyrochlore do not exhibit distinct varistor properties. [24]

Variation in varistor voltage on the surface of a varistor disk causes the current density to be different in various portions. Current density is higher in portions with lower varistor voltage and the temperature rises faster in these portions. Since the leakage current increases with the temperature, this heating increases the current again leading to “current concentration.” If the current magnitude and duration is enough this positive feedback process will take place and will concentrate the current in a narrow path through the

MOV. The temperature of that current path will reach the melting point of the ceramic (820 ˚C for Bi2O3,

1700 ˚C for ZnO) creating a puncture through the MOV and resulting in its failure. The “current concentration” is a thermal process that takes time. Therefore, it is seen in long duration surges and does not occur during short duration impulses such as 8/20 µs current surge. Moreover, the longer the impulse is, the more obvious current concentration will be. The thermal pictures of Figure 1.16 give an evidence of such facts.

Chapter 1: Introduction 16

[25]

Figure 1-16. Current concentration in an MOV, gray areas show the highest temperature areas (Colors changed, originally white); (a) 8/20 µs, 40.8 kA/1700 V, 83-88˚C (b) 10/350 µs, 2.76 kA/920 V, 82-87˚C, (c) 50 Hz, 3 s, 2.04 A/740 V, 110-126 ˚C

Understanding current localization phenomenon helps to interpret some of the MOVs characteristics, and results of this research on the topic that are presented in Chapter 2. Current concentration increases with increasing voltage, as the breakdown of some barriers in small areas enables the current to be drawn into a few paths. But eventually as the voltage is further increased, more and more barriers are overcome, thus offering more current carrying paths through the microstructure. This delocalization continues until all the bonds reach the grain conductance, and the system again becomes uniform. [20]

This phenomenon can be better explained by simulation results in [18] shown in Figure 1-17. When applied voltage is low (700 V/cm) the current distribution is almost uniform. When the voltage increases the current is concentrated into a few paths. In 2700 V/cm it is obviously seen that current is localized and most of the current flows from 2 paths. Then increase in the applied voltage can reduce the current concentration and make current distribution more uniform. As it is seen in 3900 V/cm, there are many carrying current paths in the varistor block.

Figure 1-17. Current delocalization phenomenon in simulation [18]

Chapter 1: Introduction 17

1.2.5.3. Cracking

Cracking, like puncture, is caused by high current concentration which results from varistor’s microstructural non-uniformity and heterogeneous power dissipation and heating in different portions. High amplitude short duration current surges can quickly inject a high amount of energy into a varistor block. In a short time of an impulse surge, energy diffusion does not happen inside the varistor, and the heating process can be considered adiabatic. Therefore, a thermally insulated temperature rise in a part of the varistor creates temperature gradients between different portions inside the varistor. Different rates of expansion in different parts of the varistor create thermal stresses on the grain boundaries between ZnO grains. If these thermal stresses inside the varistor exceed the stability strength of the ceramic, cracking destruction happens. The critical thermal stress (fc) which causes varistor cracking is in the range between

17.2 and 48.3 MPa. Figure 1.18 shows examples of varistor disks, failed in cracking mode.

Figure 1-18. Cracking failures in MOVs [26], [27]

1.2.5.4. Flashover

Flashover is current concentration on the edge of the MOV, instead of a current flowing through it. In low-voltage varistors, usually an insulating coating covers the whole body of the MOV and prevents a flashover. Any failure of the coating due to high temperatures during discharge activities, humidity, pollution, or material defects can lead to the flashover failure in the MOV. It is conventional wisdom that high 4/10 µs currents will cause surface flashovers (at magnitudes somewhat larger than 65 or 100kA) [28].

It is also shown in the literature that multi-pulse strikes and sustained over-voltages can cause flashover in very lower currents comparing to single pulse surges that are common in standard tests [28], [29].

Chapter 1: Introduction 18

It is a well-known fact that most of the lightning strikes have multiple strokes and up to ten strokes are quite common. The time interval between these strokes are 15 to 150 ms with an average value of 40 ms.

Darveniza and Saha [28] found that most of the distribution arresters tested by multi-pulse flashes at rated current or above fail by flashover of the varistor blocks which sometimes involved a little of the metal- oxide material as well. Figure 1.19 shows some MOVs failed by flashovers.

Figure 1-19. Flash over tracks on MOVs [26]

The probable causes of surface flashover of varistors are [29]:

a) plasma created near the edge of metallization;

b) humidity affecting the varistor outer surface;

c) material defects at the outer edges of the varistor;

d) partial discharges produced at the edges and surface possibly affecting the dielectric strength.

1.2.6. Effective Factors in MOVs’ Failure Modes

Depending on the condition of the varistor itself, ambient conditions, system specifications and surge parameters different failure modes may occur in MOVs. Although there is extensive research on MOV’s energy handling capability and failure modes [17]–[19], [23], [24], [30]–[36], not all of the effective factors are studied or well characterized. Limited surge research laboratories and high cost of tests, differences between standard tests and real-world application conditions, lack of accurate models for lightning and switching surges, and randomness of MOV’s behavior due to its complicated microstructure may be some of the reasons that hinder the research in this area. In this section we will review some of the papers

Chapter 1: Introduction 19

concentrating on the effective factors in failure modes and energy handling capability. The influencing factors are:

1) Surge Magnitude and Duration

The dependence of the failure mode on magnitude and duration of the current pulses is well studied by

Eda [19]. As it is seen from Figure 1-20, he showed that in short duration high currents the dominant failure mode is cracking and in long duration low currents is puncture. However, [19] reported that some puncture failures are caused by short duration high currents and some cracking destruction are seen in long duration low current tests. Similar results are reported by [18]. Simulation of thermal behavior of the varistor and solving heat diffusion equations and calculating the highest temperature and produced thermal stresses due to temperature differences in hot spots confirms and explains these results as well [17], [37].

Figure 1-20. Effects of magnitude and duration of the pulse on 14-mm varistor’s failure mode

2) Microstructural Non-uniformity of the MOV

As mentioned previously, microstructure of an MOV is random and non-uniform. When non-uniformity is high, it is predictable that current concentration and formation of hot spots and paths inside the varistor will cause puncture or will crack the varistor due to thermal stresses [18].

Varistor disks with hot spots of low intensities or very small effective area (or diameter) of the hot spot usually do not crack and can only fail by puncture. When the area or intensity of hot spot increases the

Chapter 1: Introduction 20

cracking, failure is seen in high currents. For higher non-uniformities, cracking failure is seen at lower currents, too [32].

3) Varistor Size

In general, for a varistor, current handling capability is proportional to the cross-sectional area and the energy handling capability is proportional to the volume, because the bigger the varistor is, the more heat it can absorb or dissipate. On the other hand, when cross-sectional area of the varistor increases, the probability of non-uniformity in the microstructure increases, too. So, the energy absorption capability per unit volume decreases accordingly. This can be seen in Figure 1-21 [18] where 50Hz voltage is applied to varistors of two different sizes with diameters of 32 mm and 52 mm. The results show a dependency on varistor size when varistor size is increasing EAC per unit volume decreases that and cracking failure becomes dominant.

Figure 1-21. Energy absorption capability for different varistor sizes [18]

4) Varistor Coating

In metal oxide varistors an external flashover can occur in currents well below the nominal discharge currents. So, a high resistance insulation collar coating of glass or other insulating material is applied on the peripheral surface. In a small varistor, an insulating coating covers the whole body as well as portions of the connecting leads in order to prevent flashover. The thickness of this coating is important in failure modes and energy absorption capability.

The coating on a varistor, while providing the electrical insulation, can act as a heat insulator too, preventing the heat from dissipating into the ambient air, thus reducing the energy absorption capability.

Chapter 1: Introduction 21

On the other hand, if the coating is too thin, it cannot provide enough insulation in higher currents and flashover takes place. So, the varistor cannot handle high currents, i.e. the energy absorption capability reduces. There is an optimal thickness for each coating depending on the material and the varistor size and specifications.

Based on this reasoning, it is predictable that when the coating is thin the dominant failure mode is flashover. The experimental tests [38] verify this. It is seen from Figure 1-22 that when the coating’s thickness increases from 100 µm to 340 µm the failure mode changes from surface flashover to other failure modes involving the ceramic itself. Furthermore, we can see that when there is an increase in the thickness of the coating, first the energy absorption capability increases but after a point it starts to decrease.

Figure 1-22. (a) The effect of coating thickness on failure mode (b) Energy absorption capability [38]

5) Varistor Contacts

Electrodes in the MOVs can be a source of failure. The impulse currents of high density can generate a large amount of heat and melt the material. Puncture at the edge of the electrode is among the most common failure mechanisms at high currents [25]. Furthermore, different thermal expansion coefficients of ceramic body and metal termination plate can cause mechanical stress and cracking along the edges of the metal plate. Two failed MOV whose failures are related to the electrodes are seen in Figure 1-23.

Chapter 1: Introduction 22

Figure 1-23. Two failures related to the electrodes [25]

The margin between the edge of the electrode and the edge of the varistor is one of the important factors in the varistor failure [39]. If the electrode extends to the edge of the varistor the current distribution in the varistor would be uniform, while the dielectric withstand capability of the varistor would decrease substantially through surface flashovers. Increasing this margin, on the other hand, will improve the flashover problem. However, spreading the current from the edge of the electrode will cause a high current density on the edge, and this will lead to high temperature difference between the elements near the electrode edge and other portions. Figure 1-24(a) shows the equithermal plots and temperature profiles, for an impulse current with 250 J/cm3 heat dissipation. In Figure 1-24(b) it is seen that when the margin is 2 mm the temperature difference between the area near the electrode edge and other areas are about 250 °C.

The highest temperature is 650 °C right under the edge of the electrode. This high temperature can be reduced to 440 °C, shown in Figure 1-24(d), by reducing the margin from 2 mm to 0.2 mm.

Figure 1-24. Equithermal plots and temperature profiles, for 250 J/cm3 heat dissipation [39]

Chapter 1: Introduction 23

Another problem of the electrode that reduces the energy absorption capability is protrusion of the metallization in axial direction into the ZnO cavities and defects in the edge of the metallization (non- smooth edge and radial protrusion toward the edge of the varistor) [20], [39]. Radial protrusion toward the edge, which is common in metal sprayed electrodes, can change the current distribution and build a hot spot near the edge. Protrusion of metal electrodes into ZnO cavities is a more severe problem that can contribute to puncture failure.

1.3. Problem Statement

High frequency overvoltages in Variable Frequency Drive (VFD) systems have adverse effects on motor and converter operation and lifetime. Also, high voltage spikes cause difficulties in SPD installation and lower surge protection performance. Originally, MOVs are built to be used in power system frequencies. In normal voltages and low frequencies of 50-60 Hz, an MOV conducts only a small current.

Over its lifetime, an MOV experiences a limited number of large current discharges due to lightning surges and TOVs. After every surge, it is assumed that the MOV has enough time to cool down to the ambient temperature and be ready for the next surge. However, in recent years with expansion of power electronics in the electrical environment and wide application of VFDs and inverters, MOVs are going to be more and more installed on lines with high frequency PWM voltages. In such networks, MOV may conduct high capacitive and resistive currents for a short duration in every pulse. Under such repetitive pulses, the MOV does not have time to cool down after each pulse. Also, lead and wire inductances may have considerable effects and may not be neglected. In this research, we investigate the MOV behavior in such high frequencies, analyze and characterize the related problems. Moreover, we develop and evaluate new approaches to allow MOVs to operate safely on the line for overvoltage protection and at the same time decrease the overvoltage in VFD systems.

This research will give insight to MOV behavior in high frequencies to SPD design engineers and facilitate the application of the MOV in VFD systems as a surge protection device. The results of this research will influence industry sections that require variable speed operation such as HVAC systems.

Application of MOVs will reduce damages due to external surges, while extending lifetime of the system by overvoltage mitigation.

Chapter 1: Introduction 24

In accordance to this vision, the main objectives and contributions of this research are as follows:

Objective 1: Develop a method for health-monitoring of Metal Oxide Varistors (MOVs)

A thorough literature review on MOV’s failure modes are conducted and all the influencing factors are studied and discussed.

A method for health-monitoring of Metal Oxide Varistors (MOVs) is developed. The proposed method uses resistive leakage current as an indicator parameter to show the status of health of MOV that can be used to alarm the user when the MOV is no longer reliable. Moreover, this method calculates the estimated remaining lifetime of the MOV. This gives the user the ability of proactive maintenance and replacement of the MOVs before they fail. In order to further increase the accuracy, the proposed method, uses MOV temperature and under/overvoltage conditions information. Furthermore, considering the behavior of

MOVs and in order to reduce the calculation error, the proposed method takes measurement errors into account.

Substantial tests on degradation of more than 150 samples of low voltage MOV are conducted and published for the first time. Surprising findings on Energy Absorption Capability and Time to Failure of

MOVs are observed, investigated and analyzed. These findings give insight to degradation of low voltage

MOVs for application engineers and manufactures.

Objective 2: Analyze, model and explain the reasons for overvoltages in Variable Frequency

Drives (VFDs)

First, a literature review on Variable Frequency Drives (VFDs) was performed to identify their possible surge related problems, and traveling wave related overvoltages. Characterization of high frequency overvoltages for the VFD system for different cable lengths, and different pulse rise times are described and modeled.

An experimental VFD setup, including a rectifier/inverter, a cable, and a three-phase motor, is built in the lab. High frequency models for cables and motors are studied. Based on the models, impedances of the

Chapter 1: Introduction 25

existing cable and motor are measured and their model parameters are extracted. The models are simulated, and results are verified by the experimental measurements.

Objective 3: Develop and evaluate a new method for surge protection in Variable Frequency

Drive systems.

A literature review of overvoltage mitigation techniques, including a comparative study of existing filters, is conducted. MOV models are presented and examined, and MOV’s behavior is studied under overvoltages. A model is built for the MOV that is used in this research. The effects of installing MOVs are analyzed and presented through simulation studies and the measurements on the experimental setup.

Two new surge protection devices are proposed and simulated. These methods reduce the let-through voltages on the VFD systems due to surges and high overvoltages. The effectiveness of the proposed devices is verified by simulations and the advantages and disadvantages of the newly proposed methods are discussed in comparison to the existing method.

1.4. Dissertation Organization

This introductory chapter provides a background on MOVs and explains the definitions and basic concepts in surge protection. Also, it delivers a literature review on MOV failure modes and factors that influence these modes. Furthermore, problem statement and objectives of the thesis research are explained.

In Chapter 2, degradation of the MOVs are investigated. Health monitoring methods are studied, and a new health monitoring and lifetime prediction method is proposed. Chapter 3 explores the high frequency overvoltage phenomenon in VFD systems and describes the experimental setup that is built in the lab, at

Northeastern University. This chapter, also, includes the modeling techniques and measurements that are used to simulate the VFD system. In Chapter 4, surge protection principals of a VFD system is described.

Also, overvoltage mitigation techniques are reviewed, and advantages and disadvantages of each approach are explained. In the final chapter, Chapter 5, MOVs behavior is investigated under pulses and PWM systems and it is explained that how overvoltages prevent an effective surge protection to be installed in the

VFD systems. Two new solutions to the problem of application of MOVs in VFD systems are developed,

Chapter 1: Introduction 26

simulated, and discussed in detail. Effectiveness of the proposed methods are demonstrated in comparison to the existing solutions.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 27

Chapter 2:

2. Degradation and Health Monitoring of Metal Oxide Varistors

In this chapter, a literature review on Energy Absorption Capability (EAC) of MOVs is presented.

Then, two experimentally conducted tests on new and degraded low voltage MOVs at Mersen in

Newburyport, MA, are described, and their results of EAC and Time to Failure (TtF) are analyzed. The research represents the first study on EAC of degraded low voltage MOVs that has been published publicly

[40]–[43]. EAC in AC and impulse currents are measured for the MOVs, and the results are compared with high voltage station-class MOV results (because there is no other data available on low voltage MOVs).

Furthermore, MOV health monitoring techniques are investigated and effects of temperature and voltage are studied on health identification parameters. Finally, a method for health monitoring of MOVs is proposed.

2.1. Energy Absorption Capability

EAC is defined as the amount of energy that an MOV can absorb in an incident of a surge or a TOV before it fails. This failure can be considered MOV’s mechanical destruction [17], [18], [30] or a predefined change in its electrical parameters [44]. It is suggested in the literature that EAC is increasing with increasing current density [17], [30]. Figure 2-1 shows EAC of station-class varistors vs. test current,

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 28

measured by experimental results in [30] and verified by simulation in [17]. It is seen that average EAC increases from 500 J/cm3 in AC currents around 1 A to above 1500 J/cm3 for pulse currents around 35 kA.

Figure 2-1. EAC of station class varistor disks vs. test current [17]

These experimental tests are conducted on 241 station-class high voltage MOVs, 23 mm high and 63 mm in diameter, with voltage rating of 3 kV [30]. Simulations yielded similar results on station-class

MOVs, as well as distribution-class MOVs of 45 mm high and 32 mm in diameter [17], [37].

Some explanations are presented in the literature to explain the increase of EAC with increase of current density. It is attributed by [45] to voltage distribution on ZnO grains and grain boundaries. At lower currents, most of the voltage drop appears on grain boundaries. Thus, most of the energy loss occurs on small regions of the microstructure. This non-uniform heating creates a temperature gradient and thermal stress inside the microstructure that causes failure. At high surge currents, on the other hand, the voltage drops increase on ZnO grains and energy loss and heating are more uniform throughout the microstructure.

However, simulations by [37] reject these explanations by showing that heat transfer at the grain size scale is too fast to allow any noticeable temperature difference between a grain and its boundaries. They conclude that increase of EAC at high current densities is a consequence of de-localization of current in the varistor. At higher currents, the voltage drop on the varistor is enough to overcome the threshold voltage of boundaries. Thus, current distribution is dependent on the grain resistances. This leads to more uniform current densities and less current localization. This is also shown by simulations in [18]. Current

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 29

distribution is more uniform in low currents (low voltage drop E=700 V/cm). Current localization increases to some point, when current increases. Then, at higher currents more current passes are created and current distribution is more uniform.

Another phenomenon that has received less attention is that, at low currents, initially EAC decreases with increase in current density [37]. This is again justifiable based on current localization as it is described in chapter 1.

(a) [18] (b) [37]

Figure 2-2. Energy Absorption Capability vs. current density (a) experimental results (b) simulation results for different failure modes

Experimental results in [18] show higher EAC at very low currents as well. Figure 2-2(a) shows EAC decreases from 550 J/cm3 at current density of 2 A/cm2 to around 250 J/cm3 at 10 A/cm2. Furthermore, simulations in [17] verify these results. Simulation results in Figure 2-2(b) shows the energy needed for any failure in a distribution-class arrester. It is seen that EAC is higher for low current densities.

In order to fulfill the objectives of the research, we conducted a large number of EAC tests both on new and degraded MOVs. More than 150 low voltage MOVs are used in the experiment. The specifications of the MOVs are summarized in Table 2.1. Figure 2-3 shows the size of test MOVs in comparison to the station-class MOVs used in the references. All the tests are conducted in a UL-certified lab at Mersen USA.

The EAC on the new MOVs are conducted at low density ac currents as well as high density impulse surge currents. For the ac current test, a constant overvoltage of 240 V is applied to twenty MOVs until their destruction. For the high current densities, single 8/20 µs impulse currents are applied to MOVs.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 30

Table 2-1. Specification of Test Specimens

MOV Voltage [V] Size [mm] Max Surge Current Max Energy [J] Power Dissipation [W] LS40K150QPK53 150 33.5 x 33.5 x 2.5 2 * 40 kA + 20 * 20 kA 360 1.4

63 mm

33.5 mm

45 mm -

23 2.5mm

Figure 2-3. Sizes of the station-class varistors [17, 28] (right) and studied low voltage MOVs (left)

2.1.1. EAC test results on new MOVs

Our test results indicate that EAC of tested low voltage MOV is decreasing when the current density increases. This is different than previously reported EAC for station-class MOVs. AC EAC test results, which are summarized in Table 2.2, show that EAC of low voltage MOV is in the same range of those of station-class MOVs reported in [44]. The average EAC per unit volume for 150-volt MOV is 954 J/cm3 when the average test current density peak is 7.8 A/cm2.

Table 2-2. AC EAC Test Results of New 150V MOVs

Average Max Min Standard Deviation EAC [J/cm3] 954 1142 691 143.7 RMS Test Current [A/cm2] 3.2 6.8 0.6 1.7 Average Peak Current Density[A/cm2] 7.8 16.0 1.6 3.8 Time to Failure [s] 0.67 2.91 0.18 0.60

Table 2-3. Impulse EAC Test Results of New 150V MOVs Sample # Surge Current [kA] Energy [J] Result A00 133 4924 Failed A0 126 4133 Failed A1 100 2493 Failed A2 90.4 1904 Passed A3 90.4 - Failed A4 84 1800 Failed A5 80.8 1577 Passed A5 80.8 1607 Passed A7 80.8 1549 Passed EAC [J] 1578 EAC [J/cm3] 574

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 31

1000 ]

3 954

500 574 EAC [J/cm EAC 7.8 7200 0 1 10 100 1000 10000 2 Current density [Apeak/cm ]

Figure 2-4. Energy Absorption Capability curve for studied MOV

The Maximum 8/20 µs surge that MOV samples can handle is 80 kA. The average energy absorption during 80 kA surges is 1578 Joules or 574 J/cm3, which is approximately one third of the values reported by references [17], [30], [44] for station-class MOVs. The Impulse EAC test results are summarized in

Table 2.3.

Analysis of the failure modes of the test samples provides more insight into the test results. All of failed samples are examined to determine the failure modes after the test. The main failure modes in impulse tests are cracking and coating peeling. In impulses higher than 120 kA, the MOVs crack into several pieces.

Usually, cracking occurs outside of the electrode area and a piece under the electrode does not crack. It seems that the heat generated in the electrode might be a contributing factor to this failure mode.

Furthermore, electrodes provide mechanical reinforcement for the portion between them, while the portions outside the electrodes are free to vibrate. In lower surge currents, the ceramic body does not crack, and the main failure mode is coating peeling. Epoxy coating is peeled off mostly in one side, where a round electrode lies between coating and the ceramic body. Again, excessive heat, generated in the electrode is the main culprit. Also, the footprints of surface flashovers are seen in some cases.

In low density AC current tests, the main failure modes are puncture and cracking. In samples with lower current densities, puncture is the only failure mode. When the test current is low, which is the case in

MOVs with high varistor voltage (V1mA), Time to Failure (TtF) is long enough for current concentration

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 32

(a) (b) (c) (d)

Figure 2-5. Dominant failure modes at (a) high impulse; (b) low impulse; (c) high ac; (d) low ac currents and formation of a puncture failure. Puncture is seen in different areas (under, outside and also on the edge of the electrode) of the ceramic in different samples. In samples with higher AC currents, cracking is seen beside puncture. The cracking failure here is different than the cracking caused by impulse current, and in most cases, it splits the whole ceramic body into two or three pieces.

Non-uniformity of the microstructure of the MOV explains the failure modes. In high impulse surges, the current is concentrated into a few paths due to non-uniform microstructure, creating hot spots and paths

[31]. Thus, high temperature gradient between grains leads to ceramic cracking. However, in lower currents, hot spots reach the melting temperature of the ceramic, before it cracks, leaving a hole through the ceramic body. In surge currents that do not crack the MOV, the rapid heating of the electrode and its expansion may cause the epoxy coating to crack and leave the MOV vulnerable to surface flashovers.

Since the EAC test is conducted with a constant voltage of 240 volts AC, the current is not controlled.

The current depends on the generator impedance and the V-I characteristics of the MOV. In order to investigate EAC of MOVs in low ac current densities, 12 new MOVs are subjected to lower overvoltages based on their varistor voltage. This range of current is important in characterizing the behavior of MOVs during a TOV. AC EACs of all new MOVs (32 in total) are plotted versus peak current density in Figure

2-6. In lower current densities around 0.7 A/cm2, the average EAC of new MOVs is 1413 J/cm3. This finding is in accordance with previous studies as described before. When the current density is low and the time to failure is long, the locally generated heat in the varistor uniformly conducts to the whole varistor, so the possibility of formation of hot spots is very low. As a result, the energy needed for destruction (i.e.

EAC) would be high.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 33

Figure 2-6. EAC of New MOVs

One possible fit for the measured data is as below:

EACNew = Ebase + E0 exp(− I peak ) Equation 2-1

Ebase, E0, and λ are constants. Ipeak is the peak value of the MOV current. For the low-density currents less than 16 A/cm2, this curve is easy to use and beneficial in study of degradation effect. In this equation, the constant term, here called the base EAC, is almost equal to the average EAC of the group for current densities higher than 2 A/cm2. For example, for new MOVs shown in Figure 2-6 EAC would be:

EACNew =947.6+705exp(−0.821I peak ) Equation 2-2

2.2. Degradation of MOVs

Metal oxide varistors are known to degrade over time when they experience surges and overvoltage transients. Generally, the leakage current of the MOV increases with degradation, thus the power dissipation increases within the varistor. Other parameters of the MOV such as varistor voltage change as well, with degradation [23], [46]. Degradation in MOVs occurs because of the changes in their microstructure. A high magnitude current surge can change the electrical properties of Schottky barriers at grain boundaries. Normally, degraded MOVs show oxygen deficiency in the grain boundaries that lower the height of barriers and increase the leakage current [47]. Also, Bismuth concentration in small areas of pulsed varistors is observed [48]. The reduction in average grain size and change in the lattice parameters have been also reported. Inhomogeneity in grain size and threshold voltages can increase the current concentrations [47]. These changes can be either asymmetrical (polarized) and change the barrier in one

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 34

direction more than the opposite direction, or symmetrical and change both sides equally [23]. Their collective effects alter the current-voltage characteristics. It is believed that degradation mostly affects the leakage current region of the V-I curve [49].

Furthermore, it should be noticed that these changes are stochastic processes and each MOV varies differently. Thus, it is necessary to use multiple test samples in MOV degradation tests to reduce the possibility of neglecting a particular behavior that might be seen only in a fraction of each production batch.

Degradation or aging of MOVs due to surges depends on the number of the surges, current magnitude discharged by the varistor, the duration of the current and its wave shape, the temperature of the varistor, the formulation of the varistor, and the polarity of the surge and varistor’s previous polarization.

In order to investigate the effects of surge degradation on EAC and TtF of low voltage MOVs, we conducted substantial tests on surge degradation of MOVs with different number and polarity of surges.

Two sets of MOVs are prepared for this test. The first set (Set A) consists of four groups as described in

Table 2.4. The MOVs of Groups 1 to 3 in Set A are commercially available (CA) ones with no thermal disconnect terminal. Group 4 MOVs are Custom Made (CM) MOVs that have the same size and specifications with one thermal disconnect terminal which is clipped to remain close for all tests. The only difference between Set A and Set B MOVs in their datasheets is that maximum surge is one 40 kA surge for CA MOVs and two 40 kA plus twenty 20 kA surges for CM ones. The second set (Set B) is composed of 6 groups of CM MOVs as described in Table 2.5.

The main findings of these tests are described as follows:

1. Average EAC of MOVs decrease with surge degradation.

Decrease in EAC is different in two sets. In Set A, EAC of degraded MOVs after 6 surges of 8/20 µs decreases 40% from 350 J/cm3 to 211.9 J/cm3. Considering the worst-case scenario, minimum EAC of degraded MOVs is 80.2 J/cm3 which shows 66% decrease from the minimum EAC of new MOVs, i.e.

236.2 J/cm3. On the other hand, in Set B, comparing Groups 5, 6 (unipolar degradation) and 10 shows that average EAC reduces 5.5% for Group 5 and 9% for Group 6, after 20 surges of 40 kA. These changes are

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 35

less than what is seen for CA MOVs. This indicates that the CM MOVs are more reliable and resistant

against surge degradation. Furthermore, for the worst case in Group 6, an EAC reduction of 9.5% is seen.

Table 2-4. Set A Samples and Test Results

Minimum Standard Set A number of Average Average Peak Current Degradation (8/20 µs Surge) EAC Deviation Group samples EAC [J/cm3] Density [A/cm2] [J/cm3] [J/cm3] 1 11 none 350.0 236.2 236.2 3.6

2 11 10% change in V1mA 345.8 195.6 195.6 2.5 3 10 6 impulse of 40 kA 211.9 80.2 80.2 1.0 4 10 none 1081.0 850.0 850 4.2

Table 2-5. Set B Samples and Test Results

Standard Average Peak Set B number of Average Minimum Degradation (8/20 µs Surge) Polarity Deviation Current Density Group samples EAC [J/cm3] EAC [J/cm3] [J/cm3] [A/cm2] 5 20 2 * 40 kA + 20 * 20 kA 900 660 161 5.0 Unipolar 6 10 20 * 40 kA 870 626 170 2.6 7 10 2 * 40 kA + 20 * 20 kA 953 701 234 3.3 Bipolar 8 10 20 * 40 kA 1036 807 165 1.2 9 20 none 954 691 144 7.8 10 12 none 1413 1029 326 0.7

Figure 2-7. EAC for Groups 5, 6, 7, 8 and new (9-10) MOVs

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 36

In order to understand the degradation effect better, EAC of Groups 5 and 6 is plotted alongside that of new MOVs in Figure 2-7. The fitted curves can be expressed as below:

EAC5 =885.8+470exp(−1.15I peak ) Equation 2-3

EAC6 =821.7+470exp(−1.15I peak ) Equation 2-4

It is noticed that Ebase has reduced to 885.8 for Group 5 and 821.7 for Group 6 from 947.6 for new

MOVs. Then, we can see 6.5% for G5 and 13.3% for G6. These numbers are a little higher than those acquired from average values. The advantage of using Ebase is that it eliminates the effect of different test currents, i.e. if a test has more samples in low current area, the average will show higher values, while Ebase will not change noticeably, since it comes from curve fitting. Including such information into the varistor datasheet by manufacturers, will give more insight into product surge capability, compared to the commonly used method of defining only maximum surge capability.

The importance of using Ebase becomes more obvious when we look at the bipolar degradation tests. It is seen from Table 2.5 that bipolar surges increase the EAC of MOVs. Although it is true when considering a constant overvoltage and different currents, using Ebase in equations (2.5) and (2.6) and curves inFigure 2-7, we realize that for a given current, EAC decreases 10.6% with twenty 40 kA bipolar surges.

EAC7 =886.2+470exp(−1.15I peak ) Equation 2-5

EAC8 =847.2+470exp(−1.15I peak ) Equation 2-6

2. Unipolar surges decrease EAC more than bipolar surges.

In degradation with unipolar surges, all surges are applied in one polarity, while in bipolar degradation, after every surge application, the polarity of surge changes for the next surge. Typically, varistor voltage decrease and leakage current increase are higher in the opposite direction of a surge. The results in Table

2.5 and equations (2.4) and (2.6) show that unipolar surges decrease EAC more than bipolar surges.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 37

3. Average EAC of CM MOVs is much higher than CA MOVs.

Comparing the average EAC of Groups A1 and A4 shows that the average EAC of CM MOVs is almost 3 times that of CA MOVs. Average EAC of CA MOVs is 350 J/cm3 that surprisingly low compared to average EAC of new CM MOVs or values reported in [17], [30] for station-class MOVs. This shows that some commercial products might have very similar nominal specifications, while they have a significant difference in their surge capabilities. Part of this may refer to lack of a standard definition for EAC of low voltage MOVs.

4. Pulsed MOVs conduct lower currents during a TOV.

It is believed that degradation mostly affects the leakage current area of V-I characteristics. However, there are mentions of changes in TOV regions, also in [49]. Our test results indicate that while surge degradation increases the leakage current of the MOV, it decreases the MOV current in the TOV region. It is seen from Table 2.5 that average peak test current for the new MOVs of Group 9 is 7.8 A/cm2, where it is

5.0 A/cm2 for Group 5 and 2.6 A/cm2 for Group 6. MOVs with higher degradation level conduct less current when a TOV occurs. This is also true for Groups 7 and 8 with bipolar degradation. Furthermore, the table shows that bipolar surges decrease the test current more than unipolar surges. This is partly because the bipolar surges, typically, increase the varistor voltage in both directions while unipolar surges decrease the varistor voltage at least in the opposite direction of the surge.

The change in the MOV’s test current is attributed to change in the TOV region of its V-I characteristic.

Figure 2-8 illustrates the change of V-I curve due to surge degradation. The change in TOV current affects the EAC of the MOV as well as its TtF, which is described in the next part.

Figure 2-8. Degradation effect on MOV’s V-I curve

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 38

5. Surge degraded MOVs have longer average Time-to-Failure (TtF) than new MOVs for a given

TOV.

Time to Failure is defined as the time from the application of 240 V overvoltage until the MOV’s failure. Surprisingly, our test results show that the degraded MOVs have longer average TtF than new ones.

This means that surge degraded MOVs tolerate TOVs longer than new MOVs. This is reflected in Table

2.6. As an example, Group 6 MOVs after experiencing 20 nominal surges have average TtF of 2.24 seconds, which means they tolerate a 240-volt TOV almost 3 times longer than new MOVs. Furthermore, comparison of Groups 5 and 7 shows that bipolar surges increase the TtF more than unipolar surges.

Again, for a thorough investigation, TtF of different groups are plotted and proper curves are fitted. The linear dependence between logarithm of current and logarithm of TtF of new MOVs is mentioned in the literature [17, 28, 38]. Figure 2-9 shows the TtF of Groups 5 and 6 compared to that of new MOVs.

Table 2-6. Time to Failures of Set B

Set B Average Minimum Standard deviation Group TtF [sec] TtF [sec]

5 1.15 0.3 1.07 6 2.24 0.66 2.04 7 3.17 0.43 3.33 8 10.64 1.11 9.12 9 0.67 0.18 0.60 10 28.2 13.2 20.1

(a) (b) Figure 2-9. Time to Failure for degraded and new MOVs

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 39

As it is seen from the Figure 2-9(a), TtF decreases at a given current with degradation. Furthermore,

Figure 2-9(b) shows average TtFs, as well as box plots and outliers for new MOVs of G9 and four degraded MOVs of Groups 5 to 8. The important point is that for a certain TOV, here 240 V, the average

Ttf for the bipolar degradation is higher than that of unipolar surge degradation.

2.3. Degraded MOV’s V-I curve

A simple and widely used model for the MOVs comprises of a nonlinear resistor in parallel with a capacitor. Degradation changes the V-I characteristics. However, the changes in the separate regions are completely different. In leakage current region, degradation increases the resistive leakage current. This has been reported in the literature [46]. Also, nonlinearity factor, α, decreases by degradation [23]. Our test results on the clamping voltage of low voltage MOVs show that degradation has little effect on the clamping voltage in high surge currents. Thus, the up-turn region should be considered unchanged with degradation. This is in accordance with the observations in [44] and [49]. Considering the changes in the discharge/TOV region is important when considering the behavior of the MOV in TOV conditions. The results presented in this document show that the VI characteristic moves toward lower currents for a given voltage in TOV region. In order to incorporate all the above-mentioned changes in a model, we propose to use the below curve to fit on the V-I data points of the degraded MOV.

ퟐ 풍풐품 푽 = 풃ퟏ + 풃ퟐ 풍풐품 푰 + 풃ퟑ 풆풙풑(− 풍풐품 푰) + 풃ퟒ 풆풙풑( 풍풐품 푰) + 풃ퟓ(풍풐품 푰) Equation 2-7

Figure 2-10. Fitted curve on degraded MOV V-I characteristic

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 40

2 When b5 = 0, the proposed curve reduces to the curve in Equation 1-2 for new MOVs. b5(log I) allows the curve to fit better in the TOV region. Using this curve fitting in simulations, allows having more realistic behavior in TOV condition for degraded MOVs.

2.4. Health Monitoring of MOVs

With the emerging concept of the smart grid and growing attention to the asset management and lowering the down time of services, monitoring of the equipment seems to be an indispensable need in the near future. Although the failure of a thermally protected varistor does not disconnect the load, it leaves the sensitive load un-protected and the next surge or transient can damage the load.

Considering the lifetime of an MOV, which is normally more than ten years, the possibility of aging and failure is not negligible. Despite the progress in the development of MOVs during past decades, a reliable assessment of their health has not been proposed [48]–[51]. A number of techniques have been proposed for online condition monitoring of MOVs. Resistive leakage current monitoring is a common method [52]. Furthermore, total leakage current [53], third and higher harmonics of the total leakage current [54] and power loss in the MOV [49] have been proposed for MOV health monitoring purposes.

Temperature monitoring has also been used for online monitoring of high voltage metal-oxide arresters

[55].

Different End-Of-Life (EOL) criteria are used by different companies such as leakage current or power loss reaching double the initial values [56]. In addition to online monitoring techniques, a number of offline diagnostic methods have been also proposed for MOV health assessment. Destructive approaches such as

Scanning Electron Microscopy (SEM) [47], and X-ray Diffraction (XRD) [48] and non-destructive methods such as reference voltage measurement [44], discharge voltage measurement [48], and partial discharge measurement [57] have been proposed in the literature.

Varistor Voltage (V1mA) basically indicates only one point on the V-I characteristic. It has no particular electro-physical significance but is commonly used as a practical standard reference by manufacturers and engineers. Usually, 10% change in V1mA from its initial value is considered as unacceptable [58].

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 41

Leakage current monitoring is a common method in Metal-Oxide (MO) arrester condition monitoring

[59]. In normal system voltages, the leakage current of an MOV is a non-sinusoidal current and mainly capacitive with a small non-linear resistive component. Figure 2-11(a) shows the leakage current of a new

MOV and its resistive and capacitive components. Although the capacitance of the MOV changes slightly with degradation [60], the capacitive leakage current is generally considered to remain almost constant through the MOV lifetime. Resistive leakage current increases with aging. However, since it is small compared to the capacitive component, the change in total leakage current might be small. Figure 2-11(b) shows the leakage current of a degraded MOV and its resistive and capacitive components. RMS value of resistive leakage current is increased from 86 µA to 298 µA after twelve surges of 40 kA. Also, the peak value of leakage current has increased from 221 µA to 787 µA. Although the resistive leakage current increased 3.5 times, the rms value of the leakage current has increased 35%.

(a) (b) Figure 2-11. Leakage current (iL), and its resistive (iR) and capacitive (iC) components

The existing methods of extracting the resistive component of the leakage current are mostly based on orthogonality of the resistive and capacitive currents, such as harmonic analysis and compensation of the capacitive current [61]. Other methods like point-on-wave [52] are used to extract the resistive leakage current without the need for the voltage waveform.

Measurement of resistive leakage current and alternatively the power loss of the varistor, need both current and voltage waveforms. In order to eliminate the need of voltage measurement, third harmonic of leakage current can be used as an indirect measure of power loss in the varistor. Typically, changes in third harmonic of leakage current are similar to resistive current. However, the amplitude of third harmonic is

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 42

low, and its measurement is more susceptible to noises in a disturbed electro-magnetic environment.

Furthermore, the third harmonic of voltage, which is unknown without knowing voltage waveform, can cause a high third harmonic current and create a huge error in third harmonic measurement [39].

In order to investigate the changes in electrical parameters of MOVs with degradation, varistor voltage and leakage current of all MOVs of Set B (Table 2.5) are measured, and their third harmonics and power losses are calculated, and the resistive leakage currents are extracted. The method used for the resistive components’ extraction is the compensation of capacitive leakage currents for all voltage harmonics [61].

Figure 2-12 shows average reference voltages (V1mA) for Groups 6 and 8 of Set B. Voltages are normalized. Difference in degradation due to unipolar and bipolar surges is obvious in the figure. After a huge change due to the first 40 kA surge in Group 6, the second unipolar surge of the same size and same polarity does not change the characteristics remarkably. However, the second bipolar surge restores V1mA to values close to the initial values.

Figure 2-12. V1mA vs. number of surges for different groups

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 43

Figure 2-13. V1mA vs. number of surges for Group 5b

The maximum V1mA reduction seen in the samples of the Group 6 is 16 percent after 20 surges of 40 kA. The minimum reference voltage in Group 4 after 20 surges of 40 kA is 0.934 that shows less than 7 percent reduction. For Groups 5 and 7, it is seen that after initial 40 kA surges, subsequent 20 kA surges increase V1mA both in bipolar and unipolar surges. In order to compare the effect of the initial surge, Group

5 is divided into two sub-groups. Group 5a receives 40 kA surges first, while Group 5b first is subjected to

20 kA surges and 40 kA surges are applied after all twenty 20 kA surges. The result for the V1mA of the

Group 5b is seen in Figure 2-13. Comparison of two Groups 5a and 5b shows that when an MOV experience small surges in the beginning, the total reduction in V1mA is smaller. However, the forward voltage change might be higher. While the minimum V1mA in Group 1a is 0.88, none of the Group 5b samples have V1mA less than 0.975. These results suggest that it is not accurate to describe the degradation of an MOV simply by the change in its V1mA from its initial value. Especially, in the case of bipolar degradation, the change in V1mA is negligible, despite the large number of surges and probable changes in microstructure and voltage current characteristics. Main observations on the electrical parameters are summarized below.

1. Varistor voltage decreases with unipolar surges of nominal amplitude (40 kA) in the reverse

polarity.

2. Smaller surges (20 kA) mostly increase the varistor voltage in forward direction.

3. The first surge has higher effect on varistor voltage and resistive leakage current.

4. A surge in reverse polarity restores the varistor voltage back to values close to the initial values.

Thus, bipolar surge degradation does not decrease V1mA.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 44

5. Resistive leakage current increases with the first surge. However, it might either increase or

decrease by subsequent surges (in both bipolar and unipolar degradation).

Figure 2-14. IR vs. number of surges for different groups

Resistive leakage current (IR) changes with surges, too. The increase in IR is larger in unipolar surges, when MOV is degraded by large 40 kA impulse currents, as seen in Figure 2-14 for Group 6. It is also seen that the resistive leakage current decreases when 20 kA surges are applied after 40 kA ones. This trend is seen in both unipolar and bipolar surge degradation process.

2.4.1. Effects of temperature and voltage on health monitoring and lifetime estimation

The temperature effect on the leakage current of MOVs is known from early days [22]. Figure 2-15 shows a typical change of V-I curve for different temperatures. Leakage current increases with temperature.

Also, increase in the applied voltage increases the leakage current. When monitoring the leakage current or any of its components, the effects of temperature and over/under-voltage should be compensated.

Figure 2-15. Temperature effect on V-I curve of an MOV [50]

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 45

250

200

150 New, 20C

100 New, 100C Voltage Voltage (V) After 80*20kA, 20C 50 After 3*50kA, 20C

0 0.001 0.01 0.1 1 10 DC Current (mA)

Figure 2-16. Measured voltage current curve for a new and a degraded MOV in 20 °C and 100°C

Voltage-Current curve of MOVs are measured in the lab by a Keithley Sourcemeter 2410. Measured voltage-current curve for a new and degraded MOV is shown in Figure 2-16. It is seen that temperature effect is similar to degradation effect in leakage current region. In order to measure the effects of temperature and voltage on the low voltage MOVs, the leakage current of 10 new MOVs are measured in temperatures -20 °C to 85 °C under voltages of 104 V to 150 V. The results are shown in Figure 2-17 for new MOVs. These temperature and voltage dependency curves change with degradation. The curves for the same MOVs are shown in Figure 2-18, after degradation process that increased their leakage current to about 90 µA in room temperature at nominal voltage. For example, if this is considered the end of life of the MOVs, the numbers from Figure 2-18 give us the upper limits for leakage current in every voltage and temperature. Interpolation between initial points and the upper limits can provide a rough estimation of the remaining lifetime of the MOVs.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 46

(a) (b) Figure 2-17. (a) Temperature, and (b) Voltage dependency of resistive leakage current for a new MOV

(a) (b) Figure 2-18. (a) Temperature, and (b) Voltage dependency of resistive leakage current for a degraded MOV

2.4.2. Proposed Health Monitoring Algorithm

Online health monitoring and lifetime estimation of MOVs provides the opportunity to plan for proactive maintenance and reduces the system down time and operation cost by limiting the maintenance activities to only near-to-fail MOVs. Based on our tests and measurements, resistive leakage current is the best parameter to monitor the health of an MOV. Also, compensation for voltage and temperature is necessary. This research proposes the algorithm in Figure 2-19 to predict the approximate remaining lifetime of an MOV.

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 47

Capture leakage current and voltage waveform

Extract resistive leakage current

Measure voltage Extract Resistive leakage current and temperature lower and upper limits (Figures 2.17 and 2.18)

Estimate lifetime (Interpolation- Compare the measured value to the reference values)

Take the proper action (Output, Alarm…)

Figure 2-19. Proposed health monitoring algorithm for low voltage MOVs

The proposed health monitoring algorithm needs the MOV’s voltage and current waveforms, as well as its temperature, to estimate lifetime prediction. Sampling frequency should be high enough to extract the harmonics of voltage and current up to 5th harmonics. The higher the order of the harmonics considered in the algorithm, the more accurate the results will become. Resistive current is extracted as discussed previously. The minimum and maximum values of the resistive current are determined based on offline measurements on new and degraded MOVs, similar to Figure 2-17 and Figure 2-18. The upper limit is decided arbitrarily based on the field data of the failed MOVs. Here, we take the values shown in Figure

2-18 as the threshold values. The temperature of the MOV should be measured at the moment of the current measurement. The temperature sensor should be installed on or close to the surface of the MOV. The remaining effective lifetime of the MOV, can be calculated using the proposed algorithm.

For example, assume that the resistive current of an MOV is 200 µA, installed in a system with a nominal voltage of 150 V. Voltage measurement shows the voltage at the moment is 145 V, and the

MOV’s temperature is 40 °C. The lower and upper limits for these conditions can be interpolated from

Figure 2-17 and Figure 2-18. The lower limit, which is the expected resistive leakage current for a new

MOV, is 97.1 µA. The threshold value can be calculated from the degraded MOV curves. Here, it is 261.3

Chapter 2: Degradation and Health Monitoring of Metal Oxide Varistors 48

µA. Interpolation of the 200 µA in this range, gives us that more than 60% of the MOV’s lifetime has passed and only about one third of the lifetime is remaining.

remaining lifetime percentage 100% 37.4% 0%

resistive leakage current 97.1 µA 261.3 µA 200 µA

Figure 2-20. Lifetime calculation of the MOV

2.5. Summary and Conclusions

In this chapter, we presented our experimental test results on EAC and TtF of new and degraded low voltage MOVs using a large number of samples. This is the first study on EAC of degraded low voltage

MOVs that has been published publicly. Our results have appeared in [40]–[43]. Surprising findings are presented. In particular, it is found that EAC of low voltage MOVs increases in low current densities. Also, it is found that TtF of degraded MOVs are longer than that of new MOVs in TOV condition. Degradation effects are explained on the behavior of the MOV and its parameters. Impulse EAC of the low voltage

MOVs are measured by experiment. It is found that impulse EAC of low voltage MOVs are lower than their AC EAC, contrary to station-class MOVs reported in the literature. Furthermore, different health monitoring methods are presented, and their advantage and disadvantages of them are explained. Effects of temperature and voltage on MOV’s electrical parameters are measured and used for estimation of remaining lifetime of the MOV. Finally, a lifetime estimation algorithm is proposed.

Chapter 3: Overvoltages in VFDs 49

Chapter 3:

3. Overvoltages in VFDs

In this chapter, a literature review on voltage wave reflection phenomenon and resulted overvoltages on motor terminals is presented. The effects of overvoltages on the motor, the drive and electric system are explained. Mathematic formulas are used to calculate the relation between overvoltages, cable length and voltage rise time. Cable and motor models are discussed, and simulation results are presented.

3.1. Introduction

Electric motors have been the main provider of mechanical energy in industrial and commercial applications since their invention in 19th century. Specifically, the fixed speed three-phase squirrel cage induction motor has been used extensively in industry, because of their rigidity and reliability. The speed of an induction motor depends on only the number of its stator poles and the frequency of the supplied voltage. Before the advent of SCR1-based speed control solutions, mechanical methods were used to change the speed of the motor. These methods were usually bulky and inefficient systems that could not provide continuous speed control over the desired range. Lack of an easy and reliable method to control the speed was considered as a major disadvantage for induction motors. After development of AC variable frequency drives in 1960s and 1970s, induction motors became even more popular and prevalent in industry [62].

1 Silicon-Controlled Rectifier

Chapter 3: Overvoltages in VFDs 50

A Variable Frequency Drive (VFD) is a solid-state electronic energy conversion system that controls the speed and the torque of an AC motor by controlling the amplitude and the frequency of the voltage that is being supplied to it. Generally, this frequency control is achieved by converting AC voltage to DC by a rectifier bridge converter, and then converting DC voltage back to AC with the desired frequency and voltage, using an inverter. Usually there is a middle section, called the DC link, comprised of one or several capacitors that reduces the DC voltage ripples. Various techniques may be used for DC/AC conversion, such as Pulse Width Modulation (PWM) and Space Vector Modulation (SVM). In these techniques, a desired output voltage is generated as the average value of multiple discrete voltage pulses, generated by means of electronic switches. Variable frequency drives may be categorized by their topology, load torque characteristics, the switching control method, and their output voltage levels. VFDs may be referred to by a variety of names such as Adjustable Speed Drives (ASD) and Variable Speed Drives (VSD). Figure 3.1 shows a typical drive system with PWM technique.

Figure 3-1. Typical motor drive system

VFDs are used in applications ranging from small electric appliances to large mine mill drives. In recent years with the increased attention to power efficiency, VFD usage also has increased dramatically in

HVAC applications for energy saving, soft starting and power factor correction purposes. In variable torque loads such as fans and pumps, VFDs offer a great opportunity for energy saving. Electric motors systems typically account for about 70% of manufacturing electricity consumption in the US. From this enormous amount of energy, about 60% are consumed by fans and pumps [63]. As a general rule for variable torque loads, their power varies with the cube of the speed, e.g., 20% speed reduction would decrease the power

49%.

Chapter 3: Overvoltages in VFDs 51

ퟑ 흎ퟐ 흎ퟐ = ퟎ. ퟖ 흎ퟏ => 푷ퟐ = ( ) 푷ퟏ = ퟎ. ퟓퟏퟐ 푷ퟏ Equation 3-1 흎ퟏ

In a typical HVAC system, a maximum air flow that the fan should be able to provide, or 100% of design air flow, is only needed 2% of the time. In 85% of the time, the required air flow is between 40% and 85% of the maximum air flow [62]. In other words, 98% of the time, energy consumption can be reduced using a VFD to control the speed of the motors. This provides a huge possibility to reduce the operating cost by a reasonable amount of investment. The actual saving and payback period depend on the specifications of the system and the cost of energy. For example, in a hardwood company in New York, with an initial investment of $46,000 to replace the motors in a ON/OFF moisture control system with a

VFD system, the payback period was less than a year and the company’s first year saving exceeded

$40,000 [62]. For other systems the payback period may vary, but the flexibility and advantages of VFDs push all industries to use them in all new applications and also replace most of older inefficient systems.

Alongside all advantages that VFDs have and all new horizons that they open up to engineers, they bring new challenges and new issues to consider, especially when they are retrofitted into existing motor systems for energy saving or speed control. In most cases, the first question might be whether the new drive can be installed using the existing wiring and enclosure. However, in some applications, the PWM voltage may cause other short-term and long-term problems, such as false over-voltage and over-current trips and reduced lifetime of the equipment [64]. Some of these problems are associated with the over-voltages that the switching of power electronic devices creates. We will discuss these issues in detail in this chapter.

Usually inverters use IGBT switches with a typical voltage rise of 600 V in 0.1 µs [65]. This rapid voltage change, alongside the mismatch between the motor and the cable surge impedances, cause high frequency transients in a VFD system. Reflection of traveling voltage wave from motor terminals creates high frequency voltage spikes [7], [66]. Figure 3-2 shows the measured overvoltage oscillations on the motor terminal on the rising and falling edges of one pulse when a 95 feet cable is connecting the inverter and the motor. Pulse amplitude at the output of the inverter is 350 V. The peak of the overvoltages on the motor terminal reaches 630 V, which can create numerous problems for the motor, the cable, the drive and the nearby susceptible electronic devices.

Chapter 3: Overvoltages in VFDs 52

Inverter output voltage

Motor terminal voltage

Figure 3-2. Overvoltages created on the motor terminal by the reflected voltage wave

In Section 3.3, it is mathematically proved that if the rise time of the voltage pulse is less than twice the voltage wave’s travel time between the inverter and the motor, a full reflection will occur, and the motor terminal voltage will reach almost double the voltage of the DC link. Other conditions, such as weak dc link, neutral point displacement and braking of the motor, might contribute to overvoltages and increase the amplitude of the voltage spikes. Overvoltages up to 3 times the voltage of DC link have been reported in the literature [67]. These overvoltages have adverse effects on the motor, the drive, and the electric system, which will be discussed later in this chapter. Approaches to eliminate or reduce the overvoltages will be reviewed in Chapter 4.

Other than negative impacts of the over-voltages on the drive system, high frequency voltage spikes create new challenges for the overvoltage protection devices. The components of the drive system, especially sensitive electronics in the control circuit need to be protected against external surges and transients, as well as, overvoltage spikes generated by the drive. Usually, overvoltage and surge protection devices are designed for occasional events, where they have enough time to cool afterward. When overvoltages are repetitive in nature, over-voltage protecting devices might experience overheating, failure or reduced lifetime. These issues, specifically MOV application as a surge protection device will be explored in Chapter 5.

Chapter 3: Overvoltages in VFDs 53

3.2. Effects of High Frequency Overvoltages

Advancements in electronic switch fabrication have increased switching frequency while reducing the losses. Drives have benefited from these fast switches in the form of higher flexibility and performance. On the downside, the major problems associated with these drives come from their three inherent features: the fast rise time of the pulses (or high dv/dt), the pulse repetition frequency, and the overvoltage (overshoots).

In a VFD, usually the output voltage of the inverter is near square wave shaped. However, the voltage wave reflection phenomenon in a cable can create a high frequency overvoltages at the end of the cable that can surpass twice the DC link voltage. These high frequency voltage spikes, created by the interaction of the inverter and the cable, have some unfavorable effects on the components of the system [68], [69].

3.2.1. Effects on Motor

Insulating tapes and materials used in induction motors are traditionally designed for power frequency,

50 or 60 Hz, sinusoidal voltage. High frequency and the fast rise time of the PWM pulses, elevate electrical and thermal stresses developed in the insulators, which can lead to a premature insulation failure. Other than the high frequency and high dv/dt, the amplitude of the overvoltages also can damage the insulators.

Usually, insulating materials in general purpose induction machines are not dimensioned to resist such high overvoltages that are imposed to motor windings in VFD systems with long cables. Partial discharge (PD) occurrence is pointed out as one of the main damaging factors in motor insulations in VFD systems [70]. A partial discharge is an electrical discharge or spark that bridges a small portion of the insulation between two conducting electrodes where the electric field strength exceeds the breakdown strength of that portion of the insulating material. It is known that partial discharge causes erosion and local heating in insulation materials. These symptoms are frequency dependent, which translates into a higher repetition rate and higher dielectric loss when higher frequency voltages are imposed. The process of deterioration can propagate and increase, until the insulation is no longer able to withstand the electrical stress. Industrial surveys show that the additional stresses generated in the stator winding insulation are one of the leading root causes of 30% to 40% of induction motor failures [71]. Also, partial discharges emit energy as electromagnetic emissions, in the form of radio waves, light and heat, and/or acoustic emissions, in the audible and ultrasonic ranges [72].

Chapter 3: Overvoltages in VFDs 54

Another phenomenon that can significantly reduce the lifetime of an induction machine is the bearing current. Bearing current breaks the insulation in the bearings which consists of thin oil film between stator and the rotor. It can create flashover that erodes and damages the bearing surface and can lead to bearing failures within a short time [73]. The occurrence of bearing currents in an induction motor has been known for decades. The early research mainly concentrates on the bearing voltage generated by the alternating flux linkage of the motor shaft. In an ideal AC motor, driven by AC sine wave power supply, the sum of the three-phase voltage is zero, thus there is no common mode voltage. However, in a real AC machine three stator coils are not exactly alike. Asymmetric flux distribution is the main cause of bearing currents inside the machine. Bearing current in ac power frequency motors has been significantly mitigated by advanced motor design and manufacturing techniques and no longer is a crucial issue in most applications. However, recently the problem is again reported when PWM VFDs were installed [74].

The bearing current is caused by the bearing voltage, a voltage generated between the neutral point and the iron core of the motor, also called common mode (CM) voltage. Unlike a sine voltage system, in a

PWM voltage, the peaks of output CM voltage are high due to the inherent instantaneous imbalance of three phase voltages. This high amplitude CM voltage in motors in a VFD system alongside with a high dv/dt creates large pulses of capacitively coupled leakage current that flow between winding and the iron core of the motor through the bearings. High bearing currents in VFDs increase the electrical erosion and wear of the bearings, finally resulting into premature failure of the bearings [73].

To understand bearing currents better, three different mechanisms can be identified that cause damage to the machine’s bearings [73].

1) The CM voltage amplitude can reach to a level that the voltage on the bearing capacitance exceeds

the threshold voltage of the lubricant. With every breakdown, a short-time current flows through

the dielectric and the stored energy in the bearing capacitances is discharged instantaneously. The

electric arc leads to small fusion craters on the inner and outer bearing race that is called electric

discharge machining (EDM). In practical applications, this form of bearing current is associated

with low-to-medium frequency harmonics of the CM voltage.

Chapter 3: Overvoltages in VFDs 55

2) High frequency ground currents are capacitive currents that are flowing from the motor frame to

rotor ground through the bearings and mainly caused by the CM voltage slope (dv/dt) applied to the

machine terminals. Capacitive bearing currents are dominantly driven by medium-to-high

frequencies of CM voltage.

3) Circulating bearing currents are another capacitive current flowing through the shaft, the two

bearings on the drive end (DE) and the non-drive end (NDE) of the shaft and the stator frame.

Figure 3.2 shows the circulating bearing current mechanism. Both stator and rotor are conductive,

the induced shaft voltage vshaft appears at the two bearings. If vshaft exceeds the dielectric strength of

the insulators in its path, the lubricant in the bearings will be punctured and an alternating high-

frequency current flows through the bearings.

Figure 3-3 Circulating bearing current

3.2.2. Effects on Drive

Due to fast voltage changes of PWM pulses in ac drives, large capacitive currents flow through line to line and line to ground cable capacitances. These current spikes that are superimposed on normal motor currents flow at every rising and falling edge of pulses and depending on the rate of the voltage change and the type of the cable, can have large amplitudes. If the peak value of the current fed by the inverter surpasses its overcurrent protection setting, an overcurrent trip will occur, and the drive will experience unexpected shut downs [69], [75], [76].

Figure 3-4 shows a cable’s capacitive charging current for a 95 feet cable. The end of the cable is not connected to any load and is left open, so all the current seen at the inverter output is cable charging current.

Chapter 3: Overvoltages in VFDs 56

L-L voltage, end of cable

L-L voltage inverter output

L-G voltage, end of cable

Cable charging current

Figure 3-4. Cable charging current, measured for a 95-feet open circuit cable

Furthermore, high repetitive peak currents are flowing through the IGBT switches and contribute to their heating. Excessive heat by pulse currents can lead to the failure of the power module that usually is sized for the normal current of the motor [76]. Even if the power switches can tolerate the current spikes, over-heating and repetitive high currents can reduce their lifetime drastically.

For low-power drives where the bus capacitors are small, the common mode current can pump up the

DC bus voltage and cause an overvoltage trip. The mechanism of feeding power into the dc bus can be explained by looking at the ground current’s path during the stages of switching of inverter’s switches and the conduction periods of the diodes in the rectifier. The average power that flows through the dc bus is nonzero due to the ground current, and can consequently, boost the DC bus level. The boost due to the common mode current is usually small compared to the bus voltage and will reach a stable amount due to power dissipation in the inverter and the cables. However, large charging current through the cable capacitance of long cables can act in the same manner and boost the DC bus voltage. Figure 3-5 shows the cable charging current path through the DC bus. The drive system comprises of a 380 V converter and inverter that feeds 75 parallel 3 kW sewing machines through cables that are approximately 100 m long.

Figure 3-6 shows a 200 V rise in the measured dc bus voltage that causes an overvoltage trip [77]. The DC

Chapter 3: Overvoltages in VFDs 57

bus voltage boost issue is more likely to occur when the drive is attempting to stop the motor or reverse its direction of rotation [76].

Figure 3-5. Circulation of common mode current

Figure 3-6. DC bus voltage rise due to common mode current [77]

3.2.3. Other Effects

As mentioned previously, in a typical inverter-fed motor, the common mode voltage is noticeable. This common mode voltage creates a common mode current that has some adverse effects. The transients create high frequency circulation current into the ground and can trigger upstream differential protection and ground fault relays [78].

Another issue concerning the ac drives are the generated electro-magnetic interferences (EMI) that can disturb the susceptible electronic devices [75], [78]. The presence of common voltage in the ac drives generates high frequency oscillatory ground current in every voltage transition. As switching frequency increases and machine zero-sequence impedances decrease, the common mode voltage causes larger common mode currents, worsening EMI problems [78]. Injection of high frequency current into ground generates radiated emissions as well as noises on the ground wire, disturbing nearby electronic devices.

Chapter 3: Overvoltages in VFDs 58

3.3. Causes and Effective Factors

When the inverter is connected to the motor through a long cable, because of the cable inductance, the stray capacitance distributed between the cable wires and the fast rise times of the pulse width modulation

(PWM) voltage from the inverter, will cause overvoltages appear at the motor terminals. Figure 3.2 shows an equivalent circuit for a transmission line or a cable. Using distributed parameters along the cable, we can derive:

휸풙 −휸풙 휸풙 −ퟐ휸풙 푽(풙) = 푽+ 풆 + 푽− 풆 = 푽+ 풆 (ퟏ + 휞 풆 ) Equation 3-2

푽 푽 푽 푰(풙) = + 풆휸풙 − − 풆−휸풙 = + 풆휸풙(ퟏ − 휞 풆−ퟐ휸풙) Equation 3-3 풁ퟎ 풁ퟎ 풁ퟎ

훾푥 where V and I are the voltage and current at point x, respectively. V+ and V- are constants. 푉+ 푒 is

−훾푥 complex representation of a forward traveling wave or the incident voltage wave. Similarly, 푉− 푒 is backward traveling wave or reflected voltage wave. Propagation coefficient, γ = α + jβ, is a complex number in which the real part, α, is called attenuation coefficient and the complex part, β, is phase change coefficient. Z0 is the characteristic impedance or Surge Impedance of the cable. Γ is the Reflection

Coefficient defined as:

푽 휞 = − Equation 3-4 푽+

Figure 3-7. Equivalent circuit for sections of a transmission line

For a lossless line, which is an acceptable approximation for a real cable, we will have:

푳 풁ퟎ = √ ⁄푪 (Ω) Equation 3-5

휸 = 휷 = 흎√푳푪 (1/m) Equation 3-6

Chapter 3: Overvoltages in VFDs 59

L and C are inductance and capacitance of the cable per unit length. Considering a ZL load at the end of the line and a source with an internal impedance of ZS in the beginning of the line, we can rewrite the reflection coefficients at the sending (훤푆) and receiving (훤퐿) ends of the cable.

풁푳−풁ퟎ 휞푳 = Equation 3-7 풁푳+풁ퟎ

풁푺−풁ퟎ 휞푺 = Equation 3-8 풁푺+풁ퟎ

Reflection coefficient at motor terminal is around 0.9 and for low impedance source is around -1. For an open line, reflection coefficient is 1 and a full reflection would occur. In this case, the voltage at the end of the line would reach 2 per unit [79]. Figure 3-8 explains the voltage wave reflection phenomenon for a voltage pulse with a finite dv/dt on a finite length of cable. Figure 3-8(b) top, shows the incident wave traveling toward the end of the cable after the switch is turned on. When the inverter switch turns on, it takes a time, τ, for the voltage wave to reach the motor. Figure 3-8(a) middle shows the incident wave being reflected on arrival at the receiving end of the line. The incident voltage will be reflected as a positive voltage, with the same amplitude for an open cable, traveling toward beginning of the cable. The reflected wave plus the incident wave will cause the voltage equal to double the voltage pulse amplitude at the terminal of the motor. At time (t1 = τ), voltage amplitude becomes almost twice the amplitude of the incident voltage, because of the reflection phenomenon. After 2τ, the reflected voltage wave will reach the inverter. Its reflection from the sending end of cable, called second incident voltage, has the negative amplitude of 훤푆 times the reflected voltage wave amplitude. Therefore, the inverter output voltage is barely affected by the reflection. At the receiving end of the cable, voltage remains almost at two times the incident voltage until the second incident wave (reflected wave from motor again reflects from inverter) reduces the voltage level to almost 1 per unit. Also, the reflection of the negative second incident is a negative wave with the same amplitude. Thus, simultaneously the reflection of the second incident wave reduces voltage amplitude even more to the minimum at t2. This reflection process repeats several times depending on the reflection coefficients and creates the voltage oscillation at each rising and falling edge of the voltage pulse.

Chapter 3: Overvoltages in VFDs 60

(a) (b) Figure 3-8. (a) Voltage wave reflection in a transmission line (b) Simulated and measured voltage waveforms [80]

We can calculate the frequency of voltage oscillations, f0, based on this explanation and the wave theory. ν is the wave travel speed in the cable. l is the cable length.

흂 = ퟏ⁄ (m/s) Equation 3-9 √푳 푪

ퟏ 풇ퟎ = ⁄ (Hz) Equation 3-10 (ퟒ 풍 √푳 푪 )

The overvoltage level depends on the modulation technique, cable length and type, motor impedance, motor’s mode of operation, and inverter output voltage rise time. Here, we briefly explain two important factors: voltage rise time, and cable length.

3.3.1. Voltage Rise Time

If we consider the theoretical extreme case of infinite slope for voltage rise, once the voltage reaches the end of an open cable, voltage rises to twice the magnitude of the voltage pulse. There is no way to avoid it.

In reality, however, the voltage rise time is finite and takes some time for IGBT switch voltage to rise. For rise times longer than twice the wave traveling time over the cable, the slower the rise time, the smaller the voltage overshoot amplitude.

The voltage amplitude on motor terminals is the sum of forward traveling voltage waves and reflected waves. The first incident and its reflection have positive amplitudes, so sum of them create a voltage larger than the original incident voltage. The second incident, which is the reflection of the reflected wave from

Chapter 3: Overvoltages in VFDs 61

inverter’s terminal, and its reflection have negative amplitudes. Thus, they decrease the amplitude of the resultant voltage. This occurs after the voltage wave propagate the cable 3 times.

The first incident and consequently the first reflection have the largest amplitudes and consequent waves have lower amplitudes. The voltage at motor terminals due to the first two forward traveling waves is as below

ퟎ 풕 < 흉 풕−흉 푽 = {( ) (ퟏ + 휞푳) 푽풅풄 흉 < 풕 < 풕풓 + 흉 Equation 3-11 풓ퟏ 풕풓 (ퟏ + 휞푳)푽풅풄 풕 > 풕풓 + 흉

ퟎ 풕 < ퟑ흉 풕−ퟑ흉 푽 = {− ( ) 휞푺휞푳 (ퟏ + 휞푳) 푽풅풄 ퟑ흉 < 풕 < 풕풓 + ퟑ흉 Equation 3-12 풓ퟐ 풕풓 − 휞푺휞푳 (ퟏ + 휞푳)푽풅풄 풕 > 풕풓 + ퟑ흉

where Vr1 is the sum of the first incident voltage and the first reflected wave. Similarly, Vr2 is the sum of second incident wave (reflected from inverter) and its reflection. Notice that it takes τ seconds for the first incident voltage to reach the motor. Thus, before this time the voltage is still zero. The same reasoning explains 3τ for Vr2. In order to avoid full reflection, rise time, tr, should be long enough to allow the second incident reach the motor before voltage reaches its maximum. This means:

풕풓 + 흉 > ퟑ흉 → 풕풓 > ퟐ흉 Equation 3-13

If the voltage pulse rise time is shorter than twice the traveling time, τ, a full reflection will occur, and motor terminal voltage will reach almost double the voltage of the DC link. At any time, t, after the first reflection, we can calculate the amplitude of the reflected wave as:

풕 푽풓풆풇풍풆풄풕풆풅 = 푽풅풄횪푳 Equation 3-14 풕풓

Calculating the receiving end voltage at time 2τ and substituting 휏 = 푙⁄푣, we can derive:

ퟐ 풍 푽풓 풑풆풂풌 = 푽풅풄 (ퟏ + 휞푳) Equation 3-15 흂 풕풓

The relative overvoltage can be calculated as:

푽풓 풑풆풂풌−푽풅풄 ퟐ 풍 휞푳 푽풐풗 = = Equation 3-16 푽풅풄 흂 풕풓

Chapter 3: Overvoltages in VFDs 62

Equation 3-16 shows the dependence of the overvoltage amplitude to the rise time, as well as the cable length. The smaller the 푡푟, the larger the 푉표푣. The relation between the rise time and the overvoltage amplitude is depicted in the Figure 3-9 based on the Equation 3-16 his figure shows the overvoltage at the end of a 100 ft cable, connected to a motor with the reflection coefficient of 1. A typical wave propagation speed of 500 ft/us is used.

1.2

1.0

0.8

0.6

0.4

0.2

0.0 NormalizedOvervoltage Peak 0 1 2 3 4 Rise Time (us)

Figure 3-9. Overvoltage on a 100-ft cable for different rise times

Motors and cables have a pre-determined level of insulation against overvoltages, given by the manufacturer. For a specific overvoltage capability, we can determine the minimum rise time of the inverter to avoid damages to the motor and cable. For example, if the maximum acceptable overvoltage level is 1.2 pu, for the cable in Figure 3-9 the minimum voltage rise time is 2 µs. Equation 3-16 can be used to calculate the minimum rise time for a given cable and acceptable overvoltage.

ퟐ 풍 휞푳 풕풓 풎풊풏 = Equation 3-17 흂 푽풐풗

3.3.2. Cable Length

Understanding the cable length’s effect on the overvoltages created in an ac drive system is easy revisiting the traveling wave theory. If the cable size is short, by the time the voltage wave rises, voltage wave has propagated the cable back and forth several time. Thus, the voltage at the end of cable is the resultant of several positive and negative incident and reflected waves that nullify the effects of each other.

Chapter 3: Overvoltages in VFDs 63

When the cable is long, the interval between the incident voltages allow the voltage to rise/fall after the subsequent incident negates the voltage change and changes it in the opposite direction.

Equation 3-16 shows a direct proportional relation between the length of the cable and the overvoltages.

It is depicted in Figure 3-10, for an inverter with rise time of 0.2 µs and a typical propagation speed of 500 ft/µs. The reflection coefficient of 1 is used for the receiving end of the cable for simplicity.

1.2

1.0

0.8

0.6

0.4

0.2

0.0

NormalizedOvervoltage Peak 0 20 40 60 80 100 Cable Length (ft)

Figure 3-10. Overvoltage for an example inverter and motor for different cable length

In practice, manufacturers specify a maximum cable length for different cable types based on their inverters. Given the above example, if the maximum acceptable overvoltage is 1.2 pu for the system, the maximum length of the cable should be limited to around 10 ft. It is also seen that for rise time of 0.2 µs, we expect a full reflection for a 50-ft cable.

From Equation 3-17, the maximum length of the cable for a given inverter and the maximum acceptable overvoltage can be calculated.

풕풓 흂 푽풐풗 풍풎풂풙 = Equation 3-18 ퟐ 휞푳

3.4. Voltage waveforms in a PWM VFD system

Typically, motor drive systems control motor speed by generating the required voltage and frequency in the form of repetitive pulses. Depending on the motor and the control strategy, different methods are used to generate voltage pulses. For an induction motor speed control, space vector modulation (SVM) is a

Chapter 3: Overvoltages in VFDs 64

common method to control PWM. In this method, as described previously, to create the reference voltage, two pulses of line to line voltages are created in each switching cycle. Width of pulses depend on what method of pulse generation or what kind of modulation in SVM is used. However, to generate the same rms voltage with the same dc link voltage and same switching frequency, the pattern of pulses would be similar in all of them. Pulses are narrower around 0 and pi and wider around pi/2 and 3pi/2, for a sinusoidal voltage output. Pulses in the second half cycle are mirror of those in the first half cycle across the zero-voltage line.

Pulse amplitudes are the same for all the pulses, neglecting small variations in dc link voltage.

When surge protection devices and MOVs are installed on 3 phase lines, they may be connected between a line and ground, between a line and neutral, or between lines. For a traditional power system, all three voltages have sinusoidal waveform and the line voltage is √3 times the phase voltage. Moreover, for a symmetrical 3-phase system, voltage between the neutral and the ground is zero. That means line to neutral and line to ground voltages are basically the same. However, for a PWM system this is not the case.

Pulses are created by connecting lines to either positive or negative dc bus. Considering a midpoint- grounded dc link, voltage of phases would be either +Vdc/2 or -Vdc/2. Thus, for the phase voltages, pulses would be Vdc high and the maximum voltage would be Vdc/2. Similarly, line to line voltages are either Vdc or -Vdc, depending on what buses two lines are connected to. The pulse height and the maximum voltage, both are Vdc. Despite the traditional systems with sinusoidal voltages, line to neutral voltages in a PWM system are not the same as line to ground voltages. Neutral voltage point is moving to +Vdc/6 or -Vdc/6 depending on the lines’ connections to the dc buses. Figure 3-11 shows a simplified diagram of an inverter feeding a motor, with the grounded DC link midpoint. This is a common configuration in 3 phase ac drives.

Figure 3-11(b) shows the same circuit, during one stage of switching pattern, when S1, S3, and S6 are closed. At this moment, Phases A and B are connected to the positive bus, and phase C is connected to the negative bus. The equivalent impedance of two parallel impedances of phases A and B is half of the phase

C impedance. So, voltage is divided 2 to 1 on the motor impedances. VAN and VBN are Vdc/3 and VCN is -

2Vdc/3. In this situation, neutral point moves Vdc/6 towards positive dc bus voltage. For neutral to ground voltage, we can write:

푽 푽 푽 푽 = 풅풄 − 풅풄 = 풅풄 Equation 3-19 푵푮 ퟐ ퟑ ퟔ

Chapter 3: Overvoltages in VFDs 65

(a) (b) Figure 3-11. (a) Midpoint grounded VFD (b) equivalent circuit when S1, S3 and S6 are closed

When the neutral point of the motor is not grounded, its voltage is equal to the common mode (CM) voltage. CM voltage is the average voltage of the three phases, by the definition. For the Figure 3-11(b) we can write:

푽풅풄 푽풅풄 푽풅풄 푽 +푽 +푽 + − 푽 푽 = 푨 푩 푪 = ퟐ ퟐ ퟐ = 풅풄 Equation 3-20 푪푴 ퟑ ퟑ ퟔ

Figure 3-12 shows a typical drive with a diode rectifier and a 3-phase inverter, simulated in OrCAD.

DC link voltage is 340 V and its midpoint is grounded. Three resistors are connected as a Y-connected load for simplicity to study the line to line, line to neutral and line to ground voltages as well as the CM voltage created on the inverter output. Figure 3-13 shows the CM voltage, which in this case is equal to the voltage between neutral point and the ground. As it can be seen from the figure, the peak of CM voltage is equal to

Vdc/2 and quite noticeable in the drive system.

APT30G100BN

V Z1 Z3 Z5 R1 R3 R5

1.5m C6 1 1 1 PWM1 PHA PWM3 PWM5 AC input V PHB V+ V+ 1.5m PHC V- 0 C8 Z2 Z4 Z6 R2 R4 R6

Rectifier 1 1 1

V PWM2 PWM4 PWM6 RLO1 RLO2 RLO3 100k 100k 100k

VNeg

V- Figure 3-12. VFD system simulated in OrCAD

Chapter 3: Overvoltages in VFDs 66

Figure 3-13. Common-mode voltage in an ac drive

400V

200V

0V

-200V

(a) -400V 0s 5ms 10ms 15ms 20ms 25ms 30ms 35ms V(PHA,PHB) Time

400V

200V

0V

-200V

(b) -400V 0s 5ms 10ms 15ms 20ms 25ms 30ms 35ms V(PHA,RLO3:1) Time

400V

200V

0V

-200V

(c) -400V 0s 5ms 10ms 15ms 20ms 25ms 30ms 35ms V(PHA) Time

Figure 3-14. VFD system voltages (a) line to line (b) line to neutral (c) line to ground

Chapter 3: Overvoltages in VFDs 67

It is seen from Figure 3-14 that line to line voltage pulses have amplitude of 340 V on both positive and negative directions. The maximum voltage for a line to neutral connection is 226 V, while some pulses have amplitudes of 113 V and others have amplitudes of 226 V. For line to neutral voltages, it is seen that all pulses are between -170V and +170V with amplitude of 340 V.

In most VFD commercial products, midpoint is grounded and above-mentioned relations between voltages are true. However, in some VFDs it is possible that the midpoint of the dc link capacitors is not connected to the ground. Such a configuration does not affect line to line and line to neural voltages. But line to ground voltages will be affected. Figure 3-15 shows a VFD system without its midpoint grounded.

This configuration saves one capacitor in the drive, with using one capacitor with higher nominal voltage, instead of two capacitors in series.

On the power system side, in most cases the neutral wire is connected to the ground. In such a system, the voltage of the dc link’s midpoint will vary regarding to the ground, causing the line to ground voltages have higher peak value near the peaks of the input sinusoidal voltage. This elevated line to ground voltage should be considered in installing appliances between lines and the ground. Figure 3-16 shows the line to line, line to neutral and line to ground voltages of the system alongside the voltage of the midpoint of capacitors to the ground. It is seen that the line to line and line to neutral voltages are the same as the waveforms with the dc link’s midpoint grounded. On the contrary, line to ground voltage is different, when the midpoint is not grounded. This is a result of dc link’s midpoint voltage swing regarding to the ground, which is shown in Figure 3-16(d).

Figure 3-15. VFD system without a grounded midpoint

Chapter 3: Overvoltages in VFDs 68

400V

200V

0V

-200V (a) -400V 0s 5ms 10ms 15ms 20ms V(PHA,PHB) Time

400V

200V

0V

-200V (b)

-400V 0s 5ms 10ms 15ms 20ms V(PHA,RLO2:1) Time

400V

200V

0V

-200V (c)

-400V 0s 5ms 10ms 15ms 20ms V(PHA) Time

400V

200V

0V

-200V (d)

-400V 0s 5ms 10ms 15ms 20ms V(C6:2,0) Time

Figure 3-16. VFD system voltages without midpoint grounding (a) line to line (b) line to neutral (c) line to ground (d) midpoint of dc link capacitors

Chapter 3: Overvoltages in VFDs 69

Considering analysis and waveforms above, any change in dc link midpoint voltage will be added to the phase-ground voltage. Figure 3-17 shows a VFD with single phase input that uses a voltage doubler stage to boost up the dc link voltage to a required level. This is a common practice in single phase input drives to create the necessary dc link voltage. In most single-phase systems, the neutral is connected to the ground somewhere in the system. So, any inadvertent input wire swapping for this drive will change the grounding connection and will result in doubling the line to ground maximum voltage. Figure 3-18 shows the line to ground voltage of phase A and DC link’s midpoint voltage. Considering the normal function of the system, line to line and line to neutral voltages will remain unchanged and the inverter and the motor will continue to work. However, at the same time, devices installed between the lines and the ground, such as SPDs, will be exposed to much higher maximum voltage that lead to their failure or faster degradation and reduced lifetime.

Figure 3-17. VFD system with voltage doubler input stage and swapped phase and neutral inputs

400V

200V

0V

-200V

-400V 0s 5ms 10ms 15ms 20ms V(PHA) V(Vs:+,0) Time

Figure 3-18. VFD system voltages with voltage doubler input stage and swapped phase and neutral inputs (red) line to ground (black) midpoint of dc link capacitors

Chapter 3: Overvoltages in VFDs 70

3.5. Simulation of a VFD system for overvoltages

Simulation study is a main part of every design and decision making in today’s industry. The key strength of simulation studies is the ability to investigate the feasibility, functionality and performance of the proposed ideas without incurring significant expenses. Simulations can answer what-if questions without interfering with the system and can prevent costly mistakes. It also allows investigation of the corner cases, where it needs a lot of time, workforce and samples to cover statistically. However, these benefits come with a few downsides. Simulation time and cost is added to the project and the outputs rely heavily on the inputs provided and the models used.

Numerous papers are published on the simulation of the ac motor drive systems with long cables [7],

[8], [81]–[87]. Main focus of the simulations might be different based on the goals and target system.

However, in general these simulations are comprised of modelling cable, modeling the motor, and system simulations. Here, cable and motor models are briefly reviewed first. Then measurements of cable and motor parameters in the lab are presented. Finally, our existing motor and drive are simulated, and the results are compared with the experimental measurements.

3.5.1. Cable model

Traditionally, four parameters of a cable are considered in the line and cable models: series resistance and series inductance that form series impedance, and parallel capacitance and parallel conductance that present the admittance of the line. In reality, these parameters are distributed over the cable length.

However, considering the lumped parameters makes the calculations easier. Figure 3-19 shows the conventional cable model which is used in most general studies. This model does not consider the dependencies of the cable parameters to frequency and consider them to be constant. Distributed model is briefly presented in section 3.3, by equations 3-2 and 3-3.

Chapter 3: Overvoltages in VFDs 71

Figure 3-19. Conventional cable model for short cables and low frequency

Most of the studies use variations of the conventional model [7], [66], [85]. Dividing the cable into n cells and using multiple back to back conventional model for each cell can increase the accuracy of the model. Paper [7] suggests that a 20-cell RLCG model is an acceptable compromise between simulation complexity and the size of the error. Another research recommends 5 cells per meter [82].

Other papers add more elements to the conventional model to reduce its errors for high frequencies.

Two of proposed improvements are shown in Figure 3-20 [87], [88].

(a) [88] (b) [87]

Figure 3-20. Improvements of conventional model of the cable

In the applications where the accuracy of the cable impedance over a frequency range is needed, more accurate models are proposed [82], [84]. Usually, other than three main wires, cables contain one or more ground wires and shielding layers. Considering capacitances among multiple wires and using the models that can cover skin effects and frequency dependency of the parameters are necessary in the modeling accuracy. The model shown in Figure 3-21 through Figure 3-23 considers capacitances between three phases, ground wire and the cable shield. For better accuracy over a large frequency range, RL and RC ladder networks are used to closely match the series and parallel impedances of the cable.

Chapter 3: Overvoltages in VFDs 72

Figure 3-21. Shielded cable model

Figure 3-22. RL ladder network for the series impedance of a cable

Figure 3-23. RC ladder network for the shunt admittance of the cable

3.5.2. Measurement of cable parameters

Generally, calculation of the model parameters is accomplished by measuring common mode, and differential mode impedances and solving the basic equations. The exact measurement method depends on the complexity of the model, number of the wires in the cable, and whether the cable is shielded or not. In the cases where the model is complex, computer programs are used to fit the measured impedance curve to the model’s impedance.

Chapter 3: Overvoltages in VFDs 73

For the commonly used 4 wire cable, the measurements of the parameters are described in the Figure

3-24. Inductance of the wire, L, the mutual inductance between 2 wires, M, and the coupling coefficient, K, can be calculated using the basic circuit theory as below.

(a) Common mode (CM) (b) Differential mode (DM)

Figure 3-24. Common Mode and Differential Mode configurations for cable inductance measurement

(a) Common mode (CM) (b) Differential mode (DM)

Figure 3-25. Inductance measurement for cable in CM and DM mode

푳 푳 ퟑ 푴 푳 = (ퟏ + ퟑ 푲) = + Equation 3-21 푪푴 ퟒ ퟒ ퟒ

푳푫푴 = 푳 (ퟏ − 푲) = 푳 − 푴 Equation 3-22

ퟑ 푳 = 푳 + 푳 Equation 3-23 푪푴 ퟒ 푫푴

ퟏ ퟑ 푲 = (푳 − 푳 ) Equation 3-24 푳 푪푴 ퟒ 푫푴

ퟑ 푴 = 푳 − 푳 Equation 3-25 푪푴 ퟒ 푫푴

Capacitance of the cable is also measured by the measurement of the line to line capacitance, Cc, and line to ground capacitance, Cg, as it is depicted in Figure 3-26. capacitances between wires of a cable and formulated in the following equations.

Chapter 3: Overvoltages in VFDs 74

Figure 3-26. capacitances between wires of a cable

First, measurement is made by connecting 3 phase wires together and measuring the total capacitance to the ground, where it will be equal to three times the line to ground capacitance, as it is sown in Figure

3-27(a). The capacitance between wires are also measured by connecting all the wires and the shield together except one wire. Capacitance between one wire and the rest of the conductors is equal to 2 Cc +

Cg, as it is seen in Figure 3-27(b).

(a) Ca (b) Cb

Figure 3-27. Measurement of capacitance of a cable

푪풂 = ퟑ 푪품 Equation 3-26

푪풃 = ퟐ 푪풄 + 푪품 Equation 3-27

Then, Cc and Cg can be calculated from the above.

ퟏ 푪 = 푪 Equation 3-28 품 ퟑ 풂

ퟏ ퟏ 푪 = 푪 − 푪 Equation 3-29 풄 ퟐ 풃 ퟔ 풂

Chapter 3: Overvoltages in VFDs 75

3.5.3. Motor Models

The effects of the high frequency voltage components introduced by the PWM technique are usually neglected when the electromechanical performance of the motor is analyzed. These low frequency behaviors of the motor can be analyzed by d-q model or conventional T-model equivalent circuit, shown in

Figure 3-28,. On the contrary, motor’s common mode and cable’s charging currents are noticeable under

PWM voltage and cannot be neglected. Overvoltages created by the voltage wave reflection cannot be simulated with acceptable precision with the conventional induction motor model. For these applications, high frequency models of the motor have been proposed [81]–[83], [86].

Figure 3-28. Conventional equivalent circuit of induction motor

In order to achieve a high accuracy in high frequency modeling, reference [82] proposes a model with a per phase equivalent circuit comprised of ten capacitor, ten inductance and ten resistor. The values of these elements are calculated by the computer programs by the means of curve fitting of the circuit impedance to the measured impedance over a wide frequency range. The proposed equivalent circuit per phase is shown in Figure 3-29.

Figure 3-29. Per phase equivalent circuit of motor [82]

Chapter 3: Overvoltages in VFDs 76

Other researchers [81], [83], [86] have proposed simpler models for high frequency model of the induction motor. Although they may vary in the configuration of the elements, they all consider the system’s capacitances to the ground, since in high frequencies the leakage current can be noticeable. A few of proposed models are shown in Figure 3-30.

(a) [81] (b) [83]

(c) [7] (d) [86]

Figure 3-30. High frequency models of induction motor

There are two main capacitances to consider in the high frequency model of an induction motor: winding to ground capacitance and winding turn to turn capacitance. In the above models, the capacitance between w and n represents the winding turn to turn capacitance. The winding to ground distributed capacitance has been represented by two lumped capacitances. The first one is connected between the phase terminal, w, and the ground, g. The second one is connected between the motor neutral, n, and the ground, g. Considering equal values for these two capacitances are favored in most models. The phase resistance (R or Rt) and the phase inductance (Ld or Lt or LM) are the low frequency parameters obtained by the locked rotor test. The core and eddy losses are represented by a resistor between w and n (Re or R2). To increase the accuracy of the high frequency damping, some models consider one or two resistors in the ground path, in series to the ground capacitors.

Chapter 3: Overvoltages in VFDs 77

High frequency equivalent circuit provides an acceptable model for the motor for voltage wave reflection simulation and study of overvoltages. However, it cannot be used to study motor’s power system frequency behavior, where stablished d-q models or symmetric equivalent are used. To resolve this limitation, some papers have proposed to include both low frequency and high frequency models as a universal model for cases that need a wide frequency range [7], [81], [86]. Two of these proposed models are shown in Figure 3-31. One major drawback of combining low frequency and high frequency models is the long simulation time. In order to have accurate results for a high frequency phenomenon it is necessary to use a small timestep. At the same time, to observe the low frequency behavior it is necessary to cover enough time span. Simulating such a long time with such a fine step, increases of the computation time.

Figure 3-31. Universal (high and low freq) model of an induction motor [81], [86]

3.5.4. Measurement of motor parameters

Generally, parameters of the high frequency model of the motor are calculated using a few measurements of common mode and differential mode impedances. Sometimes for simplicity, these measurements are made in a specific frequency in which the impedances and errors are acceptable for the purpose of the simulation. Alternatively, the parameters can be measured over a range of frequencies and the calculated impedance curve is fitted to the measured impedance. Taking the model in Figure 3-30(a) into the account, the measurements basics and equations are explained below.

Chapter 3: Overvoltages in VFDs 78

Figure 3-32. Motor impedance measurement

(a) Zwn (b) Zwg

Figure 3-33. motor impedances (a) winding to neutral (b) winding to ground [81]

Impedances are measured for these two configurations. The impedance Zwn is measured between the three phase terminals connected together and the motor neutral, with the floating ground terminal. The impedance Zwg is measured between the three phase terminals connected together and the ground terminal, with the floating motor neutral. Calculating these impedances from Figure 3-30(a) with the consideration that in high frequencies R will be negligible compared to Ld, delivers the below equations.

풔푳풅 풁 = 푪 Equation 3-30 풘풏 ퟐ 품 푳풅 ퟑ [풔 푳풅(푪풕+ )+풔 +ퟏ] ퟐ 푹풆

ퟐ 푳풅 풔 푳풅(푪풕+푪품)+풔 +ퟏ 푹풆 풁 = 푪 Equation 3-31 풘품 ퟐ 품 푳풅 ퟔ퐬푪품 [풔 푳풅(푪풕+ )+풔 +ퟏ] ퟐ 푹풆

The impedance Zwn has a zero in the origin and a pair of poles with natural frequency of fpwn.

ퟏ 풇 = Equation 3-32 풑풘풏 푪품 ퟐ흅√푳 (푪 + ) 풅 풕 ퟐ

Chapter 3: Overvoltages in VFDs 79

The impedance Zwg has two complex conjugate zeros at frequency fzwg, as well as one pole at origin and

two complex conjugate poles at fzwn that is equal to fpwn.

ퟏ 풇풛풘품 = Equation 3-33 ퟐ흅√푳풅(푪풕+푪품)

풇풛풘풏 = 풇풑풘풏 Equation 3-34

By matching the above equation to the impedance measurements shown in Figure 3-33, parameters of

the model are calculated. Cg and Ct can be calculated using the fact that at both low and frequencies the

impedance Zwg is almost purely capacitive.

ퟏ 푪품 = @ (ퟏ ~ ퟏퟎ 풌푯풛) Equation 3-35 ퟔ(ퟐ흅풇)풁풘품

Calculation of Ct shows that it is very small and can be neglected. Re can be calculated using the Zwn at the resonance frequency.

푹풆 = ퟑ 풁풘풏 @ 풇풑풘풏 Equation 3-36

Ld and R (neglected here) are obtained by the commonly used locked rotor test. This value, however, might be a little different from the high frequency inductance. For more accurate value we can use

Equation 3-33 to derive the inductance.

ퟏ 푳풅 = ퟐ Equation 3-37 (ퟐ흅풇풛풘품) 푪품

3.5.5. Experimental setup and Measurements

In order to achieve the goals of the research, an experimental setup is built in the lab in Northeastern

University. This setup is comprised of an induction motor controlled by a VFD. Motor shaft is coupled to a dynamometer as a mechanical load. As the controller, two different drives are used in different tests: one commercial drive, and one programmable motor control development board. Figure 3-34 shows the experimental setup.

Chapter 3: Overvoltages in VFDs 80

Figure 3-34. Experimental set up in Northeastern university lab

The commercial drive is WEG CFW10, a 1 hp motor controller, with a 120 V single phase input. Being a commercial product, it does not allow much control over parameters such as dc link voltage. Switching frequency is adjustable from 2.5 kHz to 15 kHz. A voltage doubler at the input creates a DC link voltage of

340 V which is not controllable. The devkit board, made by Microchip, uses a dsPIC33 EP256MC506

Microcontroller. In addition to the motor speed, switching frequency and the DC link voltage are controllable in this drive.

The cable used in this study, Belden, is a shielded 4 conductor cable specially built for VFD. The specifications of the cable are provided in the below table.

Table 3-1. Specifications of the cables used in the research Voltage Cable AWG Conductors Shield and coverage rating (V) Aluminum Foil-Polyester Tape 100% BELDEN 14 4 600 Multicore VFD Braid TC - Tinned Copper 85%

The motor used in the study is a 230 V, 60 Hz, 1760 rpm, 3-phase, 1 hp general purpose motor. The specifications of the motor are seen in Table 3-2. The shaft of the motor is connected to the shaft of a dynamometer that can provide a mechanical load for the motor.

Chapter 3: Overvoltages in VFDs 81

Table 3-2. Technical specifications of the 3-phase motor Power 1 hp Nameplate Speed 1760 rpm Voltage 230 V Full Load Amps 2.8 A Nominal Efficiency 85.5 %

Figure 3-35. Motor and cable parameter measurement with the precision LCR meter

Impedances of the cable and the motor is measured in the lab, using a Keysight Precision LCR Meter

(model: E4980A/AL). The test frequency range for this meter is 20 Hz to 1 MHz, with an accuracy of

0.01%. Figure 3-35. Motor and cable parameter measurement with the precision LCR metershows the impedance measurement of the motor and the cable with the precision LCR meter in the lab. As it is seen in the pictures, coaxial cables and BNC connectors are used to reduce the impact of unwanted electromagnetic noises. The meter’s software compensates for the measurement cable’s inductance and capacitance, using the short circuit and open circuit measurements.

3.5.5.1. Cable parameters

The Belden cable’s common mode and differential mode impedances are measured as it is described in section 3.5.2. The measurement results are seen in Figure 3-36. The self and mutual inductances can be calculated using equations 3-23 and 3-25. For example, for inductances in 1 MHz, we have:

L = 618.4 + 3/4 × 349.0 = 880.2 nH/m

M = 618.4 – 1/4 × 349.0 = 531.2 nH/m

Chapter 3: Overvoltages in VFDs 82

2000

1500

1000

(nH/m) CM

L 500

0 10 100 1K 10K 100K 1M Frequency (Hz)

1200 1000 800 600 400

LDM (nH/m) LDM 200 0 10 100 1K 10K 100K 1M Frequency (Hz)

Figure 3-36. Cable inductance measurement for Belden Cable (top) LCM (bottom) LDM

Capacitance measurements are also done for the Belden cable as described before and line to line capacitors and line to ground capacitors are calculated using equations 3-28 and 3-29. As it is seen in

Figure 3-37, measurements show that capacitances for the cable are almost constant in the range of measurement.

140 120 100 80 60 Cg 40 Cc 20 Capacitance(pF/m) 0 10 100 1K 10K 100K 1M

Frequency (Hz)

Figure 3-37. Line to line (Cc) and Line to ground capacitance of Belden Cable

Chapter 3: Overvoltages in VFDs 83

Series resistance and parallel conductance of the cable are also measured during the inductance and capacitance measurements. Parallel conductance can be measured by connecting the three lines wires and measuring the conductance to the ground. The wire series resistance will be three times the measured resistance, when 3 lines are connected in parallel. The measured parallel resistance of the cable is seen in

Figure 3-38. Parallel resistance of the Belden cable. The series resistance of the wire can be measured when measuring the common inductance. It is shown in Figure 3-39 for the Belden cable.

6000 m)

Ω. 5000 4000 3000 2000 1000

Parallel Parallel Resistance (M 0 10 100 1K 10K 100K 1M Frequency (Hz)

Figure 3-38. Parallel resistance of the Belden cable

250

200

/m) 150 Ω 100

50 Resistance(m 0 10 100 1K 10K 100K 1M Frequency (Hz)

Figure 3-39. Series resistance of the Belden cable

Although the impedance of connecting wires is accounted for by the precision LCR meter by measuring the short circuit and open circuit impedances, still noises and parasitic capacitances and inductances

Chapter 3: Overvoltages in VFDs 84

interfere with the measurements. To reduce the measurement error, the parameters are measured several times and average values are used. The cable parameters are seen in the below table for the Belden cable.

Table 3-3. Measured cable parameters for Belden cable Freq Hz 1 M 60 L nH/m 350 957 Rs mΩ/m 196 10.2 C pF/m 130 137 Rp KΩ.m 3.5 299,000 Zc Ω 51.9 83.5 τ ns 196 332

3.5.5.2. Motor parameters

For measurement of the motor parameters, the method described in 3.5.4 is used. In this method Zwn and

Zwg of the motor are measured by a precision LCR meter for frequency range of 20 Hz to 1 MHz. The measured impedances are seen in Figure 3-40 and Figure 3-41.

10,000

1,000

) Ω

100 Zwn ( Zwn 10

1 10 100 1K 10K 100K 1M Frequency (Hz)

90 45 0

10 100 1K 10K 100K 1M (Degrees)

Zwn Angle Angle Zwn -45 -90

Figure 3-40. Zwn measurement for the motor

Chapter 3: Overvoltages in VFDs 85

1.0E+07 1.0E+06

) 1.0E+05 Ω 1.0E+04

Zwg( 1.0E+03 1.0E+02 1.0E+01 10 100 1K 10K 100K 1M Frequency (Hz)

0 10 100 1K 10K 100K 1M -45

-90

(degrees) Zwg Angle ZwgAngle -135

Figure 3-41. Zwg measurement for the motor

Measurement results can be used to calculate the parameters of the motor model. Poll and zero frequencies for Zwn and Zwg are determined from the above figures.

fpwn = 145 KHz, fzwg = 105 KHz, fpwg = 180 KHz Equation 3-38

From equations 3-35 through 3-37 we can write:

Re = 3 × Zwn @ 145KHz = 9.6 KΩ Equation 3-39

ퟏ 푪 = = ퟎ. ퟑퟑ 풏푭 Equation 3-40 품 ퟔ (ퟐ흅 ×ퟏퟎ×ퟏퟎퟑ ×ퟏퟎퟎퟎ)

ퟏ 푳풅 = ퟐ = ퟕ. ퟑ 풎푯 Equation 3-41 (ퟐ흅×ퟏퟎퟓ×ퟏퟎퟑ) ×ퟎ.ퟑퟑ ×ퟏퟎ−ퟗ

R in the model, in Figure 3-42, is the high frequency frame resistance to the ground. It could not be derived from the measurements due to the limited frequency range, hence considered 100 ohms. In summary, the model of the motor can be considered as below:

Chapter 3: Overvoltages in VFDs 86

Figure 3-42. Motor model parameters extracted from measurements

3.5.6. Simulation results

The cable, the motor and the inverter are simulated in Matlab/Simulink. First, only an open circuit cable is simulated with distributed parameters. The oscillations of voltage at the motor terminals due to one pulse is compared to the measured waveforms from the lab setup. Then, the motor model is added to the simulations for completeness.

As it is seen from the captured waveforms in Figure 3-43, overvoltages are created both on the rising and falling edges. These waveforms are captured for a dc link voltage of 340 V, but the pulse from inverter output has small oscillations and its peak reaches 350 V. Using a zoom view at the rising edge allows measuring the rising time approximately. The curvy waveform of the pulse makes the determination of a precise rise time for simulations difficult. The voltage rises rapidly at first, but then the slope of the voltage waveform becomes small. The pulse’s rise time is around 150 to 200 ns until it reaches to around 300 volts and then slowly increases to 340 volts. The period of the oscillations are around 780 to 800 ns, which translates approximately to 1.25 MHz oscillations. The time between the pulses at the inverter output and motor terminal is about 200 ns. The peak value of the voltage on the end of the cable reaches 620 to 640 volts, varying for different pulses.

Chapter 3: Overvoltages in VFDs 87

Inverter output Inverter output

Motor terminal Motor terminal

Figure 3-43. Captured voltages on the experimental setup, open cable

For simplicity, in these simulations, the inverter is considered as a controlled voltage source that generates an output waveform similar to the measured voltage pulse on the inverter output.

Figure 3-44. Cable simulation with distributed petameters in Matlab Simulink

A pulse with 150 ns rising and falling time is created in the simulation. Pulse reaches 300 V in 150 ns and then with a smaller slope rises to 350 V by 400 ns after the starting point. More detailed pulse can create a better waveform on motor terminals that is closer to the captured one. However, for the simplicity, this pulse can demonstrate the concept.

Figure 3-45. Voltage oscillations on rising and falling edge of a pulse

Chapter 3: Overvoltages in VFDs 88

Figure 3-46. Voltage oscillations on rising edge of a pulse

Simulations with distributed parameters, using the measured cable parameters shows a 644 V peak value for the voltage peak and a period of 794.5 ns period of oscillations. Although, we should take into considerations that the peak value is somewhat dependent on the shape of the pulse, in addition to the rise time. The period of oscillation is solely dependent on the cable parameters which determines the speed of traveling wave in the cable. Here, it takes about 200 ns (190 to 210 ns in the measured waveforms, 198.6 ns in the simulation) to travel across a 29 m cable. A simple calculation gives us the wave speed of 29/0.1986

µs that is 146 m/µs. Calculation of theoretical wave speed, according to Equation 3-9, also shows:

ퟏ ퟏ 흂 = = = ퟏퟒퟖ. ퟐퟓ × ퟏퟎퟔ Equation 3-42 √푳푪 √ퟑퟓퟎ×ퟏퟎ−ퟗ×ퟏퟑퟎ×ퟏퟎퟏퟐ

Adding the motor model to the simulation, as it is seen in Figure 3-47, changes the waveforms slightly.

However, since the motor impedance is large, the overall changes are not substantial. In fact, some researchers considered motor as an open circuit in high frequency simulations [80].

Figure 3-47. VFD system simulation in Matlab Simulink

Chapter 3: Overvoltages in VFDs 89

During the measurements, motor is running in no-load condition. A no-load motor draws a low frequency sinusoidal current from the inverter. However, in the simulation of the high frequency voltage oscillations, only a high frequency model of the motor is used, and the low frequency current is zero. Still the high frequency current can be compared to the captured waveform, because the low frequency current appears as a dc offset in the captured waveform.

Vab Vag

Vbg

Ia

Figure 3-48. Captured waveforms on motor terminal

Figure 3-49. Simulated waveforms on motor terminal

Chapter 3: Overvoltages in VFDs 90

Figure 3-48 shows the captured waveforms of line and phase voltages of the motor in the experimental

VFD system, as well as motor’s current. When the cable is connected to the motor, for a 340 V pulse, the peak value of line to line voltage at the motor terminal reaches 610 to 620 volts. The oscillation period is

200 to 210 ns. Line to ground voltage is varying between 550 to 570 volts for the captured pulses. The waveform of the line to ground voltage is also different than the line to line voltage, as the voltage of the other phases do not remain at the dc link level because of the induced voltage. In Figure 3-49 the green waveform is the motor current with its low frequency value of 1.6 A at the time of the capture. The high frequency component of the current is super-imposed on the low frequency current and its peak reaches 1.3 amps for the first oscillation in the pulse. This value for the simulation is 1.1 amps. The high error value in capacitive current measurement might be related in part to the capacitance between the length of cable after the current probe and the motor frame. It might also be a result of not considering all the parasitic capacitances of the motor.

Figure 3-49 shows the same waveforms from the simulation. The peak value of the line to line voltage is 619.6 V, where the peak value of line to ground voltage for phase a is 582.5 V. The peak value of the induced voltage reaches 182 volts in the simulation, where its measured value was 180 volts. The main variables are summarized in Table 3-4.

Table 3-4. Comparison of simulation results to measured values

Measurement Simulation Error (%)

VL-L peak (V) 610 619.6 1.5

VL-G peak (V) 200 204.7 2.3 τ (ns) 570 582.5 2.1

iMotor (A) 1.3 1.1 -15.3

When designing a system, or analyzing the system for a particular issue, simulation of the system can prove useful. Simulation can make impossible scenarios to be studied or possible investigation easier. For example, calculating the voltage peak vs. cable length is possible to be done on a real system. However, it needs multiple cables with various length and cables are costly. Moreover, calculating the voltage peak vs. the voltage rise time might be even more difficult in practice, while in the simulation it can be easily done.

Chapter 3: Overvoltages in VFDs 91

Simulation studies also provide important details that otherwise usually are neglected in simplified calculations and approximations. For example, relation between the maximum overvoltage and the rise time of the pulse can be oversimplified neglecting the effect of the pulse waveform and assuming a linearly rising voltage pulse. Moreover, motor impedance in the end of the cable also is not an ideal pure resistive impedance or open circuit to reflect the same waveshape as the incident voltage wave. Calculating the overvoltage using Equation 3-16 leads to monotonically decreasing overvoltage values as the rise time increases. It can be seen in Figure 3-9. However, the waveform of the pulse can change this trend.

Overvoltage value can increase for certain values of rise time, if rise time increases. Figure 3-50 shows the simulation results for the peak value of the voltage on motor terminal versus pulse rise time. As it is seen, if the rise time is less than 2τ a full reflection occurs. This was expected as it is explained previously.

However, for the rise time values larger than 2τ (0.415 to 0.83 µs), the overvoltage value is less than the calculated one. This is because of the pulse and reflection waveforms have larger slope in the beginning and smaller slope near the max value. When the second reflection arrives at the end of cable, it decreases more than what the last part of the incident wave increases. This leads to a lower overvoltage on motor terminal.

Another important result is that, not always increasing the rise time can lead to the decreased overvoltage. In this case, if the rise time is about 0.9 µs and we increase it to 1.3 µs, not only we do not decrease the overvoltage, but also, we increase from about 10% to 20%.

1

0.8

0.6

0.4

0.2 NormalizedOvervoltage 0 0 0.5 1 1.5 2 2.5 3 3.5 rise time (µs)

Figure 3-50. Simulation results: motor's voltage peak vs. pulse rise time

Chapter 3: Overvoltages in VFDs 92

3.6. Summary and conclusion

In this chapter, the theory of the overvoltages in the VFD systems and their adverse effects on the components of the system are presented. Also, the effective factors on the amplitude of the overvoltages are analyzed through theoretical and simulation studies. It is shown that overvoltage values are mainly dependent on the pulse rise time and the cable length. In summary, if the voltage pulse rise time is shorter than twice the voltage wave’s traveling time for the cable, τ, a full reflection will occur, and motor terminal voltage will reach almost double the voltage of the DC link. Furthermore, in this chapter, cable’s and motor’s high frequency models are discussed, and the measurement techniques of the model parameters are presented. High frequency models are extracted for the VFD system that is built in the lab in Northeastern

University and the system is simulated. Simulation results are shown to match the captured waveforms and measured values. It is also revealed that over-simplification of some formulas might lead us to wrong conclusions, where simulation results can provide a detailed analysis with acceptable accuracy.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 93

Chapter 4:

4. Overvoltage Mitigation and Protection in VFDs

In this chapter, surge protection requirements of the ac drive systems and the locations of SPDs in a

VFD system are discussed. Furthermore, regarding the high frequency overvoltages created by the drive that are discussed in the previous chapter, here, a literature review on the existing approaches to mitigation of those overvoltages and protection against them in VFD systems are presented. Advantages and disadvantages of those techniques are discussed.

4.1. Surge Protection in VFDs

As any other electric system, VFD systems need to be protected against surges and overvoltage transients. These surges can come from the utility side or be generated by the drive itself. Typically, the surges that come from the power system are less frequent and have higher energy and amplitude. These surges could be lightning surges, or the switching surges from the power system. In addition to those surges, operation of the converter/inverter also can create overvoltages that can be detrimental to the sensitive electronic circuits. An effective surge protection of a drive system should protect the power electronic switches and control circuit as well as the motor.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 94

In a typical drive system, there are five points to place surge protection devices, shown in Figure 4-1.

The SPDs used in these locations might use protective devices with different technologies. In the commercial products, some SPDs might be integrated with other products, such as filters to provide protection against poor power quality or high harmonic distortion. In these cases, installation of those hybrid products need extra considerations. From surge protection viewpoint, each location in Figure 4-1 is discussed below.

Figure 4-1. One-line diagram of a typical drive system and its surge protection

Location 1 - Drive Input: The first and the most important point for protecting the drive system is at the drive input. Surges from utility can have large energies and/or amplitudes and can damage the rectifier diodes and inverter switches and even affect inverter output [89], [90]. An SPD at this point provides protection from events propagated on the electrical system from upstream sources and external events such as lightning and switching surges created by the utility. An IEEE application note recommends parallel

SPD with no filtering at this point in order to avoid any interaction and interference with drive’s capacitive components [91].

Location 2 - Control Circuit: The control circuit contains sensitive electronics and can be damaged by surges from external sources or by the transients created by the drive. In most cases the input voltage of the drive is 480 V, and the control circuit is isolated by a step-down transformer. It is recommended that a two port SPD with filtering to be installed at location 2 in order to provide low protection level, needed by the electronic circuit [91]. In low voltage drives, usually the control circuit and the power circuit and switches

Chapter 4: Overvoltage Mitigation and Protection in VFDs 95

are integrated into one board. In this case, the protection of the control circuit is recommended by SPDs on the board.

Locations 3 and 4 - Drive Output and Motor Input: Protecting the immediate drive output is recommended, when the length of the connecting cable between the drive and the motor is longer than 10 m (30 ft) or if the connection is routed along an external wall or outdoors. An SPD at this point diminishes the effects of direct lightning or induced voltage surges due to nearby lightning. These surges can cause damage to the drive, even if protection is provided at the motor input. Protection of the motor input is essential, too. Providing protection at this location helps extend the life of the motor and prevent damage to the windings and bearings of the motor due to surges. At these locations, a parallel connected SPD is appropriate [91].

One important issue at locations 3 and 4 arises from the fact that majority of inverters use Pulse Width

Modulation (PWM) techniques to control the frequency and the rms value of the voltage. Thus, the maximum of the waveform is equal to the DC link voltage. This maximum voltage is normally higher than the peak value of a sine waveform with the same rms value (푉푝푒푎푘 = √2푉푟푚푠). When the maximum voltage is considered to select a proper SPD, the protection level would be higher compared to a motor without a

VFD. For example, consider a 208 V three phase induction motor, fed with sinusoidal voltage. Line to Line voltage has the maximum of 294 V. Line to ground rms voltage is 120 V, with the peak value of 170 V.

Line to neutral voltage is equal to line to ground voltage. For a proper MOV selection, the maximum voltage at the point of installation is considered. Usually MOVs are marketed with a limited number of rated MCOVs that engineers select from. For example, one can choose to install 150V MOVs between lines and ground, and lines and neutral, and 275 V or 320 V MOVs between lines, depending on the study of overvoltages at that point. This is a common and well-stablished practice and all MOVs are designed and tested for those conditions. However, in an electric motor equipped with a VFD speed controller, peak values of line to line, line to neutral and line to ground voltages are dependent on the DC link voltage. For example, a drive that controls a 208V three phase motor is likely to have a 340 V dc link. In this case, the maximum line to line voltage would be 340 V, while the line to neutral and the line to ground voltages would have the peak values of 226 and 170 volts. It forces the engineer to select a higher rated MCOV, such as 420 V for line to line protection. Choosing a higher MCOV, seems as a reasonable choice because

Chapter 4: Overvoltage Mitigation and Protection in VFDs 96

it allows the system to function. However, it also increases the voltage levels that the protected system/appliance is exposed to. For example, for a 5-kA surge that is not rare in a distribution system, the clamped voltage of 420 V MOV is 400 V higher than that for 320 V MOV [58]. Even if the appliance can tolerate these higher overvoltages, it can dramatically reduce its lifetime.

Another difficulty in the surge protection for the locations 3 and 4 in the Figure 4-1 is the lack of application notes and installation guidelines. Typical datasheets of MOV manufacturers do not consider installations on PWM lines. Almost no documents are available that describe the selection criteria of

MOVs for such lines or that describe the long-term effects of PWM pulses on MOV’s reliability and lifetime expectancy. Further, typical manufacturer post production and quality check tests are based on the traditional power system sinusoidal waveforms.

Another important and troublesome issue in installation of MOV-based SPDs on motor terminal is the existence of high frequency voltage reflection overvoltages, especially when the connecting cable is long.

As discussed before, when there is no limiting device, these voltage spikes can reach the amplitudes of twice the DC link voltage. MOVs installed on the motor terminal conduct high currents during these peaks, which can result in MOV failure or an extreme reduction in MOV’s lifetime. A simple solution that engineers might use is to install an MOV with higher MCOV in order to avoid high leakage current.

Similar to what is mentioned above, this would increase the protection level which might not be acceptable, depending on the motor insulation level.

Location 5 - Series SPD at Drive Input: At location 5, a series surge protector can add extra security for the drive’s surge protection. As mentioned above, typically a parallel surge protection is needed at drive input. Moreover, a line reactor can protect the drive against lightning strikes and voltage spikes.

Reactors may be installed for use on either the drive input or drive output for different purposes such as to balance the ac lines, or to limit the short circuit current. When installed on the dive input, reactors provide a level of protection from the incoming surges from the utility side. When reactors are placed on the drive output, they help reduce the voltage fast rise time (dv/dt) and the high peak voltages which can be experienced in IGBT inverter applications when the distance between the inverter and the motor is long.

The degree of effectiveness of the reactor depends mostly on its impedance. The most common selections

Chapter 4: Overvoltage Mitigation and Protection in VFDs 97

for this are 3% and 5% impedance reactors [92]. Figure 4-2 shows the effect of an incoming voltage spike from AC power lines on DC bus voltage. Voltage spikes on the AC power lines cause elevation of the DC bus voltage which may cause the inverter to trip for protection. Reactors can absorb these line spikes and offer protection to the rectifiers and DC bus capacitors while minimizing unnecessary tripping of the inverter.

Line reactors have several side effects that need consideration. The main side effect is the voltage drop on the reactor and line loss due to the coil’s resistance. Thus, the use of a reactor is dependent on whether a

DC link choke is built into the drive.

Figure 4-2. Effect of a voltage spike from AC power line on DC bus voltage Mitigation of Transient Overvoltage in VFDs [92]

4.2. Overvoltage Mitigation in VFDs

The phenomenon of voltage wave reflection at motor terminals in a PWM AC drive creates large high frequency overvoltages and imposes severe stresses on the motor insulation. If no mitigation measures are taken, the drive system will face higher damage possibility, longer down time, reduced reliability and shortened lifetime. When the drive/motor system is small and low-cost, installing extra devices for mitigation of these overvoltages might not be economical. In these conditions, depending on the economic assessments, installing a motor and a cable with a higher insulation strength might be an acceptable decision [93]. For larger drive systems or when other engineering design constraints must be met, a

Chapter 4: Overvoltage Mitigation and Protection in VFDs 98

mitigation technique might be used, based on the severity of the overvoltages and financial assessments.

Several approaches have been proposed in literature to solve the reflected overvoltage problem in motor drive systems. Here we summarize the benefits and shortcomings of each approach.

4.2.1. Passive Filters

Passive filters are the most common choice to mitigate the overvoltages on the motor terminal. Passive filters are used either on motor terminals, inverter output or both. The design goal for the filters on motor terminals is to match the impedance of the load with the surge impedance of the cable [94]. Matching surge impedance eliminates the reflection of the voltage at the motor terminal and theoretically reduce the overvoltages to zero. Similarly, the philosophy behind using the filters on inverter output is to increase the rise time of the pulses [95]. A slower rise time provides enough time for consecutive reflections to arrive and reduce the effect of the first reflection. Hence slow-rising pulses create smaller overvoltages. Both approaches require installation of extra components on the line. Apart from the cost and the need for a space to be installed, a filter adds to the losses of the system and/or causes a voltage drop. The ultimate goal is to make an agreement among the conflicting goals: minimal overvoltage, minimal filter losses, minimal common mode currents, and better voltage distribution in the machine stator winding.

The major disadvantage of using a passive filter in general is the higher power loss in the system. When a filter is installed in parallel with the load, the filter current flows through the filter resistors and dissipates power. Series reactors also might be installed on the lines to decrease the overvoltages. Although these reactors are typically low-resistant coils, they carry the load current. They can have a high copper loss in the windings and a high core loss due to the high-frequency pulses. They also cause a voltage drop in the line that is typically 3-5%. Other disadvantages of installing a filter for overvoltage mitigation is the added cost, and the need for a bigger space. Additionally, each filter has its own advantages and disadvantages that are discussed below [86], [93–104].

Chapter 4: Overvoltage Mitigation and Protection in VFDs 99

a. Filters on Motor Terminals

a.1. RC: low voltage overshoot, high dv/dt, high power loss

An RC filter on motor terminal matches the surge impedance of the cable and damps the voltage overshoot. In its simplest form, it consists of Y-connected resistors and capacitors in series. Figure 4-3 shows an RC filter installed in a VFD system. RC filter on the motor terminal works like a snubber circuit.

For high-frequency components of the voltage pulse, capacitor acts as a short circuit, drawing current through it. The energy dissipates in the resistor of the RC filter, hence damping the oscillations.

Although the RC filter looks simple and effective, there are difficulties in its design and application.

The performance of the filter depends on the cable type and characteristics. In addition, in some applications, it is difficult, if not impossible, to access the motor terminals [100], due to the location of the installation or other concerns. Furthermore, depending on the power loss of the filter and the installation conditions, the cooling of the motor can be compromised if the filter installed close to the motor and in the same location.

Figure 4-3. RC filter on motor terminal [86], [94]

Theoretically, if the impedance at the end of the cable is equal to the surge impedance of the cable, the voltage reflection would be zero. Filter parameters are selected so that the filter impedance matches the surge impedance of the cable at the frequency of the oscillations. Parameters of the first-order RC filter can be calculated as below.

ퟐ ퟏ 푳 풁풇 = 풁풄 → √푹풇 + ퟐ ퟐ = √ Equation 4-1 풋흎풇 푪풇 푪

Chapter 4: Overvoltage Mitigation and Protection in VFDs 100

Where Rf and Cf are the filter resistance and capacitance, respectively. L and C are inductance and capacitance of the cable per unit length. One way to determine the filter parameters is to choose Rf resistance to provide over-damping in the circuit [103]. So, we can write:

푳 푹풇 > ퟐ√ Equation 4-2 푪풇

L is the lumped inductance of the cable per unit.

a.2. RLC: low voltage overshoot, high power loss, low dv/dt

Similar to above, as depicted in Figure 4-4, RLC filter at motor terminal can be designed to match the surge impedance of the cable. Adding inductors to the filter can make its response more customizable but adds to the cost and the space needed for the installation. Different approaches of calculating filter parameters are proposed to achieve low overshoot and minimize the power loss.

Figure 4-4. RLC filter on motor terminal [86], [94], [95], [103]

A simplified method of choosing Rf would be matching it to cable’s characteristic impedance when the frequency tends to infinity [95]. Depending on the model used for the cable, the parameters may be a little different.

풋흎풇푳풇푹풇 ퟏ 풁풇 = + Equation 4-3 푹풇+풋흎풇푳풇 풋흎풇푪풇

푳 푹 = 풁 = √ Equation 4-4 풇 풄 푪

This simplified method might not provide the best results, in terms of mitigating the overshoot or minimizing the power loss. Another approach [103] chooses Rf so that it results in an overdamped circuit.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 101

ퟏ 푳풇 푹풇 < √ Equation 4-5 ퟐ 푪풇

Usually filter capacitance and inductance are chosen using computer programs, aiming at the optimal point where the voltage overshoot and the filter power loss are in an acceptable range. Some other constraints may be introduced based on the experience or the specifications of the components, to narrow down the search space. For example, [103] suggest to select the resonant frequency of the filter to be five times the switching frequency of the PWM inverter. Moreover, the operating frequency of the RLC filter should be chosen to be near the resonant frequency to minimize the voltage overshoot.

b. Filters on Inverter Output

b.1. Reactor: high voltage drop, low dv/dt, low cost, simple

An easy and common way to increase the rise time of the voltage pulse and consequently reduce the voltage overshoot on the motor terminals is to add a series inductor to the inverter output, as it is shown in

Figure 4-5. Inductors are usually calculated as a percentage of base impedance of the system. Inductors of 3 to 5 percent are common in industry. Although the inductors are built to have a low resistance, the load current flows through them and the power loss might not be negligible, especially if they are installed in the inverter cabinet. Also, inductors are bulky and need a space to be installed. Moreover, higher values of series inductance can deteriorate the system’s power factor and cause a noticeable voltage drop [94].

Figure 4-5. Inductor on inverter output [94]

Another variant of the this filter can be used by adding a parallel resistor to the inductor to provide a more customizable filter [101]. The design process of such a filter is simpler compared to an RLC filter, while providing an acceptable overshoot and power loss in some applications.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 102

b.2. LC: low loss, low overshoot, complicated design due to resonance

A major disadvantage of using a passive filter is the active power loss in the filter resistor. One approach to decrease the power loss in the filter is to remove the resistor component of the filter and only use capacitors and inductors. An LC filter on inverter output can decrease the overvoltage effectively and with low power loss. This filter, similar to other filters with parallel capacitors, has the advantage of compensation of the reactive power needed by the cable and the motor. However, the design should consider the filter resonance and its complications. These filters are often designed with a resonant frequency, 푓푟푒푠, significantly below the switching frequency, and above the output fundamental frequency to avoid resonance with the load.

ퟏ 풇풓풆풔 = ⁄ Equation 4-6 ퟐ흅√푳풇푪풇

Figure 4-6. LC filter on inverter output [94], [96], [104]

LC filter commonly is installed with inductors in series and capacitors in parallel [94], [96], [104], as it is shown in Figure 4-14. Various topologies of this filter can be used depending on whether the common point of the capacitors are connected to the dc link or not [96]. Connection of the components determines the maximum voltage and current of the components and also affects the performance of the filter.

Moreover, connecting the capacitors and inductors in parallel and installing them in series with the line is also suggested [97].

b.3. RLC: high power loss, low overshoot

To avoid resonance in LC circuit, a resistor is added to the filter. Elements of an RLC filter can be connected in different ways. Traditionally, RLC filter comprises of an L in series with the load and an RC branch connected to the ground [86, 89], as seen in Figure 4-7(a). However, other combinations of the RLC

Chapter 4: Overvoltage Mitigation and Protection in VFDs 103

filter are also proposed. The disadvantageous of having the resistors in series with capacitors in traditional topology is that they conduct the filter current, thus larger high-power resistors are required. Increasing the resistors to decrease the current, will reduce the filter performance. To resolve this problem, resistors can be placed in parallel with capacitor (not shown here), or in parallel with inductors, as shown in Figure

4-7(b). In an RLC filter, it is possible to connect the neutral point of the wye connection of filter to the mid- point of the DC link to cancel the common voltage. However, in this case, circulating current that flows through filter elements and IGBTs should be considered [100].

(a)

(b)

Figure 4-7. RLC filter on inverter output [86], [94], [95]

Selection of the resistance of the filter is a compromise between the power loss and the damping. It should be selected to provide an effective damping while keeping the filter power loss in an acceptable range. One approach is to select the resistance to be equal to the cable characteristic impedance at high frequencies [95]. The reactance and the capacitance of the filter are usually calculated with the computer programs aiming to make a balance between the power loss of the filter and the voltage overshoot at the motor terminal.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 104

b.4. LC clamp: medium loss, extra components

As discussed above, resistors in an RLC filter damp oscillations to avoid high overvoltages. However, they add power loss and increase the size of the filter considerably. One alternative approach is to not use resistors and instead clamp the voltage at the DC bus with fast recovery diodes. One topology used for this filter is shown in Figure 4-8 [98]. In this filter, the LC resonating voltage is clamped to the dc-bus voltage, and the rise time of inverter output voltage can be controlled by the values of Lf and Cf of the filter.

Connecting the wye connection point of the capacitors to the midpoint of the DC link, also, helps to achieve the reduction in differential mode voltage and common mode current [102]. Although this filter would have more components compared to LC and RLC filters, its power loss is less than that of damping resistors [86], [94].

Figure 4-8. LC filter with diode clamp on inverter output [86], [94], [98], [102]

c. Filters on Both Motor Terminals and Inverter Output

c.1. RL-plus-C: low loss, low dv/dt

An RL-plus-C filter comprises of a parallel inductor-resistor that is connected in series with the line at the inverter output, as well as having a capacitor in parallel with the load. The basic idea of this filter is that

RL impedance is small at low frequencies and is equal to characteristic impedance at high frequencies. In high frequencies L impedance will be high and neglected. So, we can choose Rf = ZC [93]. Filter’s capacitor and inductor is calculated based on circuit impedances and the desired overvoltage on the load.

An RL-plus-C filter is seen in Figure 4-9.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 105

Figure 4-9. RL-plus-C filter [93]

c.2. RLC-plus-C: low loss, low dv/dt, low bearing currents

This filter can be considered a variation of RL-plus-C filter, with added capacitor to further decrease common mode and bearing currents. It also shows better performance in reducing the overvoltage spikes

[99]. RLC-plus-C filter also can be seen as a conventional RLC filter on inverter output plus extra Y connected capacitors on motor terminals, as it is seen in Figure 4-10.

Figure 4-10. RLC-plus-C filter [99]

4.2.2. Active Filters

Active filters are filters with both passive components and active components such as power electronic switches and amplifiers. They are common in power systems in power quality improvement and harmonic mitigation applications. The main concept in an active filter is to mitigate harmonics by injecting active power with the same frequency but with reverse phase. Because of the complexity and extra components, active filters can be expensive and can lower the reliability of the system. At the same time, they are more flexible and generally have lower power loss compared to passive filters. Usually for low power motor

Chapter 4: Overvoltage Mitigation and Protection in VFDs 106

drive systems, a passive filter is a common and low-cost solution. However, for high-power systems, investing in an active filter might be a solution to consider.

Some studies proposed to use active filters in motor drive systems to suppress the overvoltages [105]–

[108]. Various topologies of the active filters are proposed. Generally, active filters are comprised of parallel passive elements, a common-mode transformer to add a positive or negative voltage, a dc source and electronics switches to control the filter behavior. A general active filter used on the inverter output is shown in Figure 4-11. An active filter on inverter output

Figure 4-11. An active filter on inverter output [107]

This filter is effective in reducing the common mode voltage, common mode current and EMI. Z1 and

Z2 serve as common mode voltage detection circuit, where it used to control the switches and inject compensating voltages to the lines by the common mode transformer.

Some filters may include complex sampling, processing and control units. While more switches and more processing generally provide better results and lower power loss, these solutions may cost more and have lower reliability overall, due to extra parts adding to the possibility of the failure. An example of such a filter is shown Figure 4-12. This filter uses an energy-recycling module (ERM) to reduce the power loss in the previously proposed filter (also shown in the same figure), while effectively reducing the overvoltages on the motor terminal. Figure 4-13 shows the line to line voltage at the inverter output and the motor terminal. It is seen that peak value of the overvoltages is reduced from 570 V to 325 V, where the

Chapter 4: Overvoltage Mitigation and Protection in VFDs 107

pulse amplitude is 300 V at the inverter output. Power loss was 18% of the load power with a resistor used in the filter, instead of ERM. Although this is a huge power loss in the filter and requires a high-power resistor or possibly using a costly cooling system, still it is less than the power loss in the conventional RC and RLC filters. Using the proposed ERM, [105] reduces the power loss of the filter down to 2% of the load. Nevertheless, this reduction in the power loss comes at the expense of using an extra 6 power switches, 3 inductors and one common mode transformer, as well as higher complexity of the control system.

Figure 4-12. Active filter with energy-recycling module (ERM) proposed in [105]

Chapter 4: Overvoltage Mitigation and Protection in VFDs 108

(a) (b) Figure 4-13. Reduction of the overvoltages in a drive system with an active filter proposed in [105]: (from top) voltage at inverter output, middle of the cable and motor terminal (a) without the filter (b) with the filter

These components are an addition to the original active filter that used 10 capacitors, 3 inductors, 3 resistors, 6 power switches and 6 rectifier diodes. In practice, these extra components add to the cost of the filter and reduce its practicality in applying to a wide range of applications.

4.2.3. PWM Control

Fast rising PWM pulses causes reflection overvoltage on motor terminals. But at the same time, PWM technique provides immense flexibility in control of the motor. This capability of precise PWM control has also been used to solve the overvoltage problem. The target is to shape the inverter output voltage so that the rate of change in the filter output during voltage transitions becomes lower. Similar to active filter method, this method needs passive components as a filter, and it may need a feedback circuit and sensors.

Also, complexity of the control scheme is a disadvantage. However, its advantage is achieving the goal of active filters in reducing the motor overvoltages, using the existing power switches in the inverter.

Researchers have proposed different methods on utilizing the controllability of the inverter itself in overvoltage mitigation [106], [109]–[111]. Reference [106] uses an active dv/dt control to eliminate the overvoltages completely. Figure 4-14 and Figure 4-15 show the topology and the fundamental idea of their method. In this approach, a dead time is applied after the rising edge and before the falling edge of the pulse. After applying the pulse for the half of the rise time, it is reverted for the second half, reducing the average amplitude of the pulse to half. Thus, it will create a voltage reflection with the amplitude of one per unit. This is the same as supplying 2 pulses with a voltage of half the amplitude and duty cycle of 50%.

Chapter 4: Overvoltage Mitigation and Protection in VFDs 109

Output voltage of the filter will reach 1 pu at t = τ. At this moment no transient occurs, since the step voltage and the filter output voltage are the same. This approach needs an LC filter on inverter output.

However, filter is designed for higher cutoff frequency and makes it possible to use smaller filter components.

Figure 4-14. LC filter used in [106] to achieve active filtering with PWM control

Inverter output Filter output

With active control Without active control

Figure 4-15. Active dv/dt control for overvoltage mitigation, (top) inverter output, (bottom) motor terminal [79]

The above mentioned method assumes the load current to be zero. However, in real conditions, motor current will affect the charging and discharging sequence of filter capacitor. Inductor current will increase during the first half of the rising time when switches are on. Then, for the second half, switches turn off and filter current circulates through switch body diode. When there is a load current, the filter current needs to change direction in the charging and discharging sequence. In order to restore the current of the filter

Chapter 4: Overvoltage Mitigation and Protection in VFDs 110

inductor to the load current level, the power switch parallel to the diode that was conducting must be turned on at the point when the diode stops conducting. This adds to the complexity of gate pulse control.

This situation is depicted in Figure 4-16. Assume the load current is negative and the positive pulse edge occurs. When S1 turns on, the filter inductor’s current on phase A increases. For the second half of the rise time, S1 switch should turn off and the filter current should decrease and reach the previous stable point. Current flows through the body diode of the S2 switch until it reaches zero. At this moment, switch

S2 should turn on to provide a path for the circulating current. This is what is seen in the figure below for the lower switch. Turning on the switch in the same leg increases complexity of the control, since cautious measures should be considered for switches, not to be ON at the same time.

Figure 4-16. PWM control pulses when the load current is negative

As it is mentioned in Chapter 3, in a VFD system with a long cable and high switching frequencies it is possible that the amplitude of overvoltages exceeds 2 pu and even reach 4 pu, where 1 pu is the DC link voltage. Various reasons can create overvoltages greater than the dc bud voltage, including double pulsing and double transition. Double pulsing is when a pulse is applied shortly after another pulse and before the oscillations from the first pulse has been damped. Superposition of the oscillations from 2 pulses can create

Chapter 4: Overvoltage Mitigation and Protection in VFDs 111

overvoltages higher than 2 pu. The oscillations from the rising and falling edges of one pulse can also create a similar phenomenon. Double transition or polarity reversal occurs when the voltage of a line changes from –Vdc to Vdc. This is a pulse with an amplitude of 2Vdc that can create overvoltages of up to 3

Vdc (i.e. 4 Vdc from -Vdc).

A simple way to eliminate the over-2pu overvoltages due to double pulsing or short pulses is suppressing any pulse with width under a certain limit, tmin, to allow the oscillations to fully decay and eliminate the possibility of superposition of voltage reflection waves [7]. Another innovative way for overvoltage mitigation uses the oscillations from 2 near-by pulses, where the superposition is possible, and adjust the pulse width so that the oscillations from the next pulse event or the falling edge of a pulse superimposes the ones from the previous pulses or the rising edge of a pulse with a negative amplitude, offsetting a certain amount of the overall overvoltage oscillation [110]. The principle of this method is shown in the Figure 4-17.

(a) (b) Figure 4-17. Overvoltage oscillations from (a) switching on (b) switching off

Assume the rising edge of a pulse is on P1. If the falling edge P4 occurs at P3, the overvoltage oscillations from both events will boost each other and the overall overvoltages may go beyond the 2 pu limit. However, a Double Pulsing Offset Technique (DPOT) PWM control adjusts the adjacent pulsing events so that the falling edge P4 occurs on either P6 or P7. This way oscillations from one event will be offset by the other.

Another PWM control technique recommends avoiding double transition or polarity reversal (pulses from –Vdc to Vdc) [7]. Selecting the switching frequency with regard to the system frequency response is a crucial step in the VFD design. Considering the components, the switching frequency should be selected to

Chapter 4: Overvoltage Mitigation and Protection in VFDs 112

be far from the resonance frequency of the system [109]. These cautions along with the previously mentioned ones will limit the overvoltages to at most 2 Vdc.

4.2.4. DC Cable between Rectifier and Inverter

As it established already, a typical VFD creates multiple problems in the system, most of which are associated with the reflection phenomenon in AC long cables. It is possible to eliminate the voltage reflection overvoltages in AC drives using a DC long cable between the rectifier and the inverter, instead of using an AC long cable between the inverter and the motor. This approach is illustrated in Figure 4-18.

The main advantage of using a dc cable is obvious: no reflected wave. That also translates to no overvoltages, smaller common mode currents, smaller cable charge current, and no filter needed. There are other advantages, such as using two long cables instead of three, and low voltage drop on the cable. In very long cables, such as submarine cables, the reactive power of the cable itself, occupies the cable current capacity. Using a dc cable instead can eliminate the reactive power of the cable. However, using a dc cable has its own disadvantages, too.

Short Circuit protection is a fundamental aspect of every electrical system. The proper functioning of most of the traditional relays and fuses is based on having a zero crossing in the current. Since the dc current does not have a zero crossing, short circuit fault protection in DC link are more difficult and need more expensive protection equipment. Other disadvantages of installing a long dc cable include separate installation of the inverter and the rectifier that might not be possible in all situations, costly maintenance, and higher harmonic currents and losses [68].

Figure 4-18. Long DC cable connection between the rectifier and the inverter [112]

Other than technical advantages and disadvantages, the widespread use of a system depends on the cost analysis of that system. In regard to the above mentioned DC-Cable VFD system, it is shown that a

Chapter 4: Overvoltage Mitigation and Protection in VFDs 113

noticeable saving can be achieved in terms of copper volume and cost [112]. At the same time, installing a

DC capacitor bank across the inverter terminals is necessary to eliminate the DC transient overvoltage peaks at the inverter. Extra cost of the DC circuit breakers should also be considered.

4.2.5. Energy Varistor

Conventionally, varistors are used to protect against surges and overvoltages that occur limited times during their lifetime. Generally, during a discharge, an MOV conducts a large current and heats up for a short time. But since the event is not persistent, the MOV has time to cool down to the environment temperature and be ready for the next event. When an MOV is used to mitigate the repeating high frequency overvoltages, the MOV repeatedly discharges high-peak impulse currents when each pulse is applied. Also, after each pulse, it has a short time to cool down until the next pulse (depending on the switching frequency). Manufacturers do not design MOVs for repetitive pulse discharge. There are no application notes or standards on how to select the MOV for these situations. The effects of the repeating pulses on the MOV’s performance and lifetime are unknown. However, the possibility of the concept is proven by a research experiment that uses energy varistors in a long-cable motor drive system on motor terminals [113]–[116]. These experiments show the effectiveness of MOV application for overvoltage mitigation and serve as a proof of concept. Also, there are attempts to show robustness of the MOVs by an accelerated test of 1000 hours in 140 °C, 55 degrees above the maximum operating temperature of the

MOV. However, this experiment provides extra cooling using an aluminum plate attached to the cupper electrodes.

Figure 4-19(a) shows the reduction of the overvoltage with application of MOVs on motor terminals in line to ground connection. The DC link voltage is 600 V and the rise time of the pulse at inverter output is

100 ns. Cable length is 130 m. Without a varistor, the peak voltage on motor terminal reaches to 1200 volts.

Installing MOVs on motor terminal, reduces this voltage to 1020 volts. The power loss in the MOV depends on the amplitude of overvoltages and its ratio to the MCOV of the varistor as well as switching frequency. Figure 4-19(b) shows the power loss for an energy varistor for different switching frequency

Chapter 4: Overvoltage Mitigation and Protection in VFDs 114

and different cable types. Also, MOV cooling effect is considered in this figure. For extra cooling, an aluminum plate is attached to copper electrodes to provide better heat dissipation. It is seen that for a given condition, an MOV can be used for overvoltage mitigation up to a certain switching frequency. Cable shield should also be taken into account. For example, this set up can work continuously up to switching frequency of 8 kHz, without any extra cooling. See pink points in Figure 4-19(b).

(a) (b) Figure 4-19. Overvoltage mitigation with MOV [113]

Installing MOVs on motor terminals affects overvoltages in two distinct ways. One effect comes from the capacitance of the varistor. Varistors’ capacitance at the end of the cable act like a filter and reduces the overvoltages. If the varistors MCOV is close to peak value of the overvoltages, the resistive current will be negligible and the capacitance of the varistor is the only limiting factor of the overvoltages. In this condition, a capacitive current will flow through the MOV. The rise time of the voltage at motor terminals also increases due to the change in capacitance of the circuit. Another effect that limits the overvoltages at motor terminal is due to varistor’s voltage clamping aspect. When an overvoltage amplitude increases, a resistive current also flow through the varistor in addition to the capacitor current discussed above. This limits the voltage on motor terminal which is connected parallel to the MOVs. The higher the amplitude of the overvoltage compared to varistor’s MCOV, the higher the resistive current. It is the resistive current that causes the power loss in the varistor and heats it up. Figure 4-20 shows overvoltage mitigation by two varistors with different MCOV for a motor drive system with the DC link voltage of 800 V. The amplitude of voltage peak at motor terminal is 1600 V. In Figure 4-20(a) varistor is chosen so that it conducts a very

Chapter 4: Overvoltage Mitigation and Protection in VFDs 115

(b) (a) Figure 4-20. Current and voltage of the MOV [114] small resistive current (MCOV is close to voltage peak). Voltage peak is decreased to 1400 V. Here, the mitigation is only caused by varistor’s capacitance. In Figure 4-20(b) the varistor draws about 12 A current at the peak of the voltage. This is the peak value of the resistive component of the varistor current. In this condition, overvoltage peak value drops to 1100 V. In this condition, the power loss in the varistor depends on the amplitude of the current and voltage as well as the switching frequency.

Chapter 5: MOVs in Variable Frequency Drives 116

Chapter 5:

5. MOVs in Variable Frequency Drive Systems

In this chapter, MOV behavior in VFD systems is investigated. Conventionally, MOVs are designed to discharge infrequent currents due to lightning and occasional power system switching surges. When MOVs are used in VFD systems, they may be exposed to repetitive pulses created by switching. These pulses can discharge repetitive currents through MOVs and cause over-heating and eventually failure of the MOV. To prevent this, designers have to select MOVs with higher voltages to ensure that MOVs can tolerate the overvoltages. This approach diminishes MOVs capability and performance to protect the system efficiently, since appliances are subjected to higher overvoltages. There has been limited research and publications on the topic to explain the phenomena. This chapter provides test results and discussion on the behavior of the MOVs when repetitive pulses of different amplitude and duration are applied. Furthermore, new approaches are proposed to allow MOVs to be safely operated in the VFD systems, while providing a proper protection level for the motor and the VFD system.

Chapter 5: MOVs in Variable Frequency Drives 117

5.1. MOV installation on VFD systems

In general, MOVs provide a low impedance path to the ground for the surge current. Typically, MOVs are installed between lines and ground. When there is access to the neutral wire, additional MOVs might be placed in line to neutral mode. Occasionally, extra MOVs might be placed in line to line mode to provide an even better protection. Considering the cost and the sensitivity of the equipment and the surge environment where the system is located, the surge protection designers might choose to use a 3-mode, a 4- mode, a 7-mode, or a 10-mode protection scheme [117]. These combinations are shown in Figure 5-1.

Depending on where the MOV is installed on the VFD cable, it may be subjected to different voltage amplitudes and waveforms. Section 3.4 describes the voltages in the VFD systems in details. The most common location to install the MOV in the VFD system is on the motor terminal, between line and ground.

Some SPDs provide extra protection by adding line to line and line to neutral MOVs.

Figure 5-1. Modes of surge protection in 3-phase systems

5.2. Behavior of MOV under Repetitive Pulses

Traditionally, MOV manufacturers test some samples for standard surges for a limited number of impulses. Typically, MOVs are not designed, nor tested for repetitive high frequency pulses. With expansion of sensitive electronic devices to many fields and applications, MOVs are going to be used in systems with non-traditional voltage waveforms. One of these new applications is using MOV-based SPDs to protect VFD controlled motors. In these systems, MOVs are needed to be installed to protect the cable between the inverter and the motor against lightning surges. These MOVs are subjected to a PWM voltage.

Chapter 5: MOVs in Variable Frequency Drives 118

Little studies have been conducted on the behavior of the MOVs under such voltages. As a result, engineers usually over-design and install MOVs with a higher MCOV to avoid any possible failure. This practice reduces the protection performance and puts the expensive and sensitive appliances at risk.

As previously stated, in a VFD system, voltage pulses at inverter output are clean with minimum distortion. On the other hand, pulses at the motor terminal are distorted with high spikes. This is shown in

Figure 5-2. The amplitude and the frequency of the spikes depend on the length of the cable and the specifications of the inverter. For a motor drive system with a long cable (lcable > 10m) surge protection on both ends of the cable is recommended by IEEE [91]. In such a configuration, although both SPDs are installed on the same line (one on the inverter output and the other on motor terminals), the voltage pulses, conducted currents and the power loss that they experience are considerably different.

In order to investigate the behavior of the MOVs under voltage pulses, this chapter experimentally injects pulses from a power supply, as well as from a VFD inverter. Pulses from the power supply have long rising and falling times (100 ~ 150 µs). Pulses from the VFD have fast rising and falling edges (100 ~

150 ns).

Figure 5-2. PWM pulses in a VFD system

5.2.1. MOV’s Response to slow Pulses from a power supply

In this experiment, MOVs are subjected to millisecond range pulses from a voltage source to study their behavior under pulses. A programmable Chroma Power Supply 61504 is used to create pulses with pre- determined width, amplitude and number, in this experiment. Figure 5-3 shows the test configuration. The

Chapter 5: MOVs in Variable Frequency Drives 119

voltage source has maximum pulse voltage of 424 V, with 2 kVA of apparent power that can deliver about

50 A for a short time in ms order. The voltage rise is slow, and the rise time of the pulses is about 100 µs.

The waveform is shown in Figure 5-4. The pulse waveform is not exactly square and has an overshoot of about 10 V. The rise time, the time required for a pulse to rise from 10 percent to 90 percent of its steady value, is 110 µs.

Figure 5-3. Pulse application to 150 TPMOV with the programmable Chroma power supply

350 300 250 200 150

100 Voltage Voltage (V) 50 0 -0.5 -50 0 0.5 1 1.5 2 2.5 Time (ms)

Figure 5-4. Pulse waveform applied with the programmable Chroma power supply for a 300 V pulse

Measurement of the MOV current is a difficult task. The normal leakage current in voltages below its

MCOV is less than a milliampere, with its resistive component being around 100 µA. For higher voltages

MOV current can increase rapidly and reach several amperes for a short time.

For microampere range, having a 100-ohm current sensing resistor in series with the MOV is acceptable and can provide a good signal to noise ratio to avoid noise and errors in measurements. However, for higher currents in the range of several ampere, the 100-ohm resistor will cause a large voltage drop in the

Chapter 5: MOVs in Variable Frequency Drives 120

applied voltage to the MOV. One way to alleviate this challenge is to change the sensing resistors based on the maximum current. Another approach is to use clamp current probe with an electronic amplifier. In this experiment two Tektronix TCP312A current probes are used. One has 30 A max current with a frequency range of dc to 100 MHz. The other one is Tektronix TCP0150 with 150 A max current and frequency range of dc to 20 MHz that are used to measure currents higher than 30 A.

Since the MOV has a parasitic capacitance, every change in its voltage will cause a current through it.

We call this part of the MOV current, the capacitive component. Two spikes of capacitive currents are shown in Figure 5-5. Zooming into the rising edge of the pulse shows that a peak of 15.2 mA current as a result of 50 V change in the voltage in a time span of 25 µs. This gives an approximate capacitance of 7.6 nF.

풅풗 푰 풊 = 푪 → 푪 = ∆풕 푪 → 푪 = ퟐퟓ 흁풔 × ퟏퟓ. ퟐ 풎푨 ÷ ퟓퟎ 푽 = ퟕ. ퟔ 풏푭 Equation 5-1 풄 풅풕 ∆푽

This value is consistent with our measurements for the MOV with a precision LCR meter, as it is shown later in this chapter.

VMOV

IMOV

Figure 5-5. 150 V Pulse applied by the power supply to a 150TPMOV and MOV current

Assume a simple model of a constant capacitor in parallel with a non-linear resistor for the MOV. The current between two edges of the pulse, where the voltage is almost constant, here called the resistive

Chapter 5: MOVs in Variable Frequency Drives 121

component, is responsible for power consumption and heating. It is small when the voltage pulse amplitude is in proximity of MCOV, thus it is not accurately measurable with our current probe. Increasing the pulse voltage, increases both capacitive and resistive components of the MOV current. Capacitive current increase is proportional to voltage increase, as ΔV/Δt increases. It is expected considering Equation 5-1.

Resistive current, on the other hand, increases exponentially with increasing voltage and its behavior is affected by multiple factors. For example, Figure 5-6 shows the MOV current for 260 V pulse. The capacitive peak current has increased to 28 mA, as expected from voltage change. Resistive current, is negligible for voltages up to about 200 V and then increases nonlinearly with voltage increase. The peak of current corresponds to the peak of voltage. Then, current decreases while the voltage amplitude is almost constant.

Figure 5-6. 260 V Pulse applied by the power supply to a 150TPMOV and MOV current

(a) (b)

Figure 5-7. (a) MOVs used in the experiment with and without case (b) MOV's structure

Chapter 5: MOVs in Variable Frequency Drives 122

There are multiple factors affecting the waveform of the MOV’s current. First, the inductance of the ceramic itself. Although it is small, it causes a small delay in rising the current. The waveform of the voltage pulse also plays a role in the waveform of the current. The pulse has a 10 volts overshoot that corresponds the peak value of the current, then both voltage and current decrease. However, after the pulse voltage settles and remains constant, the current continues to decrease. This should be justified by taking the MOV’s microstructure into account. This is similar to the behavior of the MOV when a DC voltage is applied. The leakage current starts from a higher value and decreases until it settles on a value. That is why for the MOV’s leakage current measurement, it is instructed to read the value 2 seconds after applying the voltage. Another factor that also plays a role in the current waveform is the heating of the MOV. The current of an MOV increases with increasing temperature for a given voltage. For the applied pulses, in lower currents, where MOV does not heat up, the current decreases after reaching its peak. For higher currents, the heating effect has higher impact and we can see the current increases. This can be seen in the below figures. Figure 5-8 shows the waveform of the MOV current for four pulses with different amplitudes. As it is seen in Figure 5-8(a), the peak value of the current is 2.5 A. The MOV current decreases after reaching its peak. On the other hand, in Figure 5-8(d) the current peak is 34 A. Here heating of the MOV during the pulse increases its current.

(a) (b)

(c) (d)

Figure 5-8. Current waveforms of MOV for 1 ms voltage pulse (a) 2.5 A (b) 17A (c) 27 A (d) 34 A

Chapter 5: MOVs in Variable Frequency Drives 123

For pulses in-between the two, the effect of current decrease due to microstructure and the effect of current increase due to the heating affect the waveform. Depending on what factor has more influential effect the current might increase, decrease or remain constant, as it is seen the Figure 5-8(b) and Figure

5-8(c).

To investigate the behavior of the MOVs under millisecond-range pulses further, 3 MOVs are selected to be subjected to pulses with various width. These MOVs have the same MCOV of 150V, but different nominal voltages, all of which are within the standard range. They are listed in the Table 5-1. The number of pulses applied, their width and the interval between them are also listed in Table 5-2. The amplitudes of the applied pulses, start from 150 V and increases in steps of 10 V, until the power supply reaches its current limit of 50 A. The distorted pulses, where the distortion is clearly from power supply current limit, are discarded from the analysis. In total, 200 pulses are applied to 3 MOVs, and the results of each experiment are compared to others to find out if pulse width and interval between pulses have any effects on the waveform and peak value of the MOV current. The capacitive current is neglected and only the resistive current is considered.

Table 5-1. MOVs used in pulse experiment and their nominal voltages

MOV code V1mA (V) MOV230 P150-13 230 MOV240 P150_12 240 MOV250 P150_15 250

Table 5-2. Number of pulses, their width and intervals applied to MOVs Experiment Number of Pulses Pulse Amplitude Pulse Width Interval code in the sequence (V) (ms) (ms) CP1 1 150 to 370 0.4 - CP2 1 150 to 350 0.7 - CP3 1 150 to 350 1 - CP4 1 150 to 340 1.5 - CP5 2 150 to 370 0.4 3.6 CP6 2 150 to 370 0.4 0.2 CP7 2 150 to 350 1 0.2 CP8 10 150 to 300 4 36

Chapter 5: MOVs in Variable Frequency Drives 124

The first expected result of the experiment is that V1mA of the MOV is the major factor in the MOV current. For example, as it is seen in Figure 5-9, for a 310 V pulse, MOV230’s peak current is 32.8 A. This value for MOV240 and MOV250 is 26.8 and 6.0 amperes, respectively. Furthermore, we should notice that the current delivering capability of the power supply is limited, and it has an internal impedance, too. When applying the pulse by a supply, MOV will limit the output voltage by discharging a larger current. Here, although the settings of the pulse are the same for these three, the pulse voltage is not. For MOV230, where the supply delivers a large current, the pulse voltage has dropped to 294 V. For the MOV240, the pulse amplitude is 302 V, and for MOV250 it is 314 V. If the supply had higher short circuit current capability,

MOV 230 would draw even larger current from the power supply. Applying multiple pulses with different amplitude, we can measure the peak current for different voltages and draw the curves in Figure 5-10.

MOV230 MOV240 MOV250 Figure 5-9. MOV current for a 310 V pulse

MOV230 MOV240 MOV250 MOV230 MOV240 MOV250

50 50 40 40 30 30 20 20 10 10 0 0

150 200 250 300 350 400 150 200 250 300 350 400

MOV MOV current Peak (V) MOV MOV current Peak (V) Pulse voltage setting (V) MOV Voltage Peak (V)

(a) (b) Figure 5-10. MOV peak current vs. (a) pulse voltage setting (b) MOV voltage peak

Chapter 5: MOVs in Variable Frequency Drives 125

Figure 5-10(a) shows the current peak vs. the pulse voltage setting (pulse amplitude if the output of the power supply is open). This chart shows what MOV current to expect, if we know the open circuit voltage.

For example, if we know the DC link voltage in an inverter, this value will also be equal to open-circuit pulse voltage amplitude. However, this chart is dependent on the impedance of the power supply. A more useful chart would be the one shown in Figure 5-10(b) that shows the MOV current vs. MOV voltage.

These charts approximately match the datasheet VI characteristics. However, there is an important distinction: Typically, when referring to the MOV’s VI characteristics, the upper voltage limit is important for currents higher than 1 mA. That is why the manufacturers do not specify the lower limits for Amp- range currents, as they do for sub milliampere currents. This is because traditionally engineers only care about MOV’s leakage current. For higher currents, including tens of amps, the only traditionally important factor is the upper limit of clamping voltage. However, for application of the MOVs in VFD systems, they may conduct currents in the range of several amperes. It becomes important to know the highest current that an MOV will draw for a specific voltage. This is the lower limit curve that is added to Figure 5-11, just to show the concept.

Figure 5-11. VI characteristics of MOVs used in the experiment [58]

Chapter 5: MOVs in Variable Frequency Drives 126

Further experiments are conducted to study the relation of pulse width (in milli-second range) and the

MOV current. For this purpose, the above experiment is repeated for different pulse widths, as seen in

Table 5-2. The shortest pulse width that the power supply can create is about 0.4 µs. Thus, it is decided to apply pulses with width of 0.4, 0.7, 1.0 and 1.5 µs. Waveforms for the 300 V pulse applied to MOV230 are seen in Figure 5-12. It is clear that pulse width affects the maximum peak value of the current. If the pulse duration was enough for the MOV current to reach its peak, and also if the MOV current amplitude was constant after reaching its peak, MOV current would be independent of the pulse width. Since it takes some time for the MOV’s resistive current to rise, so the current peak is smaller for the short pulses. However, the second factor can affect the current both ways. It is described before that the MOV current during the pulse can be increasing, constant, or decreasing, depending on the amplitude of the MOV and its heating specifications (see Figure 5-9). Thus, if the current is increasing the peak value of the current will be higher for wider pulses, as it is seen in Figure 5-12. On the other hand, if the current is decreasing during the

pulse, as it is seen in Figure 5-13, it is not the case.

1.5 ms

1.0 ms

0.4 ms 0.7 ms

Figure 5-12. Pulse voltages and MOV currents for 300 V pulses for MOV230

Chapter 5: MOVs in Variable Frequency Drives 127

Figure 5-13. Pulse voltages and MOV currents for 310 V pulses for MOV250

It is seen in Figure 5-13 that 0.7 ms pulse is wide enough to allow the current to reach its peak. For pulses wider than that, the current peak is the same, although the average current and the average power is decreasing. For pulses narrower than 0.7 ms, the MOV current does not reach its peak, thus, has a smaller peak value.

Another question that can be asked after reviewing these results is whether adjacent pulses can affect the MOV current. For a resistor, two identical pulses create two identical current waveforms, but this is not true for an MOV. Because of the MOV’s microstructure, it takes some time for the current to rise. Also, for a constant dc voltage, the current starts to fall and settles after some time. Experiments CP5 to CP7 intend to investigate this topic. It is found that the pulses can affect the MOV’s current for the subsequent pulses.

Figure 5-14 shows an oscilloscope screen capture of two pulses of 300 V and 1 ms, applied to

MOV240. The interval between pulses is set to 0.2 ms. It is clearly seen that the MOV current during the second pulse does not rise as fast as it does during the first pulse. This effect is more seen in pulses close together, and less in the pulses that are apart from each other.

Chapter 5: MOVs in Variable Frequency Drives 128

Figure 5-14. two pulses of 300V, 1 ms, applied to MOV240

(a) (b) Figure 5-15. two 310 V, 0.4 ms pulses applied to MOV230 with interval of (a) 0.2 ms (b) 3.6 ms

Figure 5-15 shows MOV230’s current for two pulses of 310 V, 0.4 ms long. Both first pulses discharge

28.8 A currents. However, the MOV currents during the second pulses are different. When the interval between pulses are large, the second pulse’s current is almost as large as the first pulse’s one, but when the pulses are close to each other, the MOV’s current during the second pulse reaches 26 A which is almost 2 amperes less than the first one.

In the final experiment in this section, ten pulses with 4 ms width and 36 ms interval are applied to a

150V MOV. The amplitude of pulses is changed from 150 to 300 V. In all amplitudes, it is seen that the

MOV conduct higher current during the first pulse, and it decreases for the subsequent pulses. Pulses of

290 V are seen in Figure 5-16. More details can be seen in Figure 5-17, where the 1st, the 5th, and the last pulse are shown.

Chapter 5: MOVs in Variable Frequency Drives 129

Figure 5-16. ten pulses of 290 V and 4 ms applied to an MOV

Figure 5-17. MOV current during ten consecutive pulses (a) 1st (b) 5th (c) 10th

It is seen that the current peak for the first pulse is 2.32 A, where it drops to 1.55 A and 0.98 A for the

5th and 10th pulses. Not only the peak value of the MOV current decreases in consecutive pulses, but also the steady state part of the current, here called Iss, also decreases. These parameters are depicted in Figure

5-18 for 290 V pulses.

2.5 Ipeak (A) 2 Iss (A)

1.5

1 Current(A) 0.5

0 1 2 3 4 5 6 7 8 9 10 Pulse number

Figure 5-18. MOV's peak and steady state currents for ten consecutive pulses

Chapter 5: MOVs in Variable Frequency Drives 130

5.3. Behavior of MOV under PWM Pulses

When an MOV is installed to protect the cable that is connecting an inverter to a motor, it is subjected to a PWM voltage. On the inverter output these pulses are clean with almost no overshoot. On the motor terminal, the waveform depends on the length of the cable. It can be relatively clean PWM with small voltage overshoots for short cables, or there can be high frequency overvoltages occurring on all edges of pulses. In this section we are going to see how MOV behaves under PWM pulses. This sectin shows that the capacitance of the MOV has a noticeable effect on the overvoltages. Adding an MOV to the end of line, regardless of its nonlinear resistance and because of its capacitance, increases the period of overvoltage oscillations and reduces their amplitude. Also, it is discussed how to calculate the resistive current from voltage and current waveforms. Suggestions are made on how to choose a proper MOV.

Typically, the rise time of a PWM inverter with IGBT switches is in sub-microseconds. Such fast- changing voltages can discharge large capacitive currents through the MOV. A simple calculation below shows how large MOV’s typical capacitive current can become. Although large capacitive currents do not cause any power loss or heating in the MOV, they flow through the cable and cause power loss and voltage drop in the cable and take up its current capacity. Large currents also may trigger the over-current protection relays and cause unwanted trips.

Rise time = 200 ns

Pulse amplitude = 340 V => iC = C dV/dt = 6.8 A

MOV capacitance = 4 nF

For this experiment, three Y-connected MOVs are installed on the 3-phase motor terminals. The common terminal of the MOVs are connected to the ground. The test setup is shown in Figure 5-19. The current and the voltage of the MOVs are measured and captured by a Tektronix oscilloscope.

Chapter 5: MOVs in Variable Frequency Drives 131

Figure 5-19. MOVs connection for motor protection

Vag @ Inverter output

Vag @ Motor terminal

MOV current

Figure 5-20. Line to ground voltages and MOV's current

(a) (b) Figure 5-21. (a) Motor current (b) MOV and Motor currents superimposed

Figure 5-20 shows an MOV’s current, connected between a line and the ground, on the motor terminal.

As it is seen from the figure, MOV conducts large currents, mainly capacitive, at each pulse edge. The peak value of the current is 4.7 A for this MOV. The motor’s current is an ac current with the maximum value of

Chapter 5: MOVs in Variable Frequency Drives 132

about 2 A. The peak value of MOV’s current is more than twice that of the motor. The load current for the cable (MOV plus motor), reaches more than 7 A. Setting of the over-current relay on the inverter side should consider these high peak currents. In fact, in the lab setup, these high currents tripped the drive several times, so we had to change the over-current protection settings.

Other than drawing large capacitive currents from inverter, MOV’s capacitance has another effect on the VFD system, too. Installing an MOV changes the waveform of the overvoltages, because of its large capacitance. Taking the setup in the lab, the cable’s capacitance is 130 pF/m. For the 29-meter cable, the total capacitance comes to 3.77 nF. According to its datasheet, A 150 V MOV’s capacitance is about 4.8 nF at 1 KHz. This large capacitance at the end of the cable changes the reflection coefficient and the waveform of the voltage oscillations. The line to ground waveform after connecting an MOV is seen in Figure 5-22.

The frequency of the oscillations has decreased, and the period has increased to 1.5 µs from 0.8 µs.

Figure 5-22. Line to ground voltage after installing an MOV

This can be verified in the simulation, too. Adding a large capacitance to the end of the line, increases the period of the oscillations. Below in Figure 5-23, the same VFD system from section 3.5.6 is used. Three

4.8 nF capacitors are connected on the motor terminal between lines and the ground. It is seen from the simulation result in Figure 5-24 that the oscillations period has increased to 1.506 µs. Also, the current flowing through the MOV’s capacitance has the peak value of 5.45 A, which is 13% less than the measured value of 6.2 A, shown in Figure 5-22.

Chapter 5: MOVs in Variable Frequency Drives 133

Figure 5-23. VFD system simulation with MOV's capacitance

1.506 µs

Figure 5-24. Simulation results: MOV's voltage and capacitive current

Another important aspect of MOV’s behavior is its resistive current. As it is stated in the previous section, the capacitive current is proportional to the voltage change rate, and the resistive current is dependent on the MOV’s voltage amplitude. For the voltage below the MOV’s MCOV, the resistive part of the current is small and negligible.

A simple way to measure the peak value of MOV’s resistive current is to take the current value, when the voltage waveform’s derivative is zero, at its peak. Since the capacitive current is proportional to dV/dt, its value at this moment is zero. In Figure 5-24, it is seen that for only a capacitor, the momentary current at the peak of the voltage, i.e. the resistive current, is zero. For a real MOV, the resistive current is not zero, but for voltages below MCOV it is small. Figure 5-25 shows the current of a 420V MOV for a pulse. The peak value of the line to ground voltage is 390 V. As it is seen from the figure, for this voltage, the resistive current is almost zero and the current waveform is zero at the peak of the voltage.

Chapter 5: MOVs in Variable Frequency Drives 134

Figure 5-25. 420V MOV's voltage and current

Figure 5-26. 150V MOV's voltage and current

Voltage and current waveforms of a 150V MOV is shown in Figure 5-26. The peak value of the voltage is 320 V. As seen in the figure, the resistive current at the peak moment of the voltage is not zero. The resistive current’s peak at this moment is about 3 A. This resistive current causes power loss and heating of the MOV. If the average power loss is higher than the MOV’s heat dissipation capability, its temperature will rise. If there is a thermal protection, it will open and MOV will be disconnected, as it is in the lab setup. If there is no thermal protection, the temperature rise can reach a thermal runaway point and will lead to MOV’s failure. So, it is important that designers of MOV application in non-conventional applications measure the resistive current and make sure the average power loss does not exceed the power dissipation capability of the MOV. Since this power loss occurs in a pulse, obviously the switching frequency of the PWM voltage (more accurately, number of pulses in a second) also should be taken into account. As a general rule, this resistive current should preferably be negligible to have a desirable safety margin.

Chapter 5: MOVs in Variable Frequency Drives 135

5.4. High Frequency Model of MOV

In today’s industry, MOVs are not designed for high frequency voltages and there are no standard high frequency tests that manufacturers should comply. In datasheets there are no information on high frequency specs and capabilities of the MOV. However, there are a few academic publications on some aspects of the high frequency behavior of MOVs [15], [118]–[123].

One of the early efforts to characterize the low voltage MOVs under high frequencies is reported in

[122], where it is showed the resistance of an MOV decreases almost linearly with the inverse frequency.

Adding this statement to the known fact that MOV’s impedance is dependent on the current flowing through it, and the leakage current of the MOV is dependent on the temperature, they proposed a model for higher frequencies, as shown in Figure 5-27.

Figure 5-27. MOV’s frequency- and temperature-dependent model

Dependence of the MOV’s impedance on temperature is measured with DC leakage current measurement. The dependence on the frequency is measured by applying sinusoidal voltage to the MOV with a signal generator (20 V peak to peak) and sweeping the frequency.

In recent years, other models are also proposed to provide a better accuracy for high frequency. Four of the models are seen in Figure 5-28. The primary elements of these models are the same, a non-linear resistor in parallel with a capacitor. The other elements are added to account for the parameters such as the lead’s resistance and inductance, or the inductance of the ceramic body.

Chapter 5: MOVs in Variable Frequency Drives 136

(a) [120] (b) [119]

(c) [15] (d) [118]

Figure 5-28. MOV's high frequency models in literature

In models in Figure 5-28(a) and (b) an impedance is added to the model to represent the skin effect.

Skin effect can be modeled with a ladder network of resistors and inductors. In the simplest form it is considered as a resistor connected in parallel with an inductor, R1 and L1 in Figure 5-28(a) and Zskin in

Figure 5-28(b). Model parameters are determined by numerical algorithms by fitting the measured impedance to the calculated impedance.

The model shown in Figure 5-28(d), named metal-oxide surge arrester wide-range (MWR) model by the authors, is intended to provide an accurate model for a wide range of frequencies and amplitudes.

Inductance, L, is determined by experimental formulas and dependent on the height of the arrester and the number of ZnO columns in parallel. Similarly, the capacitance is considered constant and is calculated by experimental formula depending on arrester’s height, line discharge class and number of parallel internal columns of ZnO varistors. RL and Rc are added to the model only for the simulation purpose [118].

The accuracy of the model is mostly dependent on how the non-linear resistor is determined. The general approach is to measure the voltage of the MOV for different applied currents and draw the

Chapter 5: MOVs in Variable Frequency Drives 137

measured voltage vs. current curve. Then, a nonlinear curve fitting can determine the relation of voltage and current of the non-linear element in the model. There are two factor that are affecting the accuracy of the model. First, obviously, the accuracy of the model depends on how well the assumed function can fit the measured data. One common function is used in [118] as below:

휶 풗 풊 = 풌 ( ) Equation 5-2 푽풓풆풇 where i and v are the current and the voltage of the varistor. α is called the non-linear factor and k is a constant multiplier. Vref is an arbitrary reference voltage, which normalizes the equation. Although

Equation 5-2 is simple, but it does not fit well on MOV’s I-V curve. To have a better fit, [119] proposes a different function as below:

푵 풗−푽ퟎ 푽 풊 = 푰ퟎ [풍풏 (ퟏ + 풆 ퟏ )] Equation 5-3

Where N is an integer number.

Another function that is proposed in [16], used in [15], and is adopted by the manufacturer of the

MOVs used in this thesis, is as following.

−퐥퐨퐠 (퐈) 퐥퐨퐠 (퐈) 푽 = ퟏퟎ [퐛ퟏ + 퐛ퟐ 퐥퐨퐠 (퐈)+ 퐛ퟑ 퐞 + 퐛ퟒ 퐞 ] Equation 5-4

The second factor affecting the accuracy of the model is how the voltage and current are measured in different regions of the curve. As it is described in Chapter 1, an MOV’s I-V curve have 3 regions: leakage current region, normal clamping region, and up-turn region. Current is in micro- or milli-amperes in the leakage region, while it can surpass tens of thousands of amperes in normal operation and up-turn regions.

Measuring current and voltage in these regions are conducted with different methods. Typically, in the leakage current region voltage of the MOV is measured when a given dc current is applied. This is how the manufacturers create the I-V characteristic curve. For currents around the nominal discharge which can be several thousand amperes typically 8/20 µs standard surge is used. It should be noticed that the MOV’s voltage might be different for faster or slower surge waveforms. Figure 5-29 shows the normalized values for voltage measurement waveforms for a typical MOV [118]. It is seen that the clamping voltage for 4/10

µs waveform is 20% to 30% higher than that for the standard 8/20 µs. In the other end of the curve, in the

Chapter 5: MOVs in Variable Frequency Drives 138

leakage current region, it is seen that the AC curve do not match the dc curve. The ac leakage current peak value if much higher than the dc leakage current for low voltages.

Figure 5-29. waveforms used for I-V curve measurement in [118]

Figure 5-30. MOV's leakage current for different frequencies [123]

The AC leakage current itself is dependent on the frequency. The higher the frequency, the higher the leakage current. An example of the ac leakage current in different frequencies is seen in Figure 5-30. It is seen that while the dc leakage current is in micro-amperes, the leakage current in 50 Hz is in milli-amperes and the leakage current for 13 kHz can reach a tenth of an ampere.

Depending on the application, region of interest might be different for modeling. Traditional view is more concerned about the highest clamping voltage which occurs in the normal operation region. Leakage

Chapter 5: MOVs in Variable Frequency Drives 139

current region is also important for thermal analysis. However, since the MOV is selected to have a small- enough leakage current, the models do not need to be accurate in this region. In new applications, most models are interested in accurately modeling both the leakage current region and normal clamping region.

In an application like VFD protection, the MOV is subjected to repetitive pulses with high frequency and the thermal stability of the MOV is critical for selecting the proper MOV. Simulating the MOV accurate enough to be able to calculate its current and power in steady state condition can be helpful in selecting the proper MOV for VFD systems and non-sinusoidal applications, in general.

5.4.1. Simulation of MOV in VFD system

As it is discussed, the proposed models for the MOV have little difference with each other. The accuracy of the model mainly relies on the accuracy of modeling the nonlinear resistance. Here, we used the model shown in Figure 5-28(c). This model is composed of a constant capacitor in parallel with a nonlinear resistor. A small resistor and a small inductance also are connected in series to model the resistances and inductances of the leads and connecting wires. For modeling the voltage current curve of the nonlinear element, the curve given by the manufacturer is used. Previously in Section 5.2, it is shown that the current and voltage of the MOV matches this curve with an acceptable accuracy.

A nonlinear resistor can be modeled in different ways in the simulation environments. It can be modeled as a current-controlled voltage source or a voltage-controlled current source. It can also be modeled as a look up table. Any of these methods might be used in different situations. Here, since the manufacturer has provided the voltage versus current equation, it is easier to model the nonlinear resistor as a current-controlled voltage source. It is simulated in MATLAB Simulink and is shown in the figure below.

Chapter 5: MOVs in Variable Frequency Drives 140

Figure 5-31. MOV model simulated in MATLAB Simulink

Another possibility is to simulate the nonlinear characteristic as a voltage-controlled current source.

Sometimes, especially when the system is big and there are other voltage sources in the simulation, modeling the MOV as a current source have better solution convergence in simulation. To model an MOV as a voltage-controlled current source, it is needed to formulate the current in terms of the voltage. Existing equations 5-2 and 5-3 are not accurate to be used in the model. Therefore, we propose an equation that matches accurately to the voltage-current curve of the MOV, as it is shown below.

I = 10 ^ ( a1 + a2 × atan ( a3 * log(V) + a4 ) ) Equation 5-5

This equation can be fitted to MOV’s voltage-current characteristic, accurately from micro-amperes to the nominal discharge current in tens of thousands of amperes. The fitted curve is shown in Figure 5-32.

Figure 5-32. Curve fitting for voltage-controlled current source model

Chapter 5: MOVs in Variable Frequency Drives 141

The model parameters for current-controlled voltage source model and voltage-controlled current source model of 150V MOV is given in Table 5-3.

Table 5-3. 150V MOV’s VI curve parameters

Voltage -controlled current source Current-controlled voltage source

-log(I) log(I) I = 10 ^ [ a1 + a2 × atan (a3 × log(V) + a4)] V = 10 ^ [b1 + b2×log(I) + b3×e + b4×e ] a1 -0.8208119 b1 2.4620502 a2 -4.0238658 b2 0.0222331 a3 -9.4237785 b3 -0.0006945 a4 23.0242377 b4 0.0032469

The VFD system simulated in Section 3.5.6 is used here to verify the MOV simulation. Three MOVs are connected between lines and the ground on the motor terminals. The system is seen in Figure 5-33.

Figure 5-33. VFD system simulation with 150V MOV

Experimental measurement result is shown in Figure 5-26. It is seen that the installing 150V MOVs on the motor terminals reduces the line to ground overvoltages from about 380 V to about 320 V. In this condition, the MOVs conduct about 2.5 to 3 amperes of resistive current at the peak of the voltage waveform.

Simulation results are presented below. Figure 5-34 shows the motor terminal voltage and the MOV current. It is seen that the MOV conducts 2.7 A current at the peak of the voltage. It is also seen that the peak value of the line to ground voltage is 320 V.

Chapter 5: MOVs in Variable Frequency Drives 142

Figure 5-34. Simulation: 150V MOVs on motor terminal, line to ground voltage and MOV current

Simulation allows us to separate the MOV’s resistive and capacitive current, by adding current measurement inside the MOV model in the resistive and capacitive paths. These currents are shown in

Figure 5-35 alongside the line to ground voltage waveform. From Figure 5-35(a) it is seen that the resistive current is zero during the pulse, except at the first two voltage peaks.

(a) (b) Figure 5-35. Simulation: 150 MOV on motor terminal (a) resistive current (b) capacitive current

In summary, the simulation and lab measurements show that the reduction of the high frequency overvoltages is possible by installing MOVs between lines and grounds. However, the selected MOV should be able to dissipate the heat generated by the repetitive currents. Also, the capacitive currents of the

MOV, which are drawn from the line, should be considered in the overcurrent protection of the drive.

Furthermore, comparison of the simulation results to lab measurements demonstrate the accuracy and effectiveness of the simulations.

Chapter 5: MOVs in Variable Frequency Drives 143

5.5. MOV Application Solutions for VFD systems

VFD systems, like any other electrical system, needs protection against surges. One of the vulnerable locations can be the cable connecting the inverter to the motor, when it is outdoor. As discussed previously, the output of the inverter and the motor terminals are recommended locations for SPD installation. The problem is, the voltage applied to the connecting cable is a PWM voltage. When the cable is long, large high frequency overvoltages appear on the motor terminal. These overvoltages can discharge large repetitive currents through the MOV and cause it to fail due to the overheating. Aside from the overvoltages, it should be noted that MOVs are not tested for PWM voltages and there are no standard practices for the design engineers to select MOVs for systems with non-sinusoidal voltages such as VFDs.

The common traditional way of SPD design is choosing an MOV with a MCOV higher than the highest overvoltage in the point of the installation. Considering that the pulse amplitude is usually much higher than motor’s nominal voltage in an inverter, engineers have to install MOVs with high voltages to avoid failure. Higher MCOV results in a higher clamping voltage and hence, a compromised protection. For example, the 220 V motor in the lab is supplied with an inverter with 340 V dc link. To avoid the highest reflected overvoltages, at least a factor of 2x should be considered. It puts the overvoltage level at 680 V. A proper MOV installation for this situation would be a 550V MOV with a maximum operating dc voltage of

745 V. With this MOV, for the nominal discharge current of 40 kA, 8/20 µs standard waveform, the clamping voltage would be higher than 3 kV.

To reduce the protection level to an acceptable range, while avoiding the increased leakage current, some manufacturers propose a Gas Discharge Tube (GDT) to be installed in series with the MOVs. In this section, we are going to study this approach and discuss its advantages and disadvantages. Furthermore, we propose two other solutions for this application: TSPD-Switched MOVs (TSMOV) and Reconfigurable

Surge Protective Device (RSPD). We will investigate the advantages and disadvantages of each method through theoretical analysis and simulation studies.

5.5.1. MOV in series with GDT

Chapter 5: MOVs in Variable Frequency Drives 144

GDTs, evolved from traditional Spark Gap, is a commonly used protective device, mainly as the primary surge protection. It can be used in various surge protection designs with MOVs or TVS diodes. As mentioned above, one possible solution for MOV application in VFD systems and preventing it from heating due to the repetitive high leakage currents is adding a GDT in series. In this section, we discuss different aspects of this solution and compare it to the base solution of installing an MOV with a higher

MCOV, or two MOVs in series. Before that, we briefly review GDTs structure, function, advantages and disadvantages.

5.5.1.1. Gas Discharge Tube (GDT)

A GDT consists of two electrodes in a sealed tube filled with an inert gas. The distance between the electrodes, the nature of the gas, and the pressure inside the tube are all controlled to permit a discharge above a desired voltage. This device is connected in parallel with the protected load, similar to an MOV.

The GDT maintains a high impedance in the OFF state until its voltage exceeds the device’s sparkover or breakdown voltage. Once the voltage goes beyond this point, the gas molecules become ionized and conduct electricity. During the discharge, GDT is in low-impedance state and the voltage across it is low

(similar to a short circuit), protecting the load against the incoming surge. Once a GDT is in the arc mode, it will only stop conducting when the available voltage is below a certain holdover level. In that conditions, the ions recombine to form molecules and the insulation of the gas is restored. A typical voltage-current curve of a GDT is shown in Figure 5-36.

Figure 5-36. Typical voltage current characteristic of a Gas Discharge Tube [124]

Chapter 5: MOVs in Variable Frequency Drives 145

GDTs have low capacitance and leakage current, hence they have minimal effect on normal operation of the system. On the downside, GDTs are known for their slow response. In other words, the transition time between high impedance and low impedance states is long (depending on the voltage ramp rate and amplitude can be up to several tens of microseconds). This might be a serious issue in protecting against fast transients, like the high frequency overvoltages in PWM inverters. Moreover, GDTs in general have a short lifespan and can withstand a limited number of surges.

Another major concern about installing the GDTs for the protection of sensitive loads is that the impulse breakdown voltage (also called sparkover voltage or trigger voltage) is dependent on the voltage change rate of the pulse. The breakdown voltage response of a GDT to transients with ramp rates of 1 V/µs or less is referred to as the DC breakdown voltage level. Due to the nature of the gas, the same GDT will experience breakdown at a higher voltage as the transient’s ramp rate increases. For example, a specific product with the DC breakdown voltage of 300 V, has a typical value of impulse breakdown voltage of 550

V at 100 V/µs and 800 V at 1 kV/µs [125]. Another disadvantage of the GDT is its inability to stop the short circuit current after the trigger. The current continues until the voltage decreases to near zero. That is why they are mainly used in ac systems or in combination with the other devices, such as in series with an

MOV, as mentioned above.

GDTs and MOVs might be connected in different ways. Here, we focus on the series connection of these two components that might be advertised as a leakage-free SPD. Such a combination is shown in

Figure 5-37. As it is obvious from its name, the obvious advantage is that the leakage current is almost non- existent. This can protect the MOV from conducting repetitive currents and overheating. Furthermore, having an MOV in series with the GDT gives it the ability to stop the current after the breakdown.

However, this hybrid products inherits some of the negative aspects of the GDT. Slow transition to the ON state can make it ineffective in protection against fast transients.

When considering a GDT to be connected in series with an MOV as a hybrid SPD, it is important to select the breakdown voltage of the GDT so that it does not trigger for the repetitive high frequency overvoltages. The role of the GDT here is to protect the MOV by not triggering in low repetitive spikes.

Chapter 5: MOVs in Variable Frequency Drives 146

Figure 5-37. A GDT in series with an MOV

To investigate further, a simple model is developed and simulated in MATLAB Simulink. Figure 5-39 and Figure 5-40 show the model simulated and how it is connected to the surge generator. To simulate the constant voltage of the GDT in the arc state, a Zener diode is used in the model. A delay module is used in to represent the intrinsic transition delay of the GDT. The transition delay of a GDT is dependent on the applied voltage and current and can vary based on the conditions [126]. Figure 5-38 shows the impulse breakdown voltage of a GDT with DC breakdown voltage of 350 V, surged with generator open circuit voltages with different peak values. Orange points show the breakdown moments for a GDT in different tries. Blue curves are generator’s open circuit voltages, plotted to provide a point of reference. It is seen that when a 400 V, 10/1000 µs surge is applied, the transition delay can be a few tens of microseconds.

When the surge voltage is increased, the transition delay becomes more consistent, but the impulse breakdown voltage varies more. According to its datasheet, the typical impulse breakdown voltage for this

GDT is 875 V for 1000 V/µs. For higher ramp rates, the impulse breakdown voltage might be even higher.

In the simulations, the breakdown voltage and the transition delay are set to typical values based on the applied voltage and according to Figure 5-38.

Figure 5-38. GDT transition delay [127]

Chapter 5: MOVs in Variable Frequency Drives 147

Figure 5-39. Simple GDT model

Figure 5-40. Simulation: surge application to the GDT

The simulation results are shown in Figure 5-41. Three surges are applied to the GDT with surge generator open circuit voltages of 400, 500 and 1000 volts. For the 400-volt surge, the transition delay can be long. For the surges below 400 V, GDT does not trigger, even though the DC breakdown voltage is 350

V. For faster surges, this value can be higher. It should be noted that the let-through voltage of a GDT is dependent on the voltage waveform. This means for the standard 8/20 µs current surge that has a 1.2/50 µs voltage waveform, the spark is fast, but the let-through voltage of a GDT is much higher than that for an

MOV. As it is seen in Figure 5-42, the maximum let-through voltage of the GDT is 625 V, while this value for a comparable MOV is 400 V.

(c) OC = 1000 V

(b) OC = 500 V

(a) OC = 400 V

Figure 5-41. 350V GDT’s impulse breakdown voltage and transition delay for different surge generator voltages

Chapter 5: MOVs in Variable Frequency Drives 148

(a) (b) Figure 5-42. 2 kV, 1.2/50 µs voltage surge (1 kA 8/20 µs current) (a) MOV (b) GDT

5.5.1.2. Simulation Results

Assume that the SPD is installed on the motor terminals of a motor, similar to the one in the lab. The line to ground voltage of the motor is seen in Chapter 3, Figure 3-48. The voltage pulse starts from -170 V and rises to 170 V, and the oscillations bump the voltage peak to 400 V. The goal is that the designed SPD should have a low leakage current in 400 V, i.e. GDT should not trigger at the peak voltage. When a GDT is connected in series with an MOV, it tolerates a large fraction of the voltage in the OFF state, since it is an open circuit with a much smaller capacitance. Thus, the GDT should be selected to bear all the voltage, here 400 V. Since the MOV does not tolerate a big voltage in the normal condition, selecting an MOV with a lower MCOV can achieve an overall lower clamping voltage. However, after the GDT is triggered, the line voltage is applied to the MOV. The MOV should be able to withstand the full voltage at least until the next current zero-crossing occurs, where the GDT goes back to the OFF state. Selecting the best MCOV would be a compromise between the reliability and the performance. Here, it is assumed that the GDT is connected in series with a 150V MOV. For the comparison with an MOV with higher MCOV, the results are compared to those from two 150V MOVs connected in series, representing a 300V MOV.

Figure 5-43 shows the simulation of an MOV and a GDT in series. Simulation results show this hybrid

SPD is effective for the slow surges where the GDT response time is not an issue as well as for the large surges, where the clamping voltage of the MOV can be high.

Chapter 5: MOVs in Variable Frequency Drives 149

Figure 5-43. Simulation of MOV and GDT in series

(a) (b) Figure 5-44. 40 kA surge with 1kV/µs ramp, applied to (a) MOV-GDT (b) MOV-MOV in series

(a) (b) Figure 5-45. 1kA surge with 1kV/µs ramp, applied to (a) MOV-GDT (b) MOV-MOV in series

Figure 5-44 shows the response of the series MOV-GDT for a 40kA surge with a voltage ramp of 1 kV/µs. The response of the GDT depends on the voltage change rate. At this rate, the impulse breakdown voltage for the GDT is 925 V, based on the datasheet. After the GDT triggers, the voltage drops to MOV voltage level. The load is subjected to the high voltage only for a very short time, and then voltage drops almost to half. For the MOV with higher MCOV, or two MOVs in series, the residual voltage is dependent

Chapter 5: MOVs in Variable Frequency Drives 150

on the amplitude of the current. For this 40kA current surge, voltage surpasses 1500 volts and remains high for the entire surge duration. For sensitive loads with low thermal capacity, this can have adverse effects and might cause a failure.

For smaller surges with 1 kV/µs ramp, such as the one in the simulation that is shown in Figure 5-45, two MOVs in series has a better response. Although the voltage remains around 800 V for all the surge duration, the voltage peak is smaller than that for the series MOV and GDT. For this surge, GDT triggers in

925 V. After an initial peak, the voltage decreases to about 400 V. This shows that the MOV GDT combination, exposes the load to higher voltages for the smaller surges. Considering the abundance of the smaller surges, this can be an important factor in the SPD selection.

The hybrid SPD of an MOV and a GDT in series has other limitations, too. It cannot be customized for the need by connecting two in parallel. Installing MOVs in parallel is a common practice in industry to achieve higher surge current handling capability. Even though MOVs do not share the surge current equally, for large surges this sharing is acceptable in most cases. On the other hand, adding a GDT in series limits this option. If two GDTs are installed in parallel, once one of them triggers, the voltage will drop and will prevent the other one from triggering.

5.5.2. TSPD-Switched MOVs (TSMOV)

The Controllable Metal-Oxide Arrester (CMOA) and Thyristor-Switched Arrester have been proposed in the past for suppression of switching overvoltages in HV and UHV power systems [128], [129]. It is basically a metal-oxide arrester which is capable of changing the number of varistor blocks by shorting out a part of the arrester with a power electronic switch in order to maintain a low residual voltage during an overvoltage. The switch that is used is usually an anti-parallel thyristor which is controlled appropriately by gate triggering. The voltage at the protected point is monitored, and if an overvoltage is detected, a gate trigger command is sent to power electronic switch to short a part of the arrester and make the clamping voltage less. Lower overvoltage level allows to use lower insulation level and reduce the cost of the equipment.

Chapter 5: MOVs in Variable Frequency Drives 151

Figure 5-46. TSMOV configuration

The same concept can be used for surge protection purposes, too. The only difference is that the switch has to be fast enough to act during the surge overvoltage. Based on this concept, we propose to use a

Thyristor Surge Protective Device (TSPD) for reconfiguration of the MOVs in surge application. The proposed SPD is shown in Figure 5-46. In this configuration, in normal conditions, both MOVs are in the circuit, providing a higher MCOV to avoid overheating of the MOVs due to overvoltages. When a surge occurs, MOVs reduce the voltage to some extent. When the voltage across MOV2 reaches the Breakover voltage of the TSPD, it turns on and shorts out the MOV2, leaving only MOV1 in the circuit. This way, the clamping voltage seen by the protected load is equal to the voltage of only one of the MOVs. The voltage of the TSPD in the On state is only a few volts.

Although a TSPD works similar to a GDT, and in some applications, they might be interchangeable, they have practical differences. For surge protective purposes, TSMOV has several advantages over the

MOV-GDT solution. First, MOVs are connected all the time and the protection is achieved even for small surges. The second advantage is that TSPD’s switching is more deterministic compared to a GDT. For a

GDT, even though the typical value is given in the datasheet, the impulse breakdown voltage has a random distribution with a noticeable standard deviation. TSPD, being a semiconductor device, is more predictable in applications that need a certain accuracy. Another important advantage of the TSPD over a GDT is its long lifetime. A GDT has a limited number of surges defined as its lifetime, while a TSPD, if properly designed, has technically unlimited lifetime (service life in excess of 20 years is typical [130]). In this section, after a brief introduction of the TSPD device, we will discuss the benefits of the proposed method through simulation results.

Chapter 5: MOVs in Variable Frequency Drives 152

5.5.2.1. Thyristor Surge Protective Device (TSPD)

Thyristor Surge Protective Device (TSPD) is a bidirectional1 thyristor device designed to protect loads from lightning and switching over-voltages. The basic thyristor is a NPNP semiconductor device that has three PN junctions. TSPD is equivalent of a PNP transistor and an NPN transistor connected as a regenerative pair. It is like a voltage triggered Triac without a gate and operates like a spark gap. It remains non-conducting until the applied voltage meets or exceeds its rated breakover voltage. Once entering this conductive state, it continues to conduct, regardless of voltage, until the applied current falls below its rated holding current. The V-I characteristic of a typical TSPD is shown in Figure 5-47.

Figure 5-47. TSPD voltage current characteristic [130]

Figure 5-48. Switching characteristic of a TSPD

Important parameters of a TSPD are depicted in Figure 5-48. TSPD remains in the Off state as long as its voltage is below the rated repetitive peak off-state voltage (VDRM). At this voltage, the current is called repetitive peak off-state current (IDRM). The breakdown region is the high-voltage portion of the voltage- current characteristic with a low dynamic resistance. Depending on the thyristor design and temperature, the end of the breakdown region may be at a higher or lower voltage than the start. The maximum voltage that occurs in the breakdown region is defined as the breakover voltage, VBO. Additional measurements

1 There are various types of TSPDs, including unidirectional TSPDs that are not covered here.

Chapter 5: MOVs in Variable Frequency Drives 153

may be made of the voltage and current at the switching point (VS, IS). The On-state region is the low- resistance high-current portion of the voltage-current characteristic. In the On-state condition, TSPD shows the minimum voltage drop for the current flowing. The minimum current that will just maintain the On- state condition is defined as the holding current, IH. Currents below this value will cause the thyristor to switch off.

TSPD is a crowbar protective device with an on-state voltage of a few volts. Generally, a TSPD’s VBO is 20% to 30% greater than its VDRM. Thus, in this respect it provides better overvoltage protection compared to a GDT that its typical impulse breakdown voltage can be 150% or more higher than its DC breakdown voltage. TSPDs have a relatively low energy handling capability and temperature sensitive characteristics. However, as the majority of the diverted current is conducted in a low-voltage condition, their current capability is high. The peak impulse let-though voltage will increase as the impulse rate-of- rise increases. When used within their rated values, these components will have a high life expectancy

[130]. Exceeding device ratings usually results in catastrophic failure.

The junctions, formed by the device NPNP structure, are the principle elements that define the device capacitance. The capacitance of a PN junction will depend on the applied dc bias, VD, ac test signal level,

Vd, and junction temperature, TJ. At low levels of dc bias, where the ac test level is significant compared to the dc bias, the capacitance is strongly dependent on the value of ac test level. The effective capacitance value decreases as the ac test level increases [130].

5.5.2.2. Simulation Results

Similar to the previous section, we consider a VFD system with DC link voltage of 340 V for the simulations. The SPD is assumed to be installed between lines and ground. The maximum voltage for the line to ground voltage is 400 V for the system. Since the two MOVs are dividing the voltage, each MOV’s voltage will not see higher than 200 V. So, the TSPD should not conduct for the normal voltages up to 200

V. A TSPD with the VDRM of 230 V is selected for this situation. The proposed device is simulated in Orcad

Capture software. The PSpice model of the TSPD is provided by the manufacturer. The maximum dynamic breakover voltage is 295 V for this device, given in the datasheet [131].

Chapter 5: MOVs in Variable Frequency Drives 154

Figure 5-49. Simulation: applying surge to a TSPD

5.0KA

2.5KA

SEL>> 0A I(U9:PIN1) 300V

200V

100V

0V 0s 2us 4us 6us 8us 10us 12us 14us 16us 18us 20us V(U9:PIN1) Time Figure 5-50. Simulation: 2/10 µs surge applied to a 230V TSPD and its voltage response

Figure 5-49 shows the ORCAD Capture simulation of the above-mentioned TSPD. A 5 kA, 2/10 µs1 surge is applied to the 230V TSPD. It is seen from the voltage waveform that the maximum voltage is

290V. Also, a GDT-like function is seen from the waveforms. The device is in the non-conducting state for low voltages. As the voltage reaches the VBO, the device transitions to the low-impedance state and effectively becomes a short circuit with a low voltage drop.

(a) (b) Figure 5-51. Simulation: (a) proposed TSMOV (b) Series MOVs for comparison

1 2/10 µs current surge with 2/10 µs open circuit voltage waveform

Chapter 5: MOVs in Variable Frequency Drives 155

1.0KV 5.0KA 1.0KV 5.0KA 1 2 1 2

0.8KV 4.0KA 0.8KV 4.0KA

0.6KV 3.0KA 0.6KV 3.0KA

0.4KV 2.0KA 0.4KV 2.0KA

0.2KV 1.0KA 0.2KV 1.0KA

>> >> 0V 0A 0V 0A 0s 5us 10us 15us 20us 0s 5us 10us 15us 20us 1 V(MOV2:1) 2 I(Rsurge) 1 V(MOV2:1) 2 I(Rsurge) Time Time (a) (b) Figure 5-52. Simulated surge and voltage response: (a) proposed TSMOV (b) Series MOVs for comparison

The proposed TSMOV device is seen in Figure 5-51(a). The TSPD is connected in parallel with MOV2.

This MOV will be shorted out during the surge. MOV1 is connected in series and remains in the surge path all the time. The simulation result is shown in Figure 5-52(a). The simulation result of the 2 MOVs in series, also, is given in the same figure for comparison. Since the TSPD’s capacitance is much smaller than the MOV’s capacitance, it does not affect the voltage division between MOVs. TSPD’s capacitance is 65 pF, compared to 4800 pF nominal capacitance of the MOV, has almost no effect. Before TSPD triggers, the voltage is divided equally among the MOVs (neglecting the MOVs tolerance). Once the TSPD switches to

On state, MOV2 is shorted out and the let-through voltage will be equal to only MOV1’s voltage. For this example, the maximum let-through voltage is 745 V, but after a sub-microsecond transition time, voltage drops to around 500 V for the rest of the surge duration. The maximum voltage on MOV2 is the switching voltage of the TSPD, which is 290V.

For the alternative solution, two MOVs in series, the peak of the let-through voltage is 956 V. This voltage remains high for the duration of the surge. This is especially important, because generally, the voltage tolerance capability is higher for shorter overvoltages. For some applications, the let-through voltage waveform may be just as important as its peak value. They require the length of the overvoltage to be shorter than a given time, depending on its peak value. Above example shows that TSMOV, not only reduces the peak value of the let-through voltage, but also reduces the time duration of the highest part of the overvoltage, considerably.

Chapter 5: MOVs in Variable Frequency Drives 156

Simulation result shows the effectiveness of the proposed method in reducing the let-through voltage, without the shortcomings of the GDT-MOV solution. There are also some similarities. Both SPD’s clamping voltage peak is dependent on the surge ramp rate. However, for TSPD the range of the change in

VBO is smaller than the change in impulse breakdown voltage of the GDT. In summary, the proposed

TSMOV has better performance, longer lifetime, higher reliability than GDT-MOV device. The main disadvantages are higher price and lower current handling capability of the TSPD, compared to the GDT based solution. However, being a semiconductor device, TSPDs are expected to have higher surge current capability and lower price with advancement in fabrication technology.

5.5.3. Reconfigurable Surge Protective Device (RSPD)

Every surge protective device has two states: stand-by state (OFF) and conducting state (ON). In stand- by state, it is ideal to have zero leakage current and zero capacitance, to have no impact on the normal function of the system. When a surge occurs, the SPD enters the conducting state, in which the idea is to prevent the overvoltage and keep the voltage of the system in the same level as before. However, usually a given parameter has opposite impacts on these two. For example, an MOV with lower MCOV will provide a lower clamping voltage, but at the same time will have higher leakage current. This is true for two MOVs in series as well. Installing MOVs in series will decrease the leakage current, but it will increase the let- through voltage. Considering two MOVs, an ideal connection for the stand-by state is series connection to reduce the leakage current, while the ideal connection for the conducting state is parallel connection to have lowered let-through voltage and increased current handling capability.

One way to have two MOVs in series for one state and the same MOVs in parallel for another state is reconfiguring the SPD. Figure 5-53 shows one such a configuration. This proposed Reconfigurable Surge

Protective Device (RSPD) has two MOVs that can be connected either in series or in parallel.

Reconfiguration is achieved by using three switches in the SPD.

Chapter 5: MOVs in Variable Frequency Drives 157

Table 5-4. State of the switches for RSPD

State \ Switch S1 S2 S3 MOVs in series OFF OFF ON MOVs in parallel ON ON OFF

Figure 5-53. Proposed RSPD

When MOV is in stand-by mode, switches S1 and S2 are open and S3 is closed, connecting MOVs in series between two lines. In occurrence of an overvoltage event, switch S3 closes and switches S1 and S2 close, connecting the MOVs in parallel between the lines. Table 5-4 summarizes the states of the switches for both stand-by and conducting modes. Figure 5-54 shows the configuration of the SPD in each state.

(a) Stand-by (b) Conducting Figure 5-54. State of switches for reconfiguration

RSPD provides benefits over other SPDs discussed before. Compared to TSMOV, RSPD can provide lower let-through voltage, since the current through each MOV is half of that for TSMOV. In TSMOV, the switched MOV that is shorted out during the surge, experiences less degradation, while the base MOV that stays in the circuit during the surge degrades more severely. This un-even degradation can change the

Chapter 5: MOVs in Variable Frequency Drives 158

voltage division ratio for the MOVs. In RSPD both MOVs experience similar surges and system voltages, thus their degradation levels are more likely to be close to each other.

Decision to turn on or off the switches can be made based on the voltage amplitude of the SPD, which is equal to the load voltage. Depending on the maximum desired voltage on the load, a simple comparison- based circuit can be used to trigger the switches.

In order to avoid creating a voltage pulse due to disconnecting the surge current, switches should be triggered in order. In normal operating condition, S3 is closed (ON), and the other two switches are open

(OFF). In the advent of the surge, S3 will carry the surge current, when the surge is still small. Once the voltage reaches the threshold value, S1 triggers first short-circuiting MOV2 and S3. After S1 closes, S3 opens. After that, S2 closes, providing the second surge current path through MOV2, in parallel with

MOV1.

5.5.3.1. Simulation Results with Ideal Switches

Figure 5-55 shows simulation of the concept of the RSPD in Matlab Simulink. 150 V MOV model built in previous sections is used here, with ideal switches. Trigger decision of the switches are made based on the load voltage, seen as “Reconfig Ctrl” unit in the figure. This unit, compares the load voltage to a reference value, here 400 V. When voltage reaches the threshold value, it provides command signals to the gates of the switches with a small delay, as it is discussed above.

Figure 5-55. Simulation of RSPD with ideal switches

Chapter 5: MOVs in Variable Frequency Drives 159

(a) (b)

Figure 5-56. Simulation results for (a) RSPD, compared to (b) two MOVs in series

Figure 5-57. Simulation results, switching moment of the MOVs at 400V from series to parallel

Simulation results demonstrate the effectiveness of the RSPD. In Figure 5-56, let-through voltage of the proposed RSDP is compared with that of two MOVs in series. In normal conditions of the system, both of these two configurations would have low leakage current, since both have two MOVs in series. Under surge condition, two MOVs in series has higher let-through voltage. As it is seen from the figure, for a 5 kA 8/20 µs standard surge, the let-through voltage reaches 1000 volts. For the same surge, RSPD provides the protection with the let-through voltage of 453 volts. This is a substantial improvement by utilizing reconfiguration of the existing MOVs under surge condition.

Figure 5-57 shows the voltage when MOVs are switching from series to parallel connection. The threshold is set to trigger when voltage reaches 400 V. When MOVs is shorted out by S1, voltage drops to half of 400V, i.e. 200V instantly. Then, after a short delay, S2 reconnects MOV2, this time in parallel with

Chapter 5: MOVs in Variable Frequency Drives 160

MOV1. MOVs share the surge current, so their voltages decrease further. As it is seen from the figure, voltage decreases to 160V, and then increased again according to MOVs current-voltage characteristic.

Summary of the results is presented in Table 5-5.

Table 5-5. Let-through voltage reduction by RSPD for 8/20 µs surge Surge current Max voltage (V) Improvement (%) peak (kA) RSPD Two MOVs in series 5 453 1000 54.7 10 502 1142 56.0 20 576 1364 57.7 40 692 1725 59.8

5.5.3.2. Switches in RSPD

Switches in RSPD can be discussed in two aspects: switch type and its surge current handling capability. Since these switches (two of them: S1 and S2) are in the surge current path, they need to conduct the surge current without considerable voltage drop on them. They also need to have high surge current handling capability. Another important factor is the switching delay. The overall reaction time from the advent of the surge until having the first MOV on should be acceptably short. A part of this reaction time is the control circuit delay, which in case of using a simple analog comparison circuit is short. Thus, the turn on delay of the switch should be short enough to be able to turn on before surge voltage reaches a damaging level.

Considering the requirements of the switches in the RSPD, insulated-gate bipolar transistor (IGBT) seems to be a good choice. An IGBT is a three terminal, four-layer semiconductor device (PNPN) that works as a unidirectional switch. In a simple view, an IGBT is a combination of a MOSFET and a bipolar junction transistor (BJT) to achieve high current handling capability of BJTs while maintaining the simple gate drive requirements of MOSFETs. Figure 5-58 shows IGBT’s cross-section and its internal connection as a MOSFET and a BJT, as well as its circuit symbol.

Chapter 5: MOVs in Variable Frequency Drives 161

Figure 5-58. IGBT: insulated-gate bipolar transistor

The main advantages of an IGBT over a Power MOSFET and a BJT are:

1. It has a low on-state voltage drop and superior on-state current capability.

2. Low driving power and a simple drive circuit due to the input MOS gate structure.

3. It has high reverse blocking capabilities.

The main drawbacks of IGBTs are:

1. Switching speed is inferior to that of a Power MOSFET and superior to that of a BJT. The

collector current tailing causes the turnoff speed to be slow.

2. IGBT only conducts in one direction. It does not have a body diode, and usually a reverse

conducting diode should be added to the module.

Early generations of the IGBT also had other issues, such as being prone to latch up or having a negative temperature coefficient, which could lead to thermal runaway and makes the paralleling of devices difficult. However, nowadays these issues are addressed and resolved in present generations of IGBTs

[132].

One aspect of selecting a switch for a surge protection application is the fact that surges can occur in positive or negative polarity. Thus, the switch should have the capability of conducting the current in both directions. An IGBT cannot conduct current in the reverse direction (from emitter to collector). This is unlike a power MOSFET, which can conduct currents in either direction (drain to source or source to drain). Also, unlike MOSFET, an IGBT does not have an intrinsic body diode that allows the current to flow in reverse direction, when the device is off. Therefore, in most applications, a fast recovery diode is connected in anti-parallel with the IGBT, as shown in Figure 5-59(a).

Chapter 5: MOVs in Variable Frequency Drives 162

(a) (b)

Figure 5-59. IGBT (a) anti-parallel diode (b) bidirectional conducting

Another aspect of selecting switch for surge protection application is its blocking characteristic. The switch should be able to block in both directions. Although an IGBT has capability of blocking forward and reverse polarity voltages, IGBTs with anti-parallel diodes cannot block the reverse polarity voltages.

Therefore, to achieve both bidirectional conducting and bidirectional blocking, it is needed to use two

IGBTs in series, as it is shown in Figure 5-59(b). When surge flows through a bidirectional switch of this type (two IGBTs in series), the total voltage drop on the device would be sum of the forward conducting voltage of one IGBT and forward conducting voltage of one anti-parallel diode. For an acceptable switch for RSPD, this voltage should be considerably less than MOV’s voltage during the surge.

Behavior of electronic switches, among them IGBTs, are studied in literature [133]–[139]. However, in most cases, these papers investigated short circuit current handling capability and do not focus on lightning surges. Since electronic switches are not mainly designed for conducting lightning surge currents, little information is available on their surge current handling capability using 8/20 µs or other standard waveforms in microsecond ranges.

Reference [135] tested IGBT’s capability in conducting 10 ms half sine shape surge current pulse and shows that with a higher gate voltage, it can conduct 20 times the rated current without failure. If we can assume the same I2t for a 20 µs surge current, the switch would be able to handle 8/20 µs surges of 220 times its rated current. For an IGBT with 100 A rated current, the surge handling capability would succeed

20 kA. If the switch is optimized for surge current handling, it may be able to conduct even higher currents.

Figure 5-60 shows IGBT’s voltage during the half sine wave current surge from [135]. IGBT’s voltage peak reaches 20 V for a current 15 times its rated current.

Chapter 5: MOVs in Variable Frequency Drives 163

(a) (b)

Figure 5-60. IGBT characteristics during the surge (a) Voltage (b) IC-VCE [135]

As it is stated previously, in a bidirectional IGBT, shown in Figure 5-59, the fast recovery diode’s voltage, also, needs to be small for the surges. Nowadays, fast recovery diodes can conduct high surge currents with a small voltage drop. Figure 5-61 shows forward voltage of the IGBT’s anti-parallel diode during an 80 kA, 100 µs surge current, for a 1700A IGBT module. As it is seen, the peak voltage is less than 20 V during the peak voltage. This high surge current handline capability is achieved through well balanced parallelization of diode inside the module. Most of the switches are designed to have positive temperature coefficient at rated current and above, resulting in homogeneous current flow through diodes.

Figure 5-61. Fast recovery diode: forward voltage vs surge current (80 kA, 100 µs) [133]

Although there is still little published on surge current capability of electronic switches, above examples show that it is possible to achieve high surge current handling capability that can be used in RSPD.

Chapter 5: MOVs in Variable Frequency Drives 164

5.6. Summary and Conclusions

Generally, MOVs are designed to conduct occasional surges in the power systems with sinusoidal voltage waveform. However, in new electrical environment, sometimes engineers need to install surge protection devices for a system with non-conventional voltage. VFD systems, as an example, use PWM voltage over a cable connecting the inverter to the motor that needs to be protected against surges. In such conditions, repetitive overvoltages created on motor terminal can conduct large repetitive currents through

MOV and cause its failure. In this chapter, we studied MOV’s behavior under repetitive surges from a voltage supply as well as PWM pulses. It is discussed how installing MOV on a VFD system can affect the wave shape of overvoltages. Also, we studied MOV’s high frequency model and simulated the MOV used in the tests and compared the simulated waveforms to the measured ones from the lab. Furthermore, we demonstrated MOVs can reduce the high frequency overvoltages in VFD systems. However, in this condition MOVs conduct resistive currents that can raise the MOVs temperature. Further investigation of this application is suggested as a potential future research.

Furthermore, in this chapter, we proposed two new approaches for installing MOVs on PWM lines to prevent MOV failure due to high leakage current, while maintaining low protection level during the surges.

These two methods, TSPD-Switched MOVs (TSMOV) and Reconfigurable Surge Protective Device

(RSPD) are simulated and compared to existing methods of installing two MOVs in series or connecting a

GDT in series with MOVs. Through simulation studies, effectiveness of the proposed methods is shown, and their advantages and shortcomings are discussed in detail. Comparisons show the proposed method have superior performance over the existing methods.

Chapter 6: Conclusions and Future Work 165

Chapter 6:

6. Conclusions and Future Work

The research in this thesis focuses on the topic of surge protection. New methods for the lifetime estimation of Metal Oxide Varistors (MOVs) are proposed, and new approaches for surge protection of

Variable Frequency Drives (VFDs) are presented and experimentally validated. The thesis first a thorough review the failure modes of MOVs and the influential factors for their degradation are explained. Then multiple experiments with a large number of MOVs are conducted to better model and explain the Energy

Absorption Capability (EAC) of both new and degraded low voltage varistors. This represents the first extensive study on the impact of EAC degradation for low voltage MOVs. Furthermore, a new method is proposed to estimate the remaining lifetime of the MOVs.

The rest of the thesis focuses on the surge protection of the VFD systems. With the increasing importance of the energy efficiency in today’s power system, VFD applications are expanding rapidly. As a main issue in their surge protection, this thesis investigates, in detail, the problem of high frequency overvoltages that are created on the motor terminal due to the voltage reflection phenomenon in long cables. The MOV is typically built for conducting occasional surge currents, where it has enough time to cool down after a surge event. The high frequency overvoltages can conduct high repetitive currents through the MOV and cause its failure because the cool down period is eliminated. A comprehensive

Chapter 6: Conclusions and Future Work 166

literature review is carried out on the causes, effects and the mitigation of these overvoltages. Qualitative explanations of the failures, supported by analytic models, are presented.

Besides the theoretical work, an experimental VFD system is built in the lab at Northeastern University.

This setup is comprised of an induction motor controlled by a VFD. Motor shaft is coupled to a dynamometer as a mechanical load. As the controller, a programmable motor control development board is used. The cable and the motor models are extracted for the experimental setup, and the system is simulated.

Moreover, MOVs behavior are studied under pulse voltages. At the end, two new approaches are proposed for installing MOVs safely on the VFD systems, that provide superior performance over the existing methods. In this chapter, the results and the conclusions of this thesis are summarized, and some novel ideas are suggested for the future work.

6.1. Summary of Results

The main research contributions of this thesis are summarized below, and are repeated from previous chapters:

In Chapter 1, a literature review is conducted on the failure modes of the MOVs. Thermal runaway, puncture, cracking and flashover are identified as main failure modes of the MOVs. The effective factors are the surge magnitude and duration, MOV’s microstructural non-uniformity, size, coating and contacts.

In Chapter 2, experimental results on new and degraded low voltage MOVs are presented, and their

EAC and Time to Failure (TtF) are analyzed. These experiments showed that average EAC of MOVs decrease with surge degradation. It is also found that unipolar surges decrease EAC more than bipolar surges. Also, the results reveal that degraded MOVs conduct lower currents during a transient overvoltage

(TOV). This fact leads to the conclusion that surge degraded MOVs have longer average TtF than new

MOVs for a given TOV. Finally, the experimental tests performed on different MOV’s showed that some commercial products might have very similar nominal specifications, yet they have a significant difference in their surge capabilities. In addition to the test results, a new health monitoring algorithm for low voltage

MOVs is proposed. The proposed algorithm utilizes takes the MOV’s voltage and current waveforms and its temperature as inputs and then estimates its remaining lifetime.

Chapter 6: Conclusions and Future Work 167

Chapter 3 describes the voltage reflection phenomenon in a VFD system and discusses the influential factors on overvoltage in VFD. It is shown that overvoltage values are mainly dependent on the pulse rise time and the cable length. Mathematical calculations demonstrate that if the voltage pulse rise time is shorter than twice the traveling time, a full reflection will occur. In such cases, the motor terminal voltage will reach almost double the voltage of the DC link. Furthermore, in this chapter, an experimental VFD system is used to validate the analytic analysis. The detailed models for the cable and the motor are extracted from the lab test-bench, and the system is simulated. Simulation results demonstrated that not always increasing the rise time can lead to the decreased overvoltage, because of the second and subsequent voltage wave reflections. The actual relation is presented through simulations.

Chapter 4 presents a thorough literature review on the overvoltage mitigation techniques in VFD systems, and advantages and disadvantages of each method is discussed. The main approach to mitigate the overvoltages in the VFD systems is using passive or active filters. These filters can be installed on the motor terminals or the inverter output, or even both. Other mitigation techniques include PWM control, using a dc cable between the rectifier and inverter instead of an ac cable between inverter and motor, and installing high energy varistors on the motor terminal.

Chapter 5 reports the experimental findings on MOV behavior under repetitive pulses. It is shown that installing MOVs at the motor terminal can reduce the overvoltages created by voltage wave reflection. It is also shown that the resistive current during a pulse can behave in different ways. It can always be increasing, or it can increase in the beginning of the response and then decrease. The response depends on on several factors, including inductance of the ceramic, the waveform of the voltage pulse, MOV’s microstructure, and the heating of the MOV. Conducting resistive current heats up the MOV. Thus, it is necessary to make sure the average power loss does not exceed the power dissipation capability of the

MOV. For such applications, designers need to know the highest current that an MOV can draw for a specific voltage. So, it is suggested that the manufacturers provide the lower limits of voltage-current characteristic.

Furthermore, two new approaches, TSPD-Switched MOVs (TSMOV) and Reconfigurable Surge

Protective Device (RSPD), are proposed in Chapter 5. The proposed methods prevent MOV failure due to

Chapter 6: Conclusions and Future Work 168

high leakage currents, while maintaining low protection level during the surges. Effectiveness of the proposed methods are demonstrated through simulation studies. The results show the proposed methods have superior performances over the existing methods.

TSMOV is comprised of two MOVs in series, where one of them is connected in parallel with a TSPD.

During the normal condition, the two MOVs are connected in series, reducing the leakage current. Under the surge condition, TSPD is shorting out one of the MOVs, providing lower clamping voltage. In summary, the proposed TSMOV has better performance, longer lifetime, higher reliability compared to the existing method of series GDT-MOV device.

Similar to the TSMOV, RSPD has two MOVs, where it can reconfigure them into parallel or series connection. This allows MOVs to be connected in series under normal voltage, and in parallel during the surge occurrences. The possibility and effectiveness of the RSPD is demonstrated through simulation.

6.2. Suggestions for Future research

Below are a few ideas for a possible future research that might build upon the research of this thesis.

• Near-failure MOV detection using machine learning

MOVs are known to degrade over time when they experience surges and transient overvoltages.

Usually, to avoid fire hazard and damages to the system, they are thermally protected. This means that once they reach their end of life, they fail and get disconnected from the line, leaving the load unprotected. For sensitive systems, the traditional way of monitoring the health of MOVs are periodic assessment of all the installed MOVs by disconnecting them and measuring their V1mA. This costly operation can be replaced by an initial investment on the electronic monitoring system that identifies the near-failure MOVs. This goal might be achieved using Machine Learning and artificial Neural Network.

Artificial Neural Network (ANN) is an imitation of biological neural network in the brain. It is a system of nodes and neurons, that receive inputs and send outputs to each other. While in nature, inputs and outputs of the neurons are electrical signals, in the ANN, numerical values are used. Outputs of nodes are determined by calculating the weighted sum of the inputs to represent the total strength of the input signals,

Chapter 6: Conclusions and Future Work 169

adding a bias value and applying a nonlinear activation function on the sum. These outputs are then fed to the other layers as inputs.

Figure 6-1. A typical Deep Neural Network

A typical ANN is shown in Figure 6-3. Each ANN consist of input layer, hidden layers and output layers. The neural network with more than one hidden layer is called Deep Neural Network (DNN). The input layer consists of a list of input features. There are various types of layers, including fully connected, convolution and pooling that allow various types of mathematical operations. Each neuron, shown as a blue line in the figure, represent a weight.

Depending on the problem and data set, the ANN can be trained in two ways: supervised or unsupervised learning. For the supervised learning, a labeled data set is needed. Using unsupervised learning the data set will be unlabeled and the network will learn by itself. Training an ANN means finding the appropriate weights of the neural connections.

In the problem of near-failure MOV detection, the inputs are components of currents and voltages of the

MOV such as amplitude, frequency, and harmonics. Temperature of the MOV’s surface could also be used as an input. A large set of offline measurements of healthy and degraded MOVs are needed to train the

ANN. The larger the training data set, the more accurate the results, so future research would emphasize performing more extensive experimental MOV measurements than performed in this thesis. The data set should contain the MOVs voltage and current information under various voltages within the acceptable voltage range (normally -10% to +5% of nominal value) in different temperatures (for example -20 °C to

85 °C), and whether this MOV is needed to be replaced, based on the health criteria. For example, if V1mA

Chapter 6: Conclusions and Future Work 170

is below 90% of its nominal value the MOV is labeled failed. During inference, data from online measurement would be fed to the network, and the output of the network would determine if this MOV needs to be replaced. This proposed concept is depicted in Figure 6-2. Fundamental components of voltage and current (V1, I1), their harmonics (V3, I3, …), and temperature (T) is measured and fed to the network as inputs. However, future research should also investigate whether other signals should be monitored and measured for the accuracy of the neural network. The goal would be for the network to provide a simple pass/fail output for the health of MOV.

Figure 6-2. Near-failure MOV detection with ANN

• Evaluation of effects of PWM pulses on lifetime of the MOVs

With the expansion of the power electronics in power systems, more MOVs are needed to be installed on the lines with non-sinusoidal voltages. Manufacturers provide data on the life expectancy of the MOVs, based on the conventional sinusoidal voltage tests. Accelerated aging tests can be designed to study the effect of PWM voltage on the lifetime of the MOVs.

The following experiments are suggested in this regard:

The peak resistive leakage current and the power loss of new MOVs could be measured, as the initial states. Then MOVs should be installed in a VFD system, on the motor terminal for a given period. The test can be conducted in an enclosed controllable oven to allow control of the environment temperature. After the exposure period, the peak resistive current and power loss of MOVs can be measured again. These data can be compared to the data collected from a control group that were under MCOV with sinusoidal

Chapter 6: Conclusions and Future Work 171

waveform for a similar period of time. An excess increase of the power loss would indicate a faster degradation. Multiple test groups are needed to be tested under different DC link voltages and switching frequencies. A challenging obstacle of this type of research is that it requires statistical study and multiple samples should be used to compensate for the randomness of the MOV’s behavior.

• Mitigation of the high frequency overvoltages in VFD using combination of MOVs and

filters

There have been extensive studies on how to design a filter for mitigation of overvoltages on motor terminal due to the voltage wave reflection. Typically, the filters have capacitors connected between lines and a neural point. However, future research might attempt to utilize the MOV’s capacitance to reduce the size of the filter capacitors, and at the same time, allow the MOV to be installed on the line to provide the surge protection. This can resolve the installation problem for the MOV with an added benefit of having a smaller capacitance for the filter. In fact, it might lead to different types of MOVs to be designed to have increased filtering abilities specific for the VFD overvoltage mitigation.

Furthers, traditional filters in VFDs can either be passive or active filters. It would be interesting research to include the effects of the MOVs directly into the topology design of the filters. One possible topology is suggested in Figure 6-3. The MOV is connected in parallel to the filter capacitor. Filter parameters will be calculated, and perhaps even reduced, considering the MOV’s capacitance. This topology allows MOVs to be installed between a line and a neutral point. The impact on lifetime and performance of the combined MOV and passive filter design should be compared with the traditional passive filter designs now presently utilized.

Figure 6-3. Combination of MOV and passive filter

Chapter 6: Conclusions and Future Work 172

Further, it may be possible to incorporate MOVs in the active filters as well. The main requirement is that MOVs should be directly connected between lines and neutral (or ground). Resistors or inductors should be avoided in series with MOVs, because they increase the impedance of the surge path and make the surge protection ineffective. This would introduce an entire new area of research in VFD protection.

References 173

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