2012 IEEE International Conference on Power Electronics, Drives and Energy Systems December16-19, 2012, Bengaluru, India

Study of Rectifier Loss Model of the

Flyback Converter

T.Halder Assistant Professor Kalyani Government Engineering College, PIN -741235 Kalyani, West Bengal, India

Abstract—This paper advocates an inclusive power loss modeling efficiency. Reinstating the normal power diode with a and simulation of rectifier diode of the Flyback power converter synchronous rectification (SR) with a MOSFET or Schottky working under continuous conduction mode (CCM) and significantly reduces the massive conduction losses discontinuous conduction mode (DCM). The small signal model which exemplify how designers are able to renovate a standard which takes into considerations all active and passive components diode-rectified isolated Flyback mode power supply of the Flyback converter is implemented by means of MATLAB / PSIM software for computing constructive loss model of Flyback (SMPS) into a self-driven-synchronous-Flyback circuit. A diode. The simple model and computer simulations are well suited Flyback diode with high conduction losses and reduced diode for hypothetical, lucid investigations and simulation model of the reverse-recovery problems are not much predictable [1-15] for converter. Lastly, a loss of rectifier diode is studied for DCM and power circuit’s efficiency of the converter. The proposed CCM for the Flyback converter using synchronous rectification Flyback Schottkey diode can cut the conduction losses and and Schottkey diode. Spurious over voltage and over current may perk up the diode reverse-recovery problems by using a self be caused by a diode switching at the divergent points in the driven auxiliary switch and some additional componenents circuit bringing into play the different type of power diodes and incorporate to output circuit of the converter to achieve zero its rating . Voltage and current waveforms focuses along with current turn-off of the output diode and the reverse-recovery transient switching functions of different rating of diodes. Switching action of the diode is together with various energy currents of the diode is slowed down to reduce the diode losses for each operating cycle. This may give rise conspicuously reverse-recovery losses. All inductive components are realized evaluations of the huge power losses in the diode at high switching on a single magnetic core by utilizing a small series frequency and load of the Flyback converter. in addition to the leakage inductance of the coupled . Furthermore, for the use of this topology in the practical Keywords—Flyback converter; Rectifier diode, CCM, DCM, Loss design, the PWM current mode control is employed for the Model, Simulation and Result proposed Flyback converter. A detailed analysis and a simple are presented. Experimental results for 40W prototype are also I. INTRODUCTION discussed to demonstrate the performance of the proposed ow-a-days there are various customs used to make better Flyback converter under DCM or CCM of circuit operation Ninclusive loss models of the rectifier diode of the Flyback power supply topology. In a conventional diode-rectified Flyback converter, the output diode rectifier is a substantial power loss contributor. The average current in the output diode is equal to the DC output current, and the peak current can be more than a few times higher, depending on the duty cycle. The forward-voltage drop of the diode is characteristically 0.5V for Schottky diodes and 0.75V for standard junction diodes. This large forward voltage drop leads to relatively high conduction losses in the diode and a substantial power limit in Fig. 1 Flyback converter using RCD converter The Fig. 1 shows the basic Flyback converter circuit using common resistance capacitor and diode (RCD) snubber which Manuscript prepared October 9, 2011) This work was supported in part by the is used to dissipate the leakage energy and protect from the Jadavpur University, Kolkata-32 , India and Kalyani Government Engineering atypical surge voltage coming from leakage inductance and its college , Nadia , West Bengal , India. T.Halder is with Government College, Engineering & Textile Technology Berhampore, Murshidabad, (GCETTB) detrimental leakage energy. West Bengal, India e-mail: [email protected].

978-1-4673-4508-8/12/$31.00 ©2012 IEEE

II. BASIC FEATURES OF DCM proportional to the power on-time (ton) for the discontinuous case. • I t does not necessitate a larger core volume than the • CCM, for the same output power requirements. Output rectifiers are operating at zero current just prior to becoming reverse biased. Therefore, reverse recovery • Small inductor size of the Flyback converter. requirements are not critical for these rectifiers. • Easy control characteristics and hardware unit. • Similarly, the power transistor turns on to a current level this is initially zero, so its turn-on time is not critical. • Flux gets reset to zero, so there does not exist a DC flux and flux swing. • Transistor turn-on to zero current also results in low radio • frequency interference (RFI) & electromagnetic The peak current ratings of power devices are higher than interference (EMI) interference generation. those operated in CCM of operation. • • High values of ripple current make output capacitor ESR Natural commutation of the output diode, minimizing requirements quite stringent. In most practical switching loss and secondary noise discontinuous of the Flyback circuits, capacitance values • Low-noise turn-on of the primary power switch. must be increased in order to achieve an adequate ESR. Transient response is correspondingly slower. • Opportunity of a quasi-resonant mode of operation to lower noise even further, the shortcoming of the DCM III. BASIC FEATURES OF CCM operations that the currents in the circuit are higher than for CCM operation The fundamental features of the DCM compared to CCM are focused and curtly explained with consequent point • No RHPZ (Right Half Pole Zero) in the low frequency wise. portion, higher cross over frequency achievable. • Low AC ripple, smaller conduction losses compared to • Simple low-cost secondary diode does not suffer from trr DCM. (reverse recovery time of diode) losses • Low hysteresis losses due to operation on B-H minor • No turn losses on the MOSFET, Id=0 (Drain Current) at loops. the time of turn on (not considering capacitive losses). • It necessitates larger core volume than the complete • Valley switching is possible in a quasi-resonant mode. energy transfer mode (DCM) for the same output power requirements and load. • It is easier to implement synchronous rectification on the secondary side of the Flyback converter • Here, the flux does not get reset to zero, hence there exists • a DC flux and flux swing. Evidently, in this mode, the flux It is not focus to sub-harmonics oscillations in current does not utilize the whole top half of the hysteresis curve. mode control • Low ripples on the output or load side of the converter. • Large AC ripple, together with conduction losses on the MOSFET and resistive paths like ESRs (equivalent series • trr (reverse recovery time) related losses on the both resistances) of copper wires. secondary side diode and the primary side (MOSFET). • It is not focus to sub-harmonics oscillations in current • Requires ultra fast recovery (UFR) diode or mode control. to circumvent excessive losses.

• It is complete energy transfer mode, whereas inductor • Turn-on losses on the MOSFET Id ≠ 0 at time of turn-on current would be zero at a particular interval of time. overlap of vds (t) and i(t) where vds(t) and i(t) is the drain to source voltage and drain current respectively. • It is easier to implement synchronous rectification on the secondary side of the Flyback Converter. • Requires a heavy compensation ramp in peak current- mode control when duty cycle is above 50% • Bigger hysteresis losses on the magnetic core (ferrite core) • • It is more multifarious to become stable in voltage mode A small transformer can be used because the average operation or control energy storage is low. Use of fewer turns also translates into reduced I2R losses. • RHPZ ((Right Half Pole Zero) hampers the accessible bandwidth. • Stability is easier to achieve because at frequencies less than one half the switching frequency there is no net • Despite similar energy storage, the inductance amplifies inductance reflected to the transformer secondary and in CCM and so the transformer dimension will be larger hence no second pole in the input-to-output transfer than that of DCM function. Also, no right half-plane (RHP) zero appears • since energy delivered to the output each cycle is directly It is incomplete energy transfer mode whereas Inductor current would never be zero at any instant.

• Peak current of rectifier and switch is half that of the value recovery period if the reverse current falls too stridently, (low of discontinuous mode of operation. value of (S), stray circuit inductance may cause unsafe over voltage (V ) across the device. It may be required to safeguard • Low output ripple rr the diode using an RCD snubber. During the period t5 large • Recovery time rectifier losses are pronounced current and voltage be present all together in the device. At • high switching frequency this may affect in substantial Feedback loop complicated to stabilize due to two poles enhance in the total power loss. Significant parameters and right half plane zero. defining the turn off characteristics are, peak reverse recovery current (Irr), reverse recovery time (trr), reverse recovery charge IV. DESIGN SPECIFICATION OF THE FLYBACK CONVERTER (Qrr) and the snappiness factor S. Of these parameters, the Input Voltage Vs = 190V-250V (DC) snappiness factor S depends mainly on the construction of the Output Voltage, V0=48V (DC) diode (e.g. drift region width, doping lever, carrier life time Load current, I0= 3A etc.). Other parameters are interrelated and also rely on snappy Magnetizing inductance, Lm= 0.15 mH factor S. Manufacturers typically stipulate these parameters as Primary turns NP = 30, Turns ratio N=5.22 functions of diF/dt for different values of IF. Both Irr and Qrr Switching frequency fsw = 70 KHz increases with IF and diF/dt while trr increases with IF and Output Power at full load, P0 =96W decreases with diF/dt. The reverse recovery characteristics Considering efficiency, ()η at the time of design = 80% shown in the Fig. 3

Input power at full load, Pi = 120W Output capacitance C2= 1000 μ F

V. TURN ON PERFORMANCE OF A FLYBAK CONVERTER’S OUTPUT RECTIFIER DIODE Practical power Flybak Diode is habitually used in circuits with di/dt limiting . The rate of rise of the forward current through the diode during turn on has major outcome on the forward voltage drop characteristics. A typical turn on transient response is shown in the Fig. 2. Fig. 3 Reverse Recovery characteristics of a Flybak diode

It is figurative of a selective set of diodes called “normal recovery” or “soft recovery” diode (S >1) The softness factor of a diode in terms of time ratio as per Fig. 3 is given as: t S = 5 (1) t Fig. 2 Forward current and voltage waveforms of a Flyback diode during Turn 4 on operation If S<<<1, then it can be treated as excellent ultra fast recovery diode which has very less losses either DCM or CCM of VI. TURN OFF PERFORMANCE OF A FLYBAK CONVERTER Flyback converter circuit operation. Hence the proper choice OUTPUT RECTIFIER DIODE of diode plays an important role of loss optimization of the The diode current does not end at zero value instead it rises converter. in the negative trend to Irr called “peak reverse recovery The whole recovery time (trr) in this case is a few tens of current” which can be as good as to IF. In many power microseconds. While this is good enough for line frequency electronic Flyback circuit, this reverse current flows through rectifiers, these diodes are also called rectifier ultra first the main power switch in addition to the load current. recovery (UFR) diodes high switching frequency pulse width Therefore, this reverse recovery current has to be accounted for modulation (PWM) Flyback SMPS argues UFR diode. Diode while selecting the main power switch. Voltage drop across reverse recovery time can be decreased by rising the rate of the diode does not change appreciably from its steady state reduction of the forward current (i.e., by reducing stray circuit value till the diode current reaches reverse recovery level. In inductance) and by using “snappy” recovery (S<<1) diode. The many power electric circuits, this may generate an effective troubles with this process are two points to increase of diF/dt short circuit across the supply, current being limited merely by also increases the magnitude of Irr and large recovery current the stray wiring inductance. Also in high frequency switching coupled with”snappy” recovery may give rise to current and circuits (e.g., Flyback SMPS) if the time period t4 is analogous voltage oscillation in the diode due to the resonant circuit to switching cycle qualitative revision to the circuit formed by the secondary leakage inductance of the couple performance is probable. Towards the end of the reverse

inductor of the Flyback converter and the diode depletion layer consisting of small value of capacitance. On the other hand, an archetypal recovery trait of a snappy recovery diode for better study and transient analysis is shown in Fig. 4.

Fig. 6 Schematic secondary voltage waveform Flyback Power converter As per Fig. 8 v(s) is the instantaneous secondary voltage of the Flyback converter given by Vss (t) = Vin sin (wt)

Fig. 4 Snappy recovery characteristics of Flyback diode VIII. AVERAGE FORWARD POWER LOSS (PAVF) P Approximately all or conduction power loss in a Flyback In some very high switching frequency, (f ) wide applications AVF, sw diode occurs during forward conduction angle ()φ condition. (fsw >50KHZ), enrichment in the reverse recovery performance The forward power loss is therefore a very important offered by customary fast recovery diode is not satisfactory. If parameter in designing the cooling or heat sink arrangement. the required reverse blocking voltage is less (<100V) schottky Average forward power loss over a full switching cycle is diodes are preferred over fast recovery diodes. Compared to p- precised by the manufacturers as a function of the average n junction diodes schottky diodes have very little Turn OFF forward current (IAVF) for different conduction angles ()φ as transient and almost no Turn ON transient. On state voltage shown in Fig. 7 drop is also less compared to a p-n junction diode for equal forward current densities. However, reverse breakdown voltages of these diodes are less (below 200V) Power schottky diodes with forward current rating in surplus of 100A are available

VII. MAXIMUM AVERAGE FORWARD CURRENT (IFAVM) OF FLYBACK DIODE The Flyback Diode is used in rectifier circuits supplying a DC (average) current to be load. In such cases the average load current and the diode forward current commonly have an easy Fig. 7 Average forward power loss vs. average forward current of a power relationship. Therefore, it will be of importance to be on Diode. proverbial provisos with the peak average current a diode can conduct in the forward direction. This specification gives the maximum average value of high switching frequency half IX. LOSS MODELING OF THE FLYBACK DIODE cycle square wave current permitted to flow through the diode The losses of the output diode (Pd) includes the on state losses in the forward direction. Average current rating of a diode and leakage current losses. Since the diodes have the finite on decreases with diminution in conduction angle ()φ due to state voltage (Vf) of approximately 0.5V. On state loss simply can be written for a load current (Io) increase in current form factor. Both IFRMS and IFAVM ratings are given at a specified case operating temperature. If the case Pd=VfIo (2) temperature increases beyond this boundary these ratings has The reverse recovery current (If) of the typical diode is about to be reduced in the same way. De-rating curves endow with 30 μA . This gives leakage current losses. This loss is not so by the manufacturers grant the correlation between IFAVM much pronounced compared to diode rectifier losses. The (IFRMS) with permissible case temperature as shown in the voltage reflected (VR) across the rectifier diode is two times of following Fig. 5 the original input voltage times the core ratio given by as:

ND2V V = o (3) R ()− D1 Duty ratio defined as: t D= on (4) + tt offon

Here, on time= ton and off time= toff Fig. 5 Schematic De-rating curves for the forward current of a Flyback Power Rectifier Diode Hence reverse current leakage loss is given by as:

P =V I (5) π f R f 1 3 sinVVQ θ the leakage loss of the diode is given by as: P = inorr dθ (16) av π ∫ ()+ o V2P4L inim θ Vsin o PdL=IfVR (6) Once again, the charging and discharging of the junction ⎛ V2 ⎞ If M= ⎜ in ⎟ (17) capacitance (CJ) gives a loss ⎜ ⎟ Vo 2 ⎝ ⎠ PCJ=0.5VJ fswCJ (7) Maximum value of M = 7.36 Basically, this loss is an unimportant compared to forward loss of the diodes. The value of junction capacitance of the diode is Minimum value of M =5.59 very small because its value is order of Pico Farad (pF) The equation (16) is computed by Matlab program as best fit Similarly, the switching loss of the diode is given by equation approximation given as: as: V 3 ()+ ⎛ s + ⎞ inrr 0.637VQ M0.004 Psw = f ⎜ ⎟QV (8) P = (18) SW ⎜ ⎟ rr0 av ()+ ⎝ N ⎠ im 0.729M1P4L Where, Q is the diode reverse recovered charge and the rr ⎛ VQ 2 ⎞ voltage across the diode when it is turned off is given by If K = ⎜ inrr ⎟ , then equation reduces as: ⎜ ⎟ ⎝ P4L im ⎠ ⎛ V ⎞ s + (9) Voff = ⎜ V0 ⎟ The range of K in terms of Q is given by ⎝ N ⎠ rr Maximum value of K = 1.32*106 Where, Vs is the input voltage, Vo is the load voltage of the Flyback converter and N is turns ratio of couple inductor. Minimum value of K = 0.75*106 Reverse recovery switching losses are the reverse recovered ()+ = ⎡ 0.637 M0.004 ⎤ charge and the reverse voltage across the diodes. The reverse Av KP ⎢ ⎥ (19) ()+ 0.729M1 recovery switching losses over the switching cycle is given by ⎣ ⎦

()θ And considering two component of Pav (i.e. p1 and p2) Pi= QVinrr sinθ f sw (10) ()+ 6 ⎡ 0.637 M0.004 ⎤ It is assumed that secondary voltage of the Flyback P1=0.75*10 ⎢ ⎥ (20) transformer, i.e. ⎣ ()+ 0.729M1 ⎦ V (t) =V Sin ω t (11) ()+ ss in 6 ⎡ 0.637 M0.004 ⎤ P2= 1.32*10 ⎢ ⎥ (21) ()+ 0.729M1 θ = ωt (12) ⎣ ⎦ ω = π Where K can be considered as the scaling factor average power 2 fsw θ )( (13) loss of the diode Approximately, it is to be considered as sinusoidal signal for very simple analysis X. CONSERVATION OF ENERGY OF THE FLYBACK CIRCUIT VV 2 Power out cannot go above power in. Sum up output power θ )(f = ino (14) (VI) of each output at maximum steady state load plus sw ()+ V2P4L inim θ Vsin o allowances for parasitic output power losses (diode and resistive losses). Divide power in watts by operating switching Where, Vin is the secondary peak voltage of the Flyback frequency. The result is the energy in Joules that must be converter, Pi is the input power of the Flyback converter and discharged each cycle into the output storage capacitor during Lm is the magnetizing inductance of the Flyback transformer. steady state operation. It is also the amount of energy that must Combing equations (4) & (8), it yields as: be added to the Flyback transformer (or couple inductor) 3 θ during the charging stage. The energy being transferred equals inorr sinVVQ Pav = (15) ()+ 1 V2P4L inim θ Vsin o E ()2 −= IIL 2 (22) T 2 P min The equation presents the average loss of the rectifier diode. 2 2 Integrating the equation over a line cycle gives the average Where, I P and I min are the peak and minimum current of the reverse recovery switching losses in terms of converter couple inductor respectively. parameters under the contemplation of simplified linear model If operating in the continuous conduction mode (CCM), the of the inherent current fed Flyback converter given as: stored energy will exceed the energy being transferred because

the starting level of stored energy is above zero (Imin > 0). The The load versus efficiency curved focuses on power losses of Flyback transformer (or couple inductor) must be designed to Rectifier diode of the Flyback converter using synchronous handle the peak stored energy. The power source is given by rectification and Schottkey diode of the different voltage as: ratings at the various load condition under DCM of operation

1 ()2 −= 2 XII. CONCLUSION E P P IIL min (23) 2 The theoretically easy amend of adding an SR to the classic It will have to supply the transferred energy plus the Flyback topology can significantly reduce overall system parasitic switching and resistive losses of the charging circuit, power losses. The power level at which such a amendment is plus some power allowance for transient conditions. Taking realistic has been decreasing with the speedy encroachment in this value and divide by the power supply voltage. The result power MOSFET technology. Hence, synchronous rectification will be the average input current. Fig. 1 points out common is now pertinent to an ever-growing range of products. The topology of a Flyback converter with an only output terminal, lower power dissipation of an SR allows designers to take where the power network and control block to the key route, advantage of least components that have less heat sinking, thus D2 is the output diode, R2 the output resistance, C2 output increasing power density while lowering assembly costs, product size, compact PCB size and light weight. Note that if capacitance, and Vo the output voltage. To efficiently examine the Flyback converter, the following assumptions and the SR MOSFET is allowed to switch during no-load/standby oversimplifications are made, where UC-3525a is chosen as conditions, the system power performance could be pulse width modulation (PWM) controller at the gate to source compromised. The SR-MOSFET switching losses, in addition of the power MOSFET of the Flyback converter ultimate to the quiescent power required by the SR controller IC, can be phase under closed loop control operation due to its current fed limiting factors in achieving the best potential system no-load nature and inherent current fed topology to full load performances. It can be used as communication power supply and industrial product XI. RESULT REFERENCES The graphical representation (p1, p2) vs. M as shown Fig. 8 [1] P. C. Sen, “Power Electronics” Tata McGraw Hill Publishing Company which represents a pair of parallel straight line as per Limited, New Delhi, 1987. comparative and comprehensive short study [2] Jacob Millman, Christos C. Halkias, “Integrated Electronics, Analog and Digital Circuits and Systems”, Tata McGraw-Hill Publishing Company 6.8 Limited, New Delhi, 1991. [3] R.W. Erickson, D. Maksimovic, “Fundamentals of power electronics”, 6.6 Second edition, Kluwer Academic Press, 2001. 6.4 [4] I. Stojanov, S.Pasca, “Computer aided design of electronics circuits. For P1 and 6 6.2 K=0.75*10 PSPICE practical guide”, (in Romanian), Editura Teora, Bucuresti, 1997. For P2 and [5] V. Popescu, D. Lascu, D. Negoitescu, “Supply sources in 6 6 K=1.32*10 -----> (p1,p2) -----> telecommunications”, (in romanian), Editura de Vest,Timisoara, 2002.

5.8 [6] D. Alexa, F. Ionescu, L. Gatlan, A. Lazar, “Power converters with resonant circuits”, (in romanian), Editura Tehnica, Bucuresti, 1998. 5.6 [7] S. Patent 4,961,044, Oct. 1990

5.4 [8] S. Lungu, O. Pop, “Modeling of electronic circuits”, (in romanian), Editura 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5 1.6 1.7 1.8 ----->M 5 x 10 Casa Cartii de Stiinta, Cluj-Napoca, 2006. [9] N.D. Trip, “Industrial electronics”, (in Romanian), Editura Universitatii din Fig. 8 Graphical representation (p1,p2) vs. M Oradea, Oradea, 2004. [10] Ned Mohan, Tore M. Undeland, William P. Robbins, “Power electronics, Converters, Application and Design” John Wiley & Sons(Asia), Publishers. Third Edition 2003 [11] J. X chen , Y.G. Guo and J.G. Zhu, “Development of a permanent magnets brushless DC motor for driving high speed embroidery machines.ÏEEE trans , Magn, vol 43, no. 11, pp,4004-4009, Nov.2007. [12] J. X chen , J.G. Zhu and Y.G. Guo “Calculation of power loss in output diode of a flybaack switching DC-DC converter ,”in Proc. Int. Power electronics and motion control Conf. Shanghai, china , Aug,2006, vol, 1 pp 1-5 [13] R. Watson, F. C. Lee, G. C. Hua, "Utilization of an Active-Clamp Circuit to Achieve Soft Switching in Flyback Converters," IEEE Power Electronics Specialists' Conf. Rec., 1994, pp. 909-916 [14] J. X chen , Y.G. Guo and J.G. Zhu, “Development of a permanent Magnets brushless Dc motor for driving high speed embroidery machines.IEEE trans , Magn, vol 43, no. 11, pp,4004-4009, Nov.2007.

[15] T.Halder “Improved Performance Analysis of the Clamp Circuits with Fig. 9 Simplified Flyback waveforms with schottky diode and Synchronous Flyback Converter, International Journal of Emerging Technology and Rectification (SR) -MOSFET output rectification Advanced Engineering) IJETAE, ISSN (2250-2459) Volume 2, Issue 1 January-2012” pp 1-8