boy30444_ch18.qxd 3/24/06 2:53 PM Page 787 NetworkNetwork TheoremsTheorems (ac)(ac) 1818

• Be able to apply the superposition theorem to ac Objectives networks with independent and dependent sources. • Become proficient in applying Thévenin’s theorem to ac networks with independent and dependent sources. • Be able to apply Norton’s theorem to ac networks with independent and dependent sources. • Clearly understand the conditions that must be met for maximum power transfer to a load in an ac network with independent or dependent sources.

18.1 INTRODUCTION This chapter parallels Chapter 9, which dealt with network theorems as applied to dc networks. Reviewing each theorem in Chapter 9 before beginning this chapter is recommended because many of the comments offered there are not repeated here. Due to the need for developing confidence in the application of the various theorems to net- works with controlled (dependent) sources, some sections have been divided into two parts: independent sources and dependent sources. Theorems to be considered in detail include the superposition theorem, Thévenin’s and Norton’s theorems, and the maximum power transfer theorem. The substitution and reciproc- ity theorems and Millman’s theorem are not discussed in detail here because a review of Chapter 9 will enable you to apply them to sinusoidal ac networks with little difficulty.

18.2 SUPERPOSITION THEOREM You will recall from Chapter 9 that the superposition theorem eliminated the need for solv- ing simultaneous linear equations by considering the effects of each source independently. To consider the effects of each source, we had to remove the remaining sources. This was ac- complished by setting voltage sources to zero (short-circuit representation) and current sources to zero (open-circuit representation). The current through, or voltage across, a portion of the network produced by each source was then added algebraically to find the total solution for the current or voltage. The only variation in applying this method to ac networks with independent sources is that we are now working with impedances and phasors instead of just resistors and real numbers. The superposition theorem is not applicable to power effects in ac networks since we are still dealing with a nonlinear relationship. It can be applied to networks with sources of dif- ferent frequencies only if the total response for each frequency is found independently and the results are expanded in a nonsinusoidal expression, as appearing in Chapter 25. One of the most frequent applications of the superposition theorem is to electronic systems in which the dc and ac analyses are treated separately and the total solution is the sum of the Th two. It is an important application of the theorem because the impact of the reactive elements boy30444_ch18.qxd 3/24/06 2:53 PM Page 788

788 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

changes dramatically in response to the two types of independent sources. In addition, the dc analysis of an electronic system can often de- fine important parameters for the ac analysis. Example 18.4 demon- strates the impact of the applied source on the general configuration of the network. We first consider networks with only independent sources to provide a close association with the analysis of Chapter 9.

Independent Sources EXAMPLE 18.1 Using the superposition theorem, find the current I through the 4 reactance X in Fig. 18.1. 1 L2 2

I X 4 XC 3 L1

+ XL 4 2 – ∠ ° ∠ °

E1 = 10 V 0 E2 = 5 V 0 – +

FIG. 18.1 Example 18.1.

Solution: For the redrawn circuit (Fig. 18.2), I Z Z Z j X j 1 3 1 L1 4

Z j X j 2 L2 4 + Z 2 – Z3 j XC j 3

E1 E2 – Considering the effects of the voltage source E (Fig. 18.3), we have + 1 Z 2Z3 j 4 j 3 12 Z 1 21 2 j 12 2 3 Z 2 Z3 j 4 j 3 j FIG. 18.2 Assigning the subscripted impedances to the network 12 90° in Fig. 18.1. E 10 V 0° 10 V 0° I 1 s1 Z 2 3 Z1 j 12 j 4 8 90° 1.25 A 90°

I I s1 I s1 Z1 Z3 Z1

Z Z + + 2 2 3

E1 E1 – –

FIG. 18.3

Determining the effect of the voltage source E1 on the current I of the network in Fig. 18.1. boy30444_ch18.qxd 3/24/06 2:53 PM Page 789

Th SUPERPOSITION THEOREM ⏐⏐⏐ 789

and Z I 3 s1 I (current divider rule) Z2 Z3 j 3 j 1.25 A 3.75 A 1 21 2 3.75 A 90° j 4 j 3 j 1

Considering the effects of the voltage source E2 (Fig. 18.4), we have

Z1 Z3 Z3

Z2 Z12 – –

E2 E2 I Is Is 2 + 2 +

FIG. 18.4

Determining the effect of the voltage source E2 on the current I of the network in Fig. 18.1.

Z1 j 4 Z j 2 1 2 N 2 E 5 V 0° 5 V 0° I 2 5 A 90° s2 Z1 2 Z3 j 2 j 3 1 90° I s2 I′ and I 2.5 A 90° 2 X 4 I L2 The resultant current through the 4 reactance X (Fig. 18.5) is L2 I″ I I I 3.75 A 90° 2.50 A 90° j 3.75 A j 2.50 A FIG. 18.5 j 6.25 A Determining the resultant current for the network in I 6.25 A 90° Fig. 18.1.

EXAMPLE 18.2 Using superposition, find the current I through the 6 resistor in Fig. 18.6.

I

X = 6 R = 6 + L

∠ ° ∠ °

E1 = 20 V 30 I1 2 A 0 XC = 8 –

FIG. 18.6 Example 18.2. boy30444_ch18.qxd 3/24/06 2:54 PM Page 790

790 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Solution: For the redrawn circuit (Fig. 18.7), Z1 Z2 Z1 j 6 Z2 6 j 8 + I Consider the effects of the current source (Fig. 18.8). Applying the cur- E1 I1 – rent divider rule, we have Z I j 6 2 A j 12 A 1 1 1 21 2 I Z1 Z 2 j 6 6 j 8 6 j 2 12 A 90° FIG. 18.7 Assigning the subscripted impedances to the network 6.32 18.43° in Fig. 18.6. I 1.9 A 108.43° Consider the effects of the voltage source (Fig. 18.9). Applying Ohm’s Z1 Z2 law gives us I E E 20 V 30° I 1 1 I 1 ZT Z1 Z 2 6.32 18.43° 3.16 A 48.43° The total current through the 6 resistor (Fig. 18.10) is I I I FIG. 18.8 1.9 A 108.43° 3.16 A 48.43° Determining the effect of the current source I1 on the (0.60 A j 1.80 A) (2.10 A j 2.36 A) current I of the network in Fig. 18.6. 1.50 A j 4.16 A I 4.42 A 70.2°

Z1 Z2

+ I EXAMPLE 18.3 Using superposition, find the voltage across the 6 E1 resistor in Fig. 18.6. Check the results against V6 I(6 ), where I is – the current found through the 6 resistor in Example 18.2. Solution: For the current source, V 6 I 6 1.9 A 108.43° 6 11.4 V 108.43° FIG. 18.9 1 2 1 21 2

Determining the effect of the voltage source E1 on For the voltage source, the current I of the network in Fig. 18.6. V I 6 3.16 A 48.43° 6 18.96 V 48.43° 6 1 2 1 21 2 R The total voltage across the 6 resistor (Fig. 18.11) is 6 V6 V 6 V 6 ′ ″ I I 11.4 V 108.43° 18.96 V 48.43° I 3.60 V j 10.82 V 12.58 V j 14.18 V 1 2 1 2 FIG. 18.10 8.98 V j 25.0 V Determining the resultant current I for the network V 26.5 V 70.2° in Fig. 18.6. 6 ′ + V 6 – ″ + V 6 – R

6

+ V6 –

FIG. 18.11

Determining the resultant voltage V6 for the network in Fig. 18.6. boy30444_ch18.qxd 3/24/06 2:54 PM Page 791

Th SUPERPOSITION THEOREM ⏐⏐⏐ 791

Checking the result, we have V6 I(6 ) (4.42 A 70.2°)(6 ) 26.5 V 70.2° (checks)

EXAMPLE 18.4 For the network in Fig. 18.12, determine the sinu-

soidal expression for the voltage y3 using superposition.

E1 = 12 V

R2 1 k

R1 XL

0.5 k 2 k + + ∠ ° X 10 k R 3 k E2 = 4 V 0 C 3 v3 – –

FIG. 18.12 Example 18.4.

Solution: For the dc analysis, the capacitor can be replaced by an open- E1 = 12 V circuit equivalent, and the inductor by a short-circuit equivalent. The re- sult is the network in Fig. 18.13. The resistors R1 and R3 are then in parallel, and the voltage V3 can be R2 1 k determined using the voltage divider rule: R1

RR R 0.5 k 3 k 0.429 k 1 3 0.5 k RE + 1 R 3 k V and V 3 3 3 R R 2 – 0.429 k 12 V 5.148 V 1 21 2 0.429 k 1 k 1.429 FIG. 18.13 V3 3.6 V Determining the effect of the dc voltage source E1 on For the ac analysis, the dc source is set to zero and the network is re- the voltage y3 of the network in Fig. 18.12. drawn, as shown in Fig. 18.14.

R1 XL

0.5 k 2 k + + ∠ ° R2 = 1 k XC = 10 k R3 = 3 k V E2 = 4 V 0 3 – –

FIG. 18.14 Redrawing the network in Fig. 18.12 to determine the effect of the ac voltage

source E2 . boy30444_ch18.qxd 3/24/06 2:54 PM Page 792

792 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

The block impedances are then defined as in Fig. 18.15, and series- Z I 1 3 parallel techniques are applied as follows: + I + s Z 0.5 k 0° Z Z V 1 E2 2 3 3 Z R 0° X 90° – – 2 1 2 1 C 2 1 k 0° 10 k 90° 10 k 90° ZT 1 21 2 1 kj 10 k 10.05 84.29° 0.995 k 5.71° FIG. 18.15 Z R j X 3 kj 2 k3.61 k 33.69° Assigning the subscripted impedances to the network 3 3 L in Fig. 18.14. and ZT Z1 Z2 Z3

0.5 k 0.995 k 5.71° 3.61 k 33.69° 1 2 1 2 1.312 k 1.57°

Calculator Solution: Performing the above on the TI-89 calculator requires the sequence of steps in Fig. 18.16.

.5((0.995∠()5.71°)(3.61∠33.69°))((0.995∠()5.71°)(3.61∠33.69°)) Polar 1.31E0∠1.55E0

FIG. 18.16

Using the TI-89 calculator to determine ZT for the network in Fig. 18.12.

E 2 4 V 0° Is 3.05 mA 1.57° ZT 1.312 k 1.57° Current divider rule: Z I 0.995 k 5.71° 3.05 mA 1.57° 2 s 1 21 2 I3 Z2 Z3 0.995 k 5.71° 3.61 k 33.69° 0.686 mA 32.74° with

v3 32.74° V3 (I3 u)(R3 0° (0.686 mA 32.74°)(3 k 0°) 6.51 V 2.06 V 32.74°

3.6 V The total solution: 0.69 V y3 y3 (dc) y3 (ac) 0 qt 3.6 V 2.06 V 32.74° y3 3.6 2.91 sin(Vt 32.74°) FIG. 18.17

The resultant voltage y3 for the network in The result is a sinusoidal voltage having a peak value of 2.91 V riding Fig. 18.12. on an average value of 3.6 V, as shown in Fig. 18.17.

Dependent Sources For dependent sources in which the controlling variable is not determined by the network to which the superposition theorem is to be applied, the ap- plication of the theorem is basically the same as for independent sources. The solution obtained will simply be in terms of the controlling variables.

EXAMPLE 18.5 Using the superposition theorem, determine the cur-

rent I2 for the network in Fig. 18.18. The quantities m and h are constants. boy30444_ch18.qxd 3/24/06 2:54 PM Page 793

Th SUPERPOSITION THEOREM ⏐⏐⏐ 793

+ V – R1 I I 4 2 + R2 6 V hI

– XL 8

Z1 I2 + FIG. 18.18 V hI Z2 Example 18.5. – Solution: With a portion of the system redrawn (Fig. 18.19), Z1 R1 4 Z2 R2 j XL 6 j 8 FIG. 18.19 For the voltage source (Fig. 18.20), Assigning the subscripted impedances to the network mV mV mV in Fig. 18.18. I Z1 Z2 4 6 j 8 10 j 8 mV Z1 0.078 mV 38.66° 12.8 38.66° > + I For the current source (Fig. 18.21), V Z2 – Z hI 4 hI 11 2 1 21 2 I 4 0.078 hI 38.66° Z1 Z2 12.8 38.66° 1 2 0.312hI 38.66° FIG. 18.20 The current I2 is Determining the effect of the voltage-controlled voltage source on the current I2 for the network in I2 I I Fig. 18.18. 0.078 mV/ 38.66° 0.312hI 38.66° For V 10 V 0°, I 20 mA 0°, m 20, and h 100, Z1 I I2 0.078(20)(10 V 0°)/ 38.66° 0.312(100)(20 mA 0°)38.66° Z 15.60 A 38.66° 0.62 A 38.66° hI1 2 I2 16.22 A 38.66°

For dependent sources in which the controlling variable is determined FIG. 18.21 by the network to which the theorem is to be applied, the dependent Determining the effect of the current-controlled source cannot be set to zero unless the controlling variable is also zero. current source on the current I2 for the network in For networks containing dependent sources (as in Example 18.5) and de- Fig. 18.18. pendent sources of the type just introduced above, the superposition the- orem is applied for each independent source and each dependent source mV not having a controlling variable in the portions of the network under in- – + vestigation. It must be reemphasized that dependent sources are not I1 IL sources of energy in the sense that, if all independent sources are re- + moved from a system, all currents and voltages must be zero. I R1 V RL –

EXAMPLE 18.6 Determine the current IL through the resistor RL in Fig. 18.22.

Solution: Note that the controlling variable V is determined by the net- FIG. 18.22 work to be analyzed. From the above discussions, it is understood that the Example 18.6. boy30444_ch18.qxd 3/24/06 2:54 PM Page 794

794 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

dependent source cannot be set to zero unless V is zero. If we set I to zero, the network lacks a source of voltage, and V 0 with mV 0. The re-

sulting IL under this condition is zero. Obviously, therefore, the network must be analyzed as it appears in Fig. 18.22, with the result that neither source can be eliminated, as is normally done using the superposition theorem. Applying Kirchhoff’s voltage law, we have VL V mV (1 m)V V 1 m V L and IL 1 2 RL RL The result, however, must be found in terms of I since V and mV are only dependent variables. Applying Kirchhoff’s current law gives us 1 m V V I I1 IL 1 2 R1 RL 1 1 m and I V a R1 RL b I or V 1 R 1 m R 1 > 1 2 31 2> L 4 Substituting into the above yields

1 m V 1 m I I 1 2 1 2 L R R a 1 R 1 m R b L L 1 > 1 2 31 2> L 4 Therefore, 1 M R I I 1 2 1 L R 1 M R L 1 2 1

18.3 THÉVENIN’S THEOREM Thévenin’s theorem, as stated for sinusoidal ac circuits, is changed only to include the term impedance instead of resistance; that is, any two-terminal linear ac network can be replaced with an Z Th equivalent circuit consisting of a voltage source and an impedance in + series, as shown in Fig. 18.23. E Th Since the reactances of a circuit are frequency dependent, the Thévenin – circuit found for a particular network is applicable only at one frequency. The steps required to apply this method to dc circuits are repeated here with changes for sinusoidal ac circuits. As before, the only change is the replacement of the term resistance with impedance. Again, dependent FIG. 18.23 and independent sources are treated separately. Thévenin equivalent circuit for ac networks. Example 18.9, the last example of the independent source section, in- cludes a network with dc and ac sources to establish the groundwork for possible use in the electronics area.

Independent Sources 1. Remove that portion of the network across which the Thévenin equivalent circuit is to be found. boy30444_ch18.qxd 3/24/06 2:55 PM Page 795

Th THÉVENIN’S THEOREM ⏐⏐⏐ 795

2. Mark (, •, and so on) the terminals of the remaining two-terminal network.

3. Calculate ZTh by first setting all voltage and current sources to zero (short circuit and open circuit, respectively) and then finding the resulting impedance between the two marked terminals.

4. Calculate ETh by first replacing the voltage and current sources and then finding the open-circuit voltage between the marked terminals. 5. Draw the Thévenin equivalent circuit with the portion of the circuit previously removed replaced between the terminals of the Thévenin equivalent circuit.

EXAMPLE 18.7 Find the Thévenin equivalent circuit for the network external to resistor R in Fig. 18.24.

XL = 8 + ∠ ° E = 10 V 0 XC 2 R –

Thévenin

FIG. 18.24 Example 18.7.

Solution: Steps 1 and 2 (Fig. 18.25): Z1 j XL j 8 Z2 j XC j 2

Z1 Z + 1

E = 10 V ∠ 0° Z2 ZTh – Z2 Thévenin

FIG. 18.25 FIG. 18.26 Assigning the subscripted impedances to the network in Fig. 18.24. Determining the Thévenin impedance for the network in Fig. 18.24. Step 3 (Fig. 18.26):

2 Z Z j 8 j 2 j 16 16 Z1 1 2 1 21 2 + ZTh Z1 Z 2 j 8 j 2 j 6 6 90° + 2.67 90° Z E 2 ETh – Step 4 (Fig. 18.27): – Z E 2 ETh (voltage divider rule) Z1 Z2 FIG. 18.27 j 2 10 V j 20 V 1 21 2 3.33 V 180° Determining the open-circuit Thévenin voltage for j 8 j 2 j 6 the network in Fig. 18.24. boy30444_ch18.qxd 3/24/06 2:55 PM Page 796

796 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Step 5: The Thévenin equivalent circuit is shown in Fig. 18.28. ∠ ° ZTh = 2.67 –90

ZTh X = 2.67 + + C ∠ ° ∠ ° ETh = 3.33 V – 180 R ETh = 3.33 V – 180 R – –

FIG. 18.28 The Thévenin equivalent circuit for the network in Fig. 18.24.

EXAMPLE 18.8 Find the Thévenin equivalent circuit for the network external to branch a-a in Fig. 18.29.

R XL XL = 5 R 1 1 2 a 3

6 8 7 + + R2 3 ∠ ° ∠ ° E1 10 V 0 E2 30 V 15 – – XC 4

a Thévenin

FIG. 18.29 Example 18.8. Solution: Steps 1 and 2 (Fig. 18.30): Note the reduced complexity with subscripted impedances: Z R j X j 1 1 L1 6 8 Z2 R2 j XC 3 j 4 Z j X j 3 L2 5

Z1 Z3 a

+ ∠ ° E1 10 V 0 Z2 –

a Thévenin

FIG. 18.30 Assigning the subscripted impedances for the network in Fig. 18.29.

Step 3 (Fig. 18.31): Z Z 10 53.13° 5 53.13° Z Z 1 2 j 5 1 21 2 Th 3 Z Z 6 j 8 3 j 4 1 2 1 2 1 2 50 0° 50 0° j 5 j 5 9 j 4 9.85 23.96° j 5 5.08 23.96° j 5 4.64 j 2.06 ZTh 4.64 j 2.94 5.49 32.36° boy30444_ch18.qxd 3/24/06 2:55 PM Page 797

Th THÉVENIN’S THEOREM ⏐⏐⏐ 797

Z1 Z3 a

Z2 ZTh

a

FIG. 18.31 Determining the Thévenin impedance for the network in Fig. 18.29.

Step 4 a a I E (Fig. 18.32): Since - is an open circuit,Z3 0. Then Th is the voltage drop across Z2: Z E 2 ETh (voltage divider rule) Z 2 Z1 5 53.13° 10 V 0° 1 21 2 9.85 23.96° 50 V 53.13° E 5.08 V 77.09° Th 9.85 23.96°

Z1 Z3 a + + I = 0 Z3 E1 Z2 ETh – – a

FIG. 18.32 Determining the open-circuit Thévenin voltage for the network in Fig. 18.29.

Step 5: The Thévenin equivalent circuit is shown in Fig. 18.33.

X a R3 R L a R3 ZTh 7 7 4.64 2.94 + 4.64 + j2.94 + + + E ∠ ° E 5.08 V ∠ –77.09° ∠ ° Th E2 30 V 15 Th E2 30 V 15 – – – – 5.08 V ∠ –77.09°

a′ a′

FIG. 18.33 The Thévenin equivalent circuit for the network in Fig. 18.29.

The next example demonstrates how superposition is applied to elec- tronic circuits to permit a separation of the dc and ac analyses. The fact that the controlling variable in this analysis is not in the portion of the net- work connected directly to the terminals of interest permits an analysis of the network in the same manner as applied above for independent sources. boy30444_ch18.qxd 3/24/06 2:55 PM Page 798

798 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

EXAMPLE 18.9 Determine the Thévenin equivalent circuit for the

transistor network external to the resistor RL in the network in Fig. 18.34. Then determine VL .

12 V

R 2 k C C2 Thévenin RB 1 M C Rs 1 10 Transistor + + 0.5 k 10 RL = 1 k VL Ei – –

FIG. 18.34 Example 18.9.

Solution: Applying superposition. dc Conditions Substituting the open-circuit equivalent for the cou-

pling capacitor C2 will isolate the dc source and the resulting currents from the load resistor. The result is that for dc conditions, VL 0 V. Al- though the output dc voltage is zero, the application of the dc voltage is important to the basic operation of the transistor in a number of impor- tant ways, one of which is to determine the parameters of the “equivalent circuit” to appear in the ac analysis to follow. ac Conditions For the ac analysis, an equivalent circuit is substituted for the transistor, as established by the dc conditions above, that will be- have like the actual transistor. A great deal more will be said about equiv- alent circuits and the operations performed to obtain the network in Fig. 18.35, but for now we limit our attention to the manner in which the Thévenin equivalent circuit is obtained. Note in Fig. 18.35 that the equiv- alent circuit includes a resistor of 2.3 k and a controlled current source whose magnitude is determined by the product of a factor of 100 and the

current I1 in another part of the network.

Rs I 0.5 k 1 + + 100I Ei RB 1 M 2.3 k 1 RC 2 k RL 1 k VL – –

Transistor equivalent Thévenin circuit

FIG. 18.35 The ac equivalent network for the transistor amplifier in Fig. 18.34.

Note in Fig. 18.35 the absence of the coupling capacitors for the ac analysis. In general, coupling capacitors are designed to be open circuits for dc analysis and short circuits for ac analysis. The short-circuit equiv- alent is valid because the other impedances in series with the coupling ca- boy30444_ch18.qxd 3/24/06 2:55 PM Page 799

Th THÉVENIN’S THEOREM ⏐⏐⏐ 799

pacitors are so much larger in magnitude that the effect of the coupling

capacitors can be ignored. Both RB and RC are now tied to ground because the dc source was set to zero volts (superposition) and replaced by a short-circuit equivalent to ground.

For the analysis to follow, the effect of the resistor RB will be ignored since it is so much larger than the parallel 2.3 k resistor.

ZTh When Ei is set to zero volts, the current I1 will be zero amperes, and the controlled source 100I1 will be zero amperes also. The result is an open-circuit equivalent for the source, as appearing in Fig. 18.36. R 2 k Z It is fairly obvious from Fig. 18.36 that C Th ZTh 2k

ETh For ETh , the current I1 in Fig. 18.35 will be FIG. 18.36 E E E i i i Determining the Thévenin impedance for the I1 Rs 2.3 k 0.5 k 2.3 k 2.8 k network in Fig. 18.35. E and 100I 100 i 35.71 103 E 1 1 2a2.8 k b > i

Referring to Fig. 18.37, we find that – + ETh (100I1)RC

3 3 100I R 2 k E

(35.71 10 / Ei)(2 10 ) 1 C Th + ETh 71.42Ei – The Thévenin equivalent circuit appears in Fig. 18.38 with the origi-

nal load RL. FIG. 18.37 Determining the Thévenin voltage for the network in RTh

Fig. 18.35. – 2 k + ETh 71.42Ei RL 1 k VL

+ –

FIG. 18.38 The Thévenin equivalent circuit for the network in Fig. 18.35.

Output Voltage VL R E 1 k 71.42E L Th 1 21 i 2 VL RL RTh 1 k 2 k and VL 23.81Ei revealing that the output voltage is 23.81 times the applied voltage with a phase shift of 180° due to the minus sign.

Dependent Sources For dependent sources with a controlling variable not in the network un- der investigation, the procedure indicated above can be applied. However, for dependent sources of the other type, where the controlling variable is part of the network to which the theorem is to be applied, another ap- proach must be used. The necessity for a different approach is demon- strated in an example to follow. The method is not limited to dependent boy30444_ch18.qxd 3/24/06 2:55 PM Page 800

800 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

sources of the latter type. It can also be applied to any dc or sinusoidal ac Z Th network. However, for networks of independent sources, the method of + application used in Chapter 9 and presented in the first portion of this sec-

ETh tion is generally more direct, with the usual savings in time and errors. – The new approach to Thévenin’s theorem can best be introduced at this stage in the development by considering the Thévenin equivalent cir- cuit in Fig. 18.39(a). As indicated in Fig. 18.39(b), the open-circuit ter- (a) minal voltage (Eoc) of the Thévenin equivalent circuit is the Thévenin equivalent voltage; that is, ZTh + + Eoc ETh (18.1)

ETh Eoc = ETh – If the external terminals are short circuited as in Fig. 18.39(c), the result- – ing short-circuit current is determined by

(b) E Th Isc (18.2) ZTh ZTh

+ or, rearranged, ETh ETh Isc = ZTh E – Th ZTh Isc

(c) E oc and ZTh (18.3) FIG. 18.39 Isc Defining an alternative approach for determining the Thévenin impedance. Eqs. (18.1) and (18.3) indicate that for any linear bilateral dc or ac net- work with or without dependent sources of any type, if the open-circuit terminal voltage of a portion of a network can be determined along with the short-circuit current between the same two terminals, the Thévenin equivalent circuit is effectively known. A few examples will make the method quite clear. The advantage of the method, which was stressed ear- I g lier in this section for independent sources, should now be more obvious.

+ The current Isc , which is necessary to find ZTh , is in general more diffi- ZTh cult to obtain since all of the sources are present. Network Eg There is a third approach to the Thévenin equivalent circuit that is also useful from a practical viewpoint. The Thévenin voltage is found as in the – two previous methods. However, the Thévenin impedance is obtained by applying a source of voltage to the terminals of interest and determining FIG. 18.40 the source current as indicated in Fig. 18.40. For this method, the source voltage of the original network is set to zero. The Thévenin impedance is Determining ZTh using the approach ZTh Eg / Ig. then determined by the following equation:

XC R1 E g

+ ZTh (18.4)

Ig –

V R + 2 Note that for each technique, ETh Eoc , but the Thévenin impedance is found in different ways. – Thévenin EXAMPLE 18.10 Using each of the three techniques described in this FIG. 18.41 section, determine the Thévenin equivalent circuit for the network in Example 18.10. Fig. 18.41. boy30444_ch18.qxd 3/24/06 2:56 PM Page 801

Th THÉVENIN’S THEOREM ⏐⏐⏐ 801

Solution: Since for each approach the Thévenin voltage is found in ex- XC R1 actly the same manner, it is determined first. From Fig. 18.41, where I XC 0, Due to the polarity for V and defined terminal polarities R2 ZTh R mV MR V V E E 21 2 2 R1 Th oc R1 R2 R1 R2 The following three methods for determining the Thévenin impedance appear in the order in which they were introduced in this section. FIG. 18.42 Determining the Thévenin impedance for the Method 1: See Fig. 18.42. network in Fig. 18.41. ZTh R1 R2 j XC XC R1 Method 2: See Fig. 18.43. Converting the voltage source to a current source (Fig. 18.44), we have (current divider rule) Isc – mV R R mV 1 2 V R2 Isc R R 1 1 2 2 R R R a R b + I 1 1 2 1 sc R1 R2 j XC R1 R2 j XC 1 2 1 2 mR2V R1 R2 FIG. 18.43 R R j X 1 1 2 2 C Determining the short-circuit current for the network and in Fig. 18.41. mR2V XC E R R oc 1 2 1 ZTh Isc mR2V 1 Isc R R R R j X 1 2 1 1 2 2 C V R1 R2 Isc R1 R2 j XC R1 1 2 R1 R2 j XC Method 3: See Fig. 18.45. E g FIG. 18.44 Ig R1 R2 j XC Converting the voltage source in Fig. 18.43 to a 1 2 current source. E g and ZTh R1 R2 j XC Ig XC R1 Ig In each case, the Thévenin impedance is the same. The resulting Thévenin equivalent circuit is shown in Fig. 18.46. + R2 Eg ZTh = R1 R2 – jXC –

+ ZTh – R2V FIG. 18.45 ETh = Thévenin R1 + R2 Determining the Thévenin impedance for the + network in Fig. 18.41 using the approach – ZTh Eg /Ig .

FIG. 18.46 The Thévenin equivalent circuit for the network in Fig. 18.41. boy30444_ch18.qxd 3/24/06 2:56 PM Page 802

802 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

XC EXAMPLE 18.11 Repeat Example 18.10 for the network in Fig. 18.47.

Solution: From Fig. 18.47, ETh is hR R I hI R R 1 2 1 2 ETh Eoc hI R1 R2 1 2 R1 R2 Method 1: See Fig. 18.48. Thévenin ZTh R1 R2 j XC FIG. 18.47 Note the similarity between this solution and that obtained for the previ- Example 18.11. ous example.

XC Method 2: See Fig. 18.49.

R R hI 1 2 Isc 1 2 R R j X 1 1 2 2 C R1 R2 ZTh = R1 R2 – jXC E hI R R oc 1 2 and ZTh 1 2 R1 R2 j XC I R R hI sc 1 1 2 2 R R j X 1 1 2 2 C Method 3: See Fig. 18.50. FIG. 18.48 Determining the Thévenin impedance for the E g network in Fig. 18.47. Ig R R j X 1 1 2 2 C E XC g and ZTh R1 R2 j XC Ig

Isc R R I hI 1 2 sc The following example has a dependent source that will not permit the use of the method described at the beginning of this section for inde- pendent sources. All three methods will be applied, however, so that the results can be compared.

FIG. 18.49 Determining the short-circuit current for the network EXAMPLE 18.12 For the network in Fig. 18.51 (introduced in Exam- in Fig. 18.47. ple 18.6), determine the Thévenin equivalent circuit between the indi- cated terminals using each method described in this section. Compare XC Ig your results.

V – + + R1 R2 Eg – +

I R1 V – ZTh Thévenin FIG. 18.50 Determining the Thévenin impedance using the approach ZTh Eg / Ig. FIG. 18.51 Example 18.12.

Solution: First, using Kirchhoff’s voltage law, ETh (which is the same for each method) is written ETh V mV (1 m)V boy30444_ch18.qxd 3/24/06 2:56 PM Page 803

Th THÉVENIN’S THEOREM ⏐⏐⏐ 803

However, V IR1 V = 0 – + so ETh (1 M)IR1

ZTh + R1 V = 0 Method 1: See Fig. 18.52. Since I 0, V and mV 0, and – ZTh R1 (incorrect) ZTh Method 2: See Fig. 18.53. Kirchhoff’s voltage law around the indicated loop gives us FIG. 18.52 V mV 0 Determining ZTh incorrectly. and V(1 m) 0 V – + Since m is a positive constant, the above equation can be satisfied only when V 0. Substitution of this result into Fig. 18.53 yields the config- I uration in Fig. 18.54, and + sc I R1 V Isc Isc I – with E 1 m IR1 Z oc 1 2 1 M R (correct) Th I I 1 2 1 sc FIG. 18.53 Method 3: See Fig. 18.55. Determining Isc for the network in Fig. 18.51. Eg V mV (1 m)V I = 0 1 Isc Eg or V + m 1 I R1 V = 0 Isc –

V Eg and I g R 1 m R 1 1 2 1 E g FIG. 18.54 andZTh 1 M R1 (correct) Ig 1 2 Substituting V 0 into the network in Fig. 18.53. The Thévenin equivalent circuit appears in Fig. 18.56, and V I 1 M R I – + g I 1 2 1 L R 1 M R L 1 2 1 which compares with the result in Example 18.6. + R1 V Eg – (1 + m)R1 Z IL Th + FIG. 18.55 ETh = (1 + m)IR1 RL Determining Z using the approach Z E / I . – Th Th g g

FIG. 18.56 The Thévenin equivalent circuit for the network in Fig. 18.51.

The network in Fig. 18.57 is the basic configuration of the transistor equivalent circuit applied most frequently today (although most texts in boy30444_ch18.qxd 3/24/06 2:56 PM Page 804

804 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

R1

+ I + + Thévenin

Vi k1V2 k2I R2 V2 – – –

FIG. 18.57 Example 18.13: Transistor equivalent network.

electronics use the circle rather than the diamond outline for the source). Obviously, it is necessary to know its characteristics and to be adept in its use. Note that there are both a controlled voltage and a controlled current source, each controlled by variables in the configuration.

EXAMPLE 18.13 Determine the Thévenin equivalent circuit for the in- dicated terminals of the network in Fig. 18.57. Solution: Apply the second method introduced in this section.

ETh Eoc V2 V k V V k E I i 1 2 i 1 oc R1 R1 V k E i 1 oc and Eoc k2IR2 k2R2 a R1 b k R V k k R E 2 2 i 1 2 2 oc R1 R1 k k R k R V 1 2 2 2 2 i or Eoc 1 a R1 b R1 R k k R k R V 1 1 2 2 2 2 i and Eoc a R1 b R1

k R V 2 2 i so Eoc ETh (18.5) R1 k1k2R2

Isc For the network in Fig. 18.58, where

R1

+ I Isc

I Vi k2I R2 sc

–

FIG. 18.58

Determining Isc for the network in Fig. 18.57. boy30444_ch18.qxd 3/24/06 2:57 PM Page 805

Th THÉVENIN’S THEOREM ⏐⏐⏐ 805

V i V2 0 k1V2 0 I R1 k V 2 i and Isc k2I R1 k2R2Vi E R k k R oc 1 1 2 2 so ZTh Isc k2Vi

R1

R R 1 2 and ZTh (18.6) R1 k1k2R2

Frequently, the approximation k1 0 is applied. Then the Thévenin voltage and impedance are

k R V 2 2 i ETh k1 0 (18.7) R1

ZTh R2 k1 0 (18.8)

Apply ZTh Eg /Ig to the network in Fig. 18.59, where k V I 1 2 R1

R1 Ig

I + +

k1V2 k2I R2 ZTh Eg – –

FIG. 18.59 Determining ZTh using the procedure ZTh Eg / Ig .

But V2 Eg k1Eg so I R1 Applying Kirchhoff’s current law, we have E k E E g 1 g g Ig k2I k2 R2 a R1 b R2 k k 1 1 2 Eg a R2 R1 b boy30444_ch18.qxd 3/24/06 2:57 PM Page 806

806 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Ig R k k R and 1 1 2 2 Eg R1R2

Eg R R 1 2 or ZTh Ig R1 k1k2R2 as obtained above.

The last two methods presented in this section were applied only to networks in which the magnitudes of the controlled sources were depen- dent on a variable within the network for which the Thévenin equivalent circuit was to be obtained. Understand that both of these methods can also be applied to any dc or sinusoidal ac network containing only inde- pendent sources or dependent sources of the other kind.

18.4 NORTON’S THEOREM The three methods described for Thévenin’s theorem will each be altered to permit their use with Norton’s theorem. Since the Thévenin and Nor- ton impedances are the same for a particular network, certain portions of the discussion are quite similar to those encountered in the previous sec- tion. We first consider independent sources and the approach developed in Chapter 9, followed by dependent sources and the new techniques de-

IN ZN veloped for Thévenin’s theorem. You will recall from Chapter 9 that Norton’s theorem allows us to re- place any two-terminal linear bilateral ac network with an equivalent cir- cuit consisting of a current source and an impedance, as in Fig. 18.60. The Norton equivalent circuit, like the Thévenin equivalent circuit, FIG. 18.60 is applicable at only one frequency since the reactances are frequency The Norton equivalent circuit for ac networks. dependent.

Independent Sources The procedure outlined below to find the Norton equivalent of a sinu- soidal ac network is changed (from that in Chapter 9) in only one respect: the replacement of the term resistance with the term impedance. 1. Remove that portion of the network across which the Norton equivalent circuit is to be found. 2. Mark (, •, and so on) the terminals of the remaining two-terminal network.

3. Calculate ZN by first setting all voltage and current sources to zero (short circuit and open circuit, respectively) and then finding the resulting impedance between the two marked terminals.

4. Calculate IN by first replacing the voltage and current sources and then finding the short-circuit current between the marked terminals. 5. Draw the Norton equivalent circuit with the portion of the circuit previously removed replaced between the terminals of the Norton equivalent circuit. The Norton and Thévenin equivalent circuits can be found from each other by using the source transformation shown in Fig. 18.61. The source transformation is applicable for any Thévenin or Norton equivalent cir- cuit determined from a network with any combination of independent or dependent sources. boy30444_ch18.qxd 3/24/06 2:57 PM Page 807

Th NORTON’S THEOREM ⏐⏐⏐ 807

ZTh = ZN

ZTh

+ E Th Z Z = Z IN = N N Th ETh = INZN ZTh –

FIG. 18.61 Conversion between the Thévenin and Norton equivalent circuits.

EXAMPLE 18.14 Determine the Norton equivalent circuit for the net- work external to the 6 resistor in Fig. 18.62.

R1 XL

4 + 3 ∠ ° E = 20 V 0 XC 5 RL 6 – Z1

Norton +

E Z2 FIG. 18.62 – Example 18.14.

Solution: Norton Steps 1 and 2 (Fig. 18.63): FIG. 18.63 Assigning the subscripted impedances to the network Z1 R1 j XL 3 j 4 5 53.13° in Fig. 18.62. Z2 j XC j 5

Step 3 (Fig. 18.64): Z1 Z Z 5 53.13° 5 90° 25 36.87° Z 1 2 1 21 2 N Z Z1 Z2 3 j 4 j 5 3 j 1 2 ZN 25 36.87° 7.91 18.44° 7.50 j 2.50 3.16 18.43°

Step 4 (Fig. 18.65): FIG. 18.64 E 20 V 0° Determining the Norton impedance for the network IN I1 4 A 53.13° Z1 5 53.13° in Fig. 18.62.

Z1 I I1 N +

E Z2 IN –

FIG. 18.65

Determining IN for the network in Fig. 18.62. boy30444_ch18.qxd 3/24/06 2:57 PM Page 808

808 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Step 5: The Norton equivalent circuit is shown in Fig. 18.66.

7.50 – j2.50 R 7.50 ∠ ° ZN R 6 ∠ ° R 6 IN = 4 A – 53.13 IN = 4 A – 53.13 L XC 2.50

FIG. 18.66 The Norton equivalent circuit for the network in Fig. 18.62.

EXAMPLE 18.15 Find the Norton equivalent circuit for the network external to the 7 capacitive reactance in Fig. 18.67.

XL R 2 5 1 R 2 1 ∠ ° XC = 7 I = 3 A 0 2 X 4 C1

FIG. 18.67 Example 18.15.

Solution: Steps 1 and 2 (Fig. 18.68): Z R j X j 1 1 C1 2 4 Z2 R2 1 Z3 j XL j 5

Z3

Z2

∠ ° Z1 I = 3 A 0

FIG. 18.68 Assigning the subscripted impedances to the network in Fig. 18.67.

Step 3 (Fig. 18.69): Z3 Z1 Z 2 Z 1 2 N Z Z Z 3 1 1 2 2 Z1 Z2 2 j 4 1 3 j 4 5 53.13° 5 90° 5 53.13° 25 36.87° Z 1 21 2 N j 5 3 j 4 3 j 1 boy30444_ch18.qxd 3/24/06 2:57 PM Page 809

Th NORTON’S THEOREM ⏐⏐⏐ 809

25 36.87° 3.16 18.43° ZN 7.91 18.44° 7.50 j 2.50

Z3 Z Z2 2

Z3 ZN ZN Z1 Z1

FIG. 18.69 Finding the Norton impedance for the network in Fig. 18.67.

Calculator Solution: Performing the above on the TI-89 calculator results in the sequence in Fig. 18.70:

((5i(24i1)))((5i24i1)) Polar 7.91E0∠18.43E0

FIG. 18.70

Step 4 (Fig. 18.71): Z I 1 IN I1 (current divider rule) Z1 Z2 2 j 4 3 A 6 A j 12 A 13.4 A 63.43° 1 21 2 3 j 4 5 53.13° 5 53.13° IN 2.68 A 10.3°

Z3

Z2 I1

IN ∠ ° Z1 I = 3 A 0

FIG. 18.71

Determining IN for the network in Fig. 18.67.

Step 5: The Norton equivalent circuit is shown in Fig. 18.72.

7.50 + j2.50 R 7.50

∠ ° ZN X 7 ∠ ° X 7 IN = 2.68 A – 10.3 C2 IN = 2.68 A – 10.3 C2 XL 2.50

FIG. 18.72 The Norton equivalent circuit for the network in Fig. 18.67. boy30444_ch18.qxd 3/24/06 2:57 PM Page 810

810 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

ZTh EXAMPLE 18.16 Find the Thévenin equivalent circuit for the network + ZTh = ZN external to the 7 capacitive reactance in Fig. 18.67. ETh INZN Solution: Using the conversion between sources (Fig. 18.73), we obtain – ZTh ZN 7.50 j 2.50 ETh INZN (2.68 A 10.3°)(7.91 18.44°) 21.2 V 8.14° FIG. 18.73 Determining the Thévenin equivalent circuit for the The Thévenin equivalent circuit is shown in Fig. 18.74. Norton equivalent in Fig. 18.72.

R XL Dependent Sources + 7.50 2.50 As stated for Thévenin’s theorem, dependent sources in which the con- E 21.2 V ∠ 8.14° X 7 trolling variable is not determined by the network for which the Norton Th C2 – equivalent circuit is to be found do not alter the procedure outlined above. For dependent sources of the other kind, one of the following proce- dures must be applied. Both of these procedures can also be applied to networks with any combination of independent sources and dependent FIG. 18.74 sources not controlled by the network under investigation. The Thévenin equivalent circuit for the network in The Norton equivalent circuit appears in Fig. 18.75(a). In Fig. 18.75(b), Fig. 18.67. we find that

I = 0 +

IN ZN IN ZN Isc IN ZN Eoc = INZN

–

(a) (b) (c)

FIG. 18.75

Defining an alternative approach for determining ZN .

Isc IN (18.9)

and in Fig. 18.75(c) that Eoc INZN Or, rearranging, we have E oc ZN IN

I E g oc and ZN (18.10) Isc + Network ZN Eg The Norton impedance can also be determined by applying a source

– of voltage Eg to the terminals of interest and finding the resulting Ig , as shown in Fig. 18.76. All independent sources and dependent sources not controlled by a variable in the network of interest are set to zero, and

FIG. 18.76 E Determining the Norton impedance using the g ZN (18.11) approach ZN Eg / Ig . Ig boy30444_ch18.qxd 3/24/06 2:58 PM Page 811

Th NORTON’S THEOREM ⏐⏐⏐ 811

For this latter approach, the Norton current is still determined by the I short-circuit current.

R1 R2 EXAMPLE 18.17 Using each method described for dependent sources, + find the Norton equivalent circuit for the network in Fig. 18.77. E hI – Solution:

IN For each method, IN is determined in the same manner. From Fig. 18.78 using Kirchhoff’s current law, we have Norton 0 I hI Isc FIG. 18.77 Example 18.17. or Isc (1 h)I

Applying Kirchhoff’s voltage law gives us I + V – R2 E IR1 Isc R2 0 R R 1 2 Isc and IR1 Isc R2 E + I R E sc 2 E hI Isc or I – R1

I R E sc 2 so Isc 1 h I 1 h 1 2 1 2a R1 b FIG. 18.78 or R1Isc 1 h Isc R2 1 h E 1 2 1 2 Determining Isc for the network in Fig. 18.77. I R 1 h R 1 h E sc 3 1 1 2 2 4 1 2 1 h E I 1 2 I sc R 1 h R N 1 1 2 2 ZN

Method 1: Eoc is determined from the network in Fig. 18.79. By Kirch- I V = 0 hoff’s current law, + – + 0 I hI or 1(h 1) 0 R1 + For h, a positive constant I must equal zero to satisfy the above. Therefore, E hI Eoc I 0 and hI 0 – – and Eoc E E E R 1 h R oc 1 1 2 2 with ZN FIG. 18.79 Isc 1 h E 1 h 1 2 1 2 Determining Eoc for the network in Fig. 18.77. R 1 h R 1 1 2 2 Method 2: Note Fig. 18.80. By Kirchhoff’s current law, – V + – V + I R1 I R2 g Ig I hI (I h)I R R By Kirchhoff’s voltage law, 1 2 + E I R IR g g 2 1 0 hI Eg ZN – Eg Ig R2 or I R1 Substituting, we have E I R FIG. 18.80 g g 2 Ig 1 h I 1 h Determining the Norton impedance using the 1 2 1 2a R b 1 approach ZN Eg / Eg. and Ig R1 (1 h)Eg (1 h)Ig R2 boy30444_ch18.qxd 3/24/06 2:58 PM Page 812

812 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

so Eg(1 h) Ig[R1 (1 h)R2] E R 1 h R g 1 1 2 2 or ZN Ig 1 h which agrees with the above.

EXAMPLE 18.18 Find the Norton equivalent circuit for the network configuration in Fig. 18.57. Solution: By source conversion, k2 R2Vi E R k k R Th 1 1 2 2 IN ZTh R1R2 R1 k1k2R2

k V 2 i and IN (18.12) R1

which is Isc as determined in Example 18.13, and

R Z Z 2 (18.13) N Th k k R 1 1 2 2 R1 For k1 0, we have

k V 2 i IN k1 0 (18.14) R1

ZN R2 k1 0 (18.15)

18.5 MAXIMUM POWER TRANSFER THEOREM When applied to ac circuits, the maximum power transfer theorem states that maximum power will be delivered to a load when the load impedance is the conjugate of the Thévenin impedance across its terminals. That is, for Fig. 18.81, for maximum power transfer to the load,

ZTh Z ∠ v Th Thz E = E ∠ v ∠ Th Th Ths ZL = ZL vL

FIG. 18.81 Defining the conditions for maximum power transfer to a load. boy30444_ch18.qxd 3/24/06 2:58 PM Page 813

Th MAXIMUM POWER TRANSFER THEOREM ⏐⏐⏐ 813

Z Z u u L Th and L ThZ (18.16)

or, in rectangular form,

RL RTh and j Xload j XTh (18.17)

The conditions just mentioned will make the total impedance of the cir- cuit appear purely resistive, as indicated in Fig. 18.82: ZT (R jX) (R j X)

and ZT 2R (18.18)

± ZTh = RTh jXTh

I +

E = E ∠ v Z = R ± jX Th Th Ths L – ZT

FIG. 18.82

Conditions for maximum power transfer to ZL.

Since the circuit is purely resistive, the power factor of the circuit un- der maximum power conditions is 1; that is,

Fp 1 (maximum power transfer) (18.19)

The magnitude of the current I in Fig. 18.82 is E E I Th Th ZT 2R The maximum power to the load is

E 2 P I 2R Th R max a 2R b

2 ETh and P (18.20) max 4R

R XC EXAMPLE 18.19 Find the load impedance in Fig. 18.83 for maximum power to the load, and find the maximum power. 6 8 + Solution: Determine ZTh [Fig. 18.84(a)]: E = 9 V ∠ 0° XL 8 ZL Z 1 R j XC 6 j 8 10 53.13° – Z 2 j XL j 8 Z Z 10 53.13° 8 90° 80 36.87° 1 2 1 21 2 ZTh Z1 Z2 6 j 8 j 8 6 0° FIG. 18.83 13.33 36.87° 10.66 j 8 Example 18.19. boy30444_ch18.qxd 3/24/06 2:58 PM Page 814

814 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Z1 Z1

+ + Z2 ZTh E Z2 ETh – –

(a) (b)

FIG. 18.84

Determining (a) ZTh and (b) ETh for the network external to the load in Fig. 18.83.

and ZL 13.3 36.87° 10.66 j 8

XL XL To find the maximum power, we must first find ETh [Fig. 18.84(b)], as 9 9 + follows: E = 9 ∠ ° 10 V 0 Z E 2 – XL ETh (voltage divider rule) R Z2 Z1 ZL 8 8 90° 9 V 0° 72 V 90° 1 21 2 12 V 90° j 8 6 j 8 6 0° E 2 12 V 2 Th 144 Then Pmax 1 2 3.38 W FIG. 18.85 4R 4 10.66 42.64 1 2 Example 18.20. EXAMPLE 18.20 Find the load impedance in Fig. 18.85 for maximum 2 power to the load, and find the maximum power.

Solution: First we must find ZTh (Fig. 18.86). Z1 Z 1 Z1 j XL j 9 Z2 R 8

1 Z1 3 Converting from a to a Y (Fig. 18.87), we have Z Z 1 2 Z 1 j 3 Z2 8 ZTh 3 The redrawn circuit (Fig. 18.88) shows Z 1 Z 1 Z2 FIG. 18.86 Z Z 1 2 Th 1 Z Z Z Defining the subscripted impedances for the network 1 1 1 2 2 in Fig. 18.85. 3 90° j 3 8 j 3 1 2 j 6 8 2

Z 1 Z 1

Z 1 Z 1 Z 1 Z 1 1 3 ZTh

Z2 ZTh Z2

FIG. 18.87 Substituting the Y equivalent for the upper FIG. 18.88

configuration in Fig. 18.86. Determining ZTh for the network in Fig. 18.85. boy30444_ch18.qxd 3/24/06 2:59 PM Page 815

Th MAXIMUM POWER TRANSFER THEOREM ⏐⏐⏐ 815

3 90° 8.54 20.56° j 3 1 21 2 10 36.87° 25.62 110.56° j 3 j 3 2.56 73.69° 10 36.87° j 3 0.72 j 2.46 ZTh 0.72 j 5.46 and ZL 0.72 j 5.46

For ETh , use the modified circuit in Fig. 18.89 with the voltage source I1 = 0 replaced in its original position. Since I1 0, ETh is the voltage across the Z 1 + series impedance of Z 1 and Z2 . Using the voltage divider rule gives us Z Z Z Z E j 3 8 10 V 0° 1 1 1 2 ETh 1 2 1 21 2 Z Z Z 8 j 6 ETh 1 2 1 + 8.54 20.56° 10 V 0° 1 21 2 Z2 E 10 36.87° – – ETh 8.54 V 16.31° E2 8.54 V 2 Th 72.93 and Pmax 1 2 W FIG. 18.89 4R 4 0.72 2.88 1 2 Finding the Thévenin voltage for the network in 25.32 W Fig. 18.85.

If the load resistance is adjustable but the magnitude of the load reac- tance cannot be set equal to the magnitude of the Thévenin reactance, then the maximum power that can be delivered to the load will occur when the load reactance is made as close to the Thévenin reactance as possible and the load resistance is set to the following value:

R 2R 2 X X 2 (18.21) L Th 1 Th load 2 where each reactance carries a positive sign if inductive and a negative sign if capacitive. The power delivered will be determined by

P E 2 4R (18.22) Th> av

RTh RL where R (18.23) av 2

The derivation of the above equations is given in Appendix G of the text. The following example demonstrates the use of the above.

EXAMPLE 18.21 For the network in Fig. 18.90:

a. Determine the value of RL for maximum power to the load if the load reactance is fixed at 4 . b. Find the power delivered to the load under the conditions of part (a). c. Find the maximum power to the load if the load reactance is made adjustable to any value, and compare the result to part (b) above. boy30444_ch18.qxd 3/24/06 2:59 PM Page 816

816 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

RTh XTh

4 7

+ RL ∠ ° ETh = 20 V 0 – XC = 4

FIG. 18.90 Example 18.21.

Solutions:

a. Eq. (18.21): R 2R 2 X X 2 L Th 1 Th load 2 2 4 2 7 4 2 1 2 1 2 216 9 225 RL 5 R R 4 5 b. Eq. (18.23): R Th L av 2 2 4.5

E 2 Eq. (18.22): P Th 4Rav 20 V 2 400 1 2 W 4 4.5 18 1 2 22.22 W c. For ZL 4 j 7 , E 2 20 V 2 P Th 1 2 max 4R 4 4 Th 1 2 25 W exceeding the result of part (b) by 2.78 W.

18.6 SUBSTITUTION, RECIPROCITY, AND MILLMAN’S THEOREMS As indicated in the introduction to this chapter, the substitution and reci- procity theorems and Millman’s theorem will not be considered here in detail. A careful review of Chapter 9 will enable you to apply these the- orems to sinusoidal ac networks with little difficulty. A number of prob- lems in the use of these theorems appear in the Problems section at the end of the chapter.

18.7 APPLICATION Electronic Systems One of the blessings in the analysis of electronic systems is that the superposition theorem can be applied so that the dc analysis and ac analy- boy30444_ch18.qxd 3/24/06 2:59 PM Page 817

Th APPLICATION ⏐⏐⏐ 817

sis can be performed separately. The analysis of the dc system will affect the ac response, but the analysis of each is a distinct, separate process. Even though electronic systems have not been investigated in this text, a number of important points can be made in the description to follow that support some of the theory presented in this and recent chapters, so in- clusion of this description is totally valid at this point. Consider the net- work in Fig. 18.91 with a transistor power amplifier, an 8 speaker as the load, and a source with an internal resistance of 800 . Note that each component of the design was isolated by a color box to emphasize the fact that each component must be carefully weighed in any good design.

VCC = 22 V

RC 100 C C RB 47 k C

0.1 μ F Rs CC B β = 200 + 0.1 μ F RL 8 Vs 1V(p-p) E –

Source Amplifier Load

FIG. 18.91 Transistor amplifier.

As mentioned above, the analysis can be separated into a dc and an ac + + component. For the dc analysis, the two capacitors can be replaced by an VCC 22 V VCC 22 V open-circuit equivalent (Chapter 10), resulting in an isolation of the am- – RC 100 – RB 47 k plifier network as shown in Fig. 18.92. Given the fact that VBE will be C IC about 0.7 V dc for any operating transistor, the base current I can be + B B β = 200 found as follows using Kirchhoff’s voltage law: V + – CE IB V VR V V 22 V 0.7 V BE B CC BE M – IB 453.2 A E RB RB 47 k

For transistors, the collector current IC is related to the base current by IC bIB , and FIG. 18.92 dc equivalent of the transistor network in Fig. 18.91. IC bIB (200)(453.2 mA) 90.64 mA Finally, through Kirchhoff’s voltage law, the collector voltage (also the collector-to-emitter voltage since the emitter is grounded) can be de- termined as follows: VC VCE VCC IC RC 22 V (90.64 mA)(100 ) 12.94 V For the dc analysis, therefore, IB 453.2 MA IC 90.64 mA VCE 12.94 V which will define a point of dc operation for the transistor. This is an im- portant aspect of electronic design since the dc operating point will have an effect on the ac gain of the network. boy30444_ch18.qxd 3/24/06 2:59 PM Page 818

818 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

Now, using superposition, we can analyze the network from an ac view- point by setting all dc sources to zero (replaced by ground connections) RC 100 R 47 k and replacing both capacitors by short circuits as shown in Fig. 18.93. Sub- B C stituting the short-circuit equivalent for the capacitors is valid because at β = 200 Rs 10 kHz (the midrange for human hearing response), the reactance of the B capacitor is determined by XC 1/2p fC 15.92 which can be ignored RL 8 800 when compared to the series resistors at the source and load. In other Vs 1V(p-p) E words, the capacitor has played the important role of isolating the ampli- fier for the dc response and completing the network for the ac response. Redrawing the network as shown in Fig. 18.94(a) permits an ac in- FIG. 18.93 vestigation of its response. The transistor has now been replaced by an ac equivalent of the transistor network in Fig. 18.91. equivalent network that represents the behavior of the device. This process will be covered in detail in your basic electronics courses. This transistor configuration has an input impedance of 200 and a current source whose magnitude is sensitive to the base current in the input cir- cuit and to the amplifying factor for this transistor of 200. The 47 k re- sistor in parallel with the 200 input impedance of the transistor can be

ignored, so the input current Ii and base current Ib are determined by V 1 V p-p 1 V p-p s 1 2 1 2 Ii Ib 1 mA p-p Rs Ri 800 200 1 k 1 2

The collector current IC is then IC bIb (200)(1 mA (p-p)) 200 mA (p-p) and the current to the speaker is determined by the current divider rule as follows: 100 IC I 1 2 0.926I 0.926 200 mA p-p L 100 8 C 1 1 22 185.2 mA p-p 1 2

I R i s B C ≅ + + 800 I 0 A Ib βI b RC 100 Vs 1V(p-p) RB Ri 200 200I RL 8 b VL 47 kΩ – – E IC IL Transistor equivalent circuit (a)

100 mA 100 100 100 mA + 8 VL 100 100 – 200 mA Impedance matching transformer (b)

FIG. 18.94 (a) Network in Fig. 18.93 following the substitution of the transistor equivalent network; (b) effect of the matching transformer. boy30444_ch18.qxd 3/24/06 2:59 PM Page 819

Th APPLICATION ⏐⏐⏐ 819

with the voltage across the speaker being VL ILRL (185.2 mA (p-p))(8 ) 1.48 V The power to the speaker is then determined as follows:

VL p-p IL p-p 1.48 V 185.2 mA p-p P V I 1 1 2 21 1 2 2 1 21 1 22 speaker L rms L rms 8 8 34.26 mW which is relatively low. It initially appears that the above was a good design for distribution of power to the speaker because a majority of the collector current went to the speaker. However, you must always keep in mind that power is the product of voltage and current. A high current with a very low voltage results in a lower power level. In this case, the voltage level is too low. However, if we introduce a matching transformer that makes the 8 resistive load “look like” 100 as shown in Fig. 18.94(b), establishing maximum power conditions, the current to the load drops to half of the previous amount because cur- rent splits through equal resistors. But the voltage across the load in- creases to VL ILRL (100 mA (p-p))(100 ) 10 V (p-p) and the power level to

VL p-p IL p-p 10 V 100 mA P 1 1 2 21 1 2 2 1 21 2 125 mW speaker 8 8 which is 3.6 times the gain without the matching transformer. For the 100 load, the dc conditions are unaffected due to the isola-

tion of the capacitor CC , and the voltage at the collector is 12.94 V as shown in Fig. 18.95(a). For the ac response with a 100 load, the out- put voltage as determined above will be 10 V peak-to-peak (5 V peak) as shown in Fig. 18.95(b). Note the out-of-phase relationship with the input

due to the opposite polarity of VL . The full response at the collector ter- minal of the transistor can then be drawn by superimposing the ac re- sponse on the dc response as shown in Fig. 18.95(c) (another application of the superposition theorem). In other words, the dc level simply shifts the ac waveform up or down and does not disturb its shape. The peak-to- peak value remains the same, and the phase relationship is unaltered. The

vc (volts) ac + dc

VC (volts) dc 17.94

12.94 vc (volts) ac 12.94

7.94 5

02TtT 02TtT 02TtT

–5

(a) (b) (c)

FIG. 18.95 Collector voltage for the network in Fig. 18.91: (a) dc; (b) ac; (c) dc and ac. boy30444_ch18.qxd 3/24/06 2:59 PM Page 820

820 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

total waveform at the load will include only the ac response of Fig. 18.95(b) since the dc component has been blocked out by the capacitor. The voltage at the source appears as shown in Fig. 18.96(a), while the voltage at the base of the transistor appears as shown in Fig. 18.96(b) be- cause of the presence of the dc component. A number of important concepts were presented in the above example, with some probably leaving a question or two because of your lack of experience with transistors. However, you should understand that the superposition theorem has the power to permit an isolation of the dc and ac responses and the ability to combine both if the total response is desired.

vb (volts) 1.2

v (volts) s 0.7 0.5 0.2

T 3 T 3 0 –2 Tt–2T 0 –2 Tt–2T

–0.5 (a) (b)

FIG. 18.96 Applied signal: (a) at the source; (b) at the base of the transistor.

18.8 COMPUTER ANALYSIS PSpice Thévenin’s Theorem This application parallels the methods used to determine the Thévenin equivalent circuit for dc circuits. The network in Fig. 18.29 appears as shown in Fig. 18.97 when the open-circuit Thévenin voltage is to be determined. The open circuit is simulated by using a resistor of 1 T (1 million M). The resistor is necessary to es-

tablish a connection between the right side of inductor L2 and ground— nodes cannot be left floating for OrCAD simulations. Since the magnitude and the angle of the voltage are required, VPRINT1 is introduced as shown in Fig 18.97. The simulation was an AC Sweep simulation at

FIG. 18.97 Using PSpice to determine the open-circuit Thévenin voltage. boy30444_ch18.qxd 3/24/06 2:59 PM Page 821

Th COMPUTER ANALYSIS ⏐⏐⏐ 821

1 kHz, and when the Orcad Capture window was obtained, the results appearing in Fig. 18.98 were taken from the listing resulting from the PSpice-View Output File. The magnitude of the Thévenin voltage is 5.187 V to compare with the 5.08 V of Example 18.8, while the phase an- gle is 77.13° to compare with the 77.09° of the same example— excellent results.

FIG. 18.98 The output file for the open-circuit Thévenin voltage for the network in Fig. 18.97.

Next, the short-circuit current is determined using IPRINT as shown in Fig. 18.99, to permit a determination of the Thévenin impedance. The resistance Rcoil of 1 m had to be introduced because inductors cannot be treated as ideal elements when using PSpice; they must all show some se- ries internal resistance. Note that the short-circuit current will pass di- rectly through the printer symbol for IPRINT. Incidentally, there is no need to exit the SCHEMATIC1 developed above to determine the Thévenin voltage. Simply delete VPRINT and R3, and insert IPRINT. Then run a new simulation to obtain the results in Fig. 18.100. The

FIG. 18.99 Using PSpice to determine the short-circuit current.

FIG. 18.100 The output file for the short-circuit current for the network in Fig. 18.99. boy30444_ch18.qxd 3/24/06 2:59 PM Page 822

822 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

magnitude of the short-circuit current is 936.1 mA at an angle of 108.6°. The Thévenin impedance is then defined by

E Th 5.187 V 77.13° ZTh 5.54 31.47° Isc 936.1 mA 108.6° which is an excellent match with 5.49 32.36° obtained in Example 18.8.

VCVS The next application will verify the results in Example 18.12 and provide some practice using controlled (dependent) sources. The net- work in Fig. 18.51, with its voltage-controlled voltage source (VCVS), will have the schematic appearance in Fig. 18.101. The VCVS appears as E in the ANALOG library, with the voltage E1 as the controlling volt- age and E as the controlled voltage. In the Property Editor dialog box, change the GAIN to 20, but leave the rest of the columns as is. After Display-Name and Value, select Apply and exit the dialog box. This re- sults in GAIN 20 near the controlled source. Take particular note of the second ground inserted near E to avoid a long wire to ground that may overlap other elements. For this exercise, the current source ISRC is used because it has an arrow in its symbol, and frequency is not important for this analysis since there are only resistive elements present. In the Property Editor dialog box, set the AC level to 5 mA and the DC level to 0 A; both are displayed using Display-Name and value. VPRINT1 is set up as in past exercises. The resistor Roc (open circuit) was given a very large value so that it appears as an open circuit to the rest of the net- work. VPRINT1 provides the open circuit Thévenin voltage between the points of interest. Running the simulation in the AC Sweep mode at 1 kHz results in the output file appearing in Fig. 18.102, revealing that the Thévenin voltage is 210 V 0°. Substituting the numerical values of this example into the equation obtained in Example 18.12 confirms the result: ETh (1 m)IR1 (1 20) (5 mA 0°)(2 k ) 210 V0°

FIG. 18.101 Using PSpice to determine the open-circuit Thévenin voltage for the network in Fig. 18.51. boy30444_ch18.qxd 3/24/06 2:59 PM Page 823

Th COMPUTER ANALYSIS ⏐⏐⏐ 823

FIG. 18.102 The output file for the open-circuit Thévenin voltage for the network in Fig. 18.101.

Next, determine the short-circuit current using the IPRINT option. Note in Fig. 18.103 that the only difference between this network and that in Fig. 18.102 is the replacement of Roc with IPRINT and the removal of VPRINT1. Therefore, you do not need to completely “redraw” the network. Just make the changes and run a new simulation. The result of the new simulation as shown in Fig. 18.104 is a current of 5 mA at an an- gle of 0°.

FIG. 18.103 Using PSpice to determine the short-circuit current for the network in Fig. 18.51.

FIG. 18.104 The output file for the short-circuit current for the network in Fig. 18.103. boy30444_ch18.qxd 3/24/06 2:59 PM Page 824

824 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

The ratio of the two measured quantities results in the Thévenin impedance: E E oc Th 210 V 0° ZTh 42 k Isc Isc 5 mA 0° which also matches the longhand solution in Example 18.12: ZTh (1 m)R1 (1 20)2 k (21)2 k 42 k

Multisim Superposition This analysis begins with the network in Fig. 18.12 from Example 18.4 because it has both an ac and a dc source. You will find in the analysis to follow that it is not necessary to set up a separate network for each source. Once the network is set up, the dc levels will ap- pear during simulation, and the ac response can be found from a View option. The resulting schematic appears in Fig. 18.105. The construction is quite straightforward with the parameters of the ac source set as follows: Voltage(RMS): 4 V; AC Analysis Magnitude: 4 V; Phase: 0 Degrees; AC Analysis Phase: 0 Degrees; Frequency (F): 1 kHz; Voltage Offset: 0 V; and Time Delay: 0 Seconds. The dc voltmeter Indicator is listed as VOLTMETER_V under Component in the Select a Component dia- log box. Recall that the indicators appear on the keypad on the left edge of the screen that looks like a red 8 LED display. To perform the analysis, use the following sequence to obtain the AC Analysis dialog box: Simulate-Analyses-AC Analysis. In the dialog box, make the following settings under the Frequency Parameters heading: Start frequency: 1 kHz; Stop frequency: 1 kHz; Sweep Type: Decade; Number of points: 1000; Vertical scale: Linear. Then shift to

FIG. 18.105 Using Multisim to apply superposition to the network in Fig. 18.12. boy30444_ch18.qxd 3/24/06 2:59 PM Page 825

Th COMPUTER ANALYSIS ⏐⏐⏐ 825

the Output option and select $4 (note node 4 on the constructed network) under Variables in circuit followed by Add to place it in the Selected variables for analysis column. Move any other variables in the selected list back to the variable list using the Remove option. Then select Simulate, and the Grapher View response of Fig. 18.106 results. Dur- ing the simulation process, the dc solution of 3.6 V appears on the volt- meter display (an exact match with the longhand solution). There are two plots in Fig. 18.106: one of magnitude versus frequency and the other of phase versus frequency. Left-click to select the upper graph, and a red ar- row shows up along the left edge of the plot. The arrow reveals which plot is currently active. To change the label for the vertical axis from Magnitude to Voltage (V) as shown in Fig. 18.106, select the Properties key from the top toolbar and choose Left Axis. Then change the label to Voltage (V) followed by OK, and the label appears as shown in Fig. 18.106. Next, to read the levels indicated on each graph with a high de- gree of accuracy, select the Show/Hide Cursor keypad on the toolbar. The keypad has a small red sine wave with two vertical markers. The re- sult is a set of markers at the left edge of each figure. By selecting a marker from the left edge of the voltage plot and moving it to 1 kHz, you can find the value of the voltage in the accompanying table. Note that at a frequency of 1.006 kHz or essentially 1 kHz, the voltage is 2.06 V which is an exact match with the longhand solution in Example 18.4. If you then drop down to the phase plot, you find at the same frequency that the phase angle is 32.65, which is very close to the 32.74 in the long- hand solution. In general, therefore, the results are an excellent match with the solu- tions in Example 18.4 using techniques that can be applied to a wide va- riety of networks that have both dc and ac sources.

FIG. 18.106 The output results from the simulation of the network in Fig. 18.105. boy30444_ch18.qxd 3/24/06 2:59 PM Page 826

826 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

PROBLEMS SECTION 18.2 Superposition Theorem 1. Using superposition, determine the current through the in-

ductance XL for each network in Fig. 18.107.

I I L L R 3 XC 6 XC 5

I ∠ 6 ° + XL 8 + = 0.3 A 0 XL 8 + ∠ 3 ° ∠ 1 ° ∠ ° E1 = 30 V 0 E2 = 60 V 0 E = 10 V 0 – – –

(a) (b)

FIG. 18.107 Problem 1.

*2. Using superposition, determine the current IL for each net- work in Fig. 18.108.

IL R

X 4 X L C1 XL 3 R 1 ∠ 6 ° IL 3 I = 0.5 A 0 + 6 + + ∠ 12 ° E = 10 V ∠ 90° X 7 I = 0.6 A 0 ∠ ° ∠ ° C2 E1 = 20 V 0 E2 = 120 V 0 – – –

(a) (b)

FIG. 18.108 Problem 2.

*3. Using superposition, find the sinusoidal expression for the 4. Using superposition, find the sinusoidal expression for the

current i for the network in Fig. 18.109. voltage yC for the network in Fig. 18.110.

R R XL 12 V 1 2

4 6 3 + i + E = 10 V ∠0° R 6 ∠ ° 1 1 I 4 A 0 X 1 v C C – R2 8 + – E = 4 V XC 2 2 –

FIG. 18.110 FIG. 18.109 Problems 4, 16, 31, and 43. Problems 3, 15, 30, and 42. boy30444_ch18.qxd 3/24/06 2:59 PM Page 827

Th PROBLEMS ⏐⏐⏐ 827

*5. Using superposition, find the current I for the network in Fig. 18.111. E = 20 V ∠ 0° – +

I R2 5 k I = 5 mA ∠ 0° R 10 k 1 XL 5 k XC 5 k

FIG. 18.111 Problems 5, 17, 32, and 44.

6. Using superposition, determine the current IL (h 100) for E = 10 V ∠ 0° the network in Fig. 18.112. – +

IL I = 2 mA ∠ 0°

hI R 20 k XL 10 k

FIG. 18.112 Problems 6 and 20. 7. Using superposition, for the network in Fig. 18.113, deter- mine the voltage VL (m 20).

X R1 C

5 k + 1 k – + V = 2 V ∠ 0° I = 2 mA ∠ 0°

mV R2 4 k VL – + –

FIG. 18.113 Problems 7, 21, and 35.

*8. Using superposition, determine the current IL for the net- work in Fig. 18.114 (m 20; h 100).

I = 1 mA ∠ 0° R2 I

– + 5 k L + mV

V = 10 V ∠ 0°

hI R1 20 k XL 5 k –

FIG. 18.114 Problems 8, 22, and 36. boy30444_ch18.qxd 3/24/06 2:59 PM Page 828

828 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

*9. Determine VL for the network in Fig. 18.115 (h 50). I

R1 = 2 k + + ∠ ° E = 20 V 53 hI RL 2 k VL – –

FIG. 18.115 Problems 9 and 23.

*10. Calculate the current I for the network in Fig. 18.116.

20V + –

I + I = 1 mA ∠ 0° I = 2 mA ∠ 0° 1 R1 2 k V R2 5 k 2 –

FIG. 18.116 Problems 10, 24, and 38.

11. Find the voltage V for the network in Fig. 18.117. s – Vx R1 10 R2 2 + + I 5 A∠0° V + + s ∠ ° – E1 10 V 0 4Vx – –

FIG. 18.117 Problem 11.

SECTION 18.3 Thévenin’s Theorem 12. Find the Thévenin equivalent circuit for the portions of the networks in Fig. 18.118 external to the elements between points a and b.

R R XL a 3 a 2 k 6 k + + 3 k E = 100 V ∠ 0° XL 4 XC 2 E = 20 V ∠ 0° XC RL – – b b

(a) (b)

FIG. 18.118 Problems 12 and 26. boy30444_ch18.qxd 3/24/06 2:59 PM Page 829

Th PROBLEMS ⏐⏐⏐ 829

*13. Find the Thévenin equivalent circuit for the portions of the networks in Fig. 18.119 external to the elements between points a and b.

XC R1 1 + 2 8 R2 a E ∠ ° R 20 = 50 V 0 1 – 10 X ∠ ° C2 a I = 0.1 A 0 XC 32 R2 = 68 XL XL 20 4 b 6 b

(a) (b)

FIG. 18.119 Problems 13 and 27.

*14. Find the Thévenin equivalent circuit for the portions of the networks in Fig. 18.120 external to the elements between points a and b.

R XC

a 10 8 + ∠ ° I = Z E1 = 120 V 0 XL 8 0.5 A ∠ 60° L – b

(a)

XC R2

a 10 10 + I = ∠ ° 2 ∠ ° I = 0.6 A 90 R1 9 0.8 ∠ 60° E = 20 V 40 – b

(b)

FIG. 18.120 Problems 14 and 28.

*15. a. Find the Thévenin equivalent circuit for the network ex- b. Using the results of part (a), determine the voltage VC for ternal to the resistor R2 in Fig. 18.109. the same figure. b. Using the results of part (a), determine the current i of *17. a. Find the Thévenin equivalent circuit for the network ex- the same figure. ternal to the inductor in Fig. 18.111. 16. a. Find the Thévenin equivalent circuit for the network ex- b. Using the results of part (a), determine the current I of ternal to the capacitor in Fig. 18.110. the same figure. boy30444_ch18.qxd 3/24/06 3:00 PM Page 830

830 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

18. Determine the Thévenin equivalent circuit for the network *24. Find the Thévenin equivalent circuit for the network exter- external to the 5 k inductive reactance in Fig. 18.121 (in nal to the resistor R1 in Fig. 18.116. terms of V). *25. Find the Thévenin equivalent circuit for the network to the left of terminals a-a in Fig. 18.123. XC E R2 – + 1 k a R1 10 k Th 1 k I ∠ ° x 8 V 0 Thévenin – R2 10 k XL 5 k 5Ix R1 2 k R3 3.3 k 20V + a´

FIG. 18.123 FIG. 18.121 Problem 25. Problems 18 and 33.

19. Determine the Thévenin equivalent circuit for the network SECTION 18.4 Norton’s Theorem external to the 4 k inductive reactance in Fig. 18.122 (in 26. Find the Norton equivalent circuit for the network external terms of I). to the elements between a and b for the networks in Fig. 18.118. XC 27. Find the Norton equivalent circuit for the network external to the elements between a and b for the networks in Fig. 18.119. 0.2 k 28. Find the Norton equivalent circuit for the network external to Th the elements between a and b for the networks in Fig. 18.120. 100I X R1 40 k R2 5 k L 4 k *29. Find the Norton equivalent circuit for the portions of the net- works in Fig. 18.124 external to the elements between points a and b. *30. a. Find the Norton equivalent circuit for the network exter-

nal to the resistor R2 in Fig. 18.109. FIG. 18.122 b. Using the results of part (a), determine the current I of Problems 19 and 34. the same figure. *31. a. Find the Norton equivalent circuit for the network exter- 20. Find the Thévenin equivalent circuit for the network exter- nal to the capacitor in Fig. 18.110. nal to the 10 k inductive reactance in Fig. 18.112. b. Using the results of part (a), determine the voltage VC for 21. Determine the Thévenin equivalent circuit for the network the same figure. external to the 4 k resistor in Fig. 18.113. *32. a. Find the Norton equivalent circuit for the network exter- *22. Find the Thévenin equivalent circuit for the network exter- nal to the inductor in Fig. 18.111. nal to the 5 k inductive reactance in Fig. 18.114. b. Using the results of part (a), determine the current I of *23. Determine the Thévenin equivalent circuit for the network the same figure. external to the 2 k resistor in Fig. 18.115.

R1 XL XL a

6 8 a 4 X 8 + R2 9 R1 = 3 C R4 40 ∠ ° Z ∠ ° E 20 V 0 L I2 = 0.4 A 20 + R3 68 +

– X E = 120 V ∠ 30° E = 108 V ∠ 0° C 12 1 R2 8 2 b – – b

(a) (b)

FIG. 18.124 Problem 29. boy30444_ch18.qxd 3/24/06 3:00 PM Page 831

Th PROBLEMS ⏐⏐⏐ 831

33. Determine the Norton equivalent circuit for the network ex- *36. Find the Norton equivalent circuit for the network external ternal to the 5 k inductive reactance in Fig. 18.121. to the 5 k inductive reactance in Fig. 18.114. 34. Determine the Norton equivalent circuit for the network ex- *37. For the network in Fig. 18.125, find the Norton equivalent ternal to the 4 k inductive reactance in Fig. 18.122. circuit for the network external to the 2 k resistor. 35. Find the Norton equivalent circuit for the network external *38. Find the Norton equivalent circuit for the network external to the 4 k resistor in Fig. 18.113. to the I1 current source in Fig. 18.116.

( μ = 20) μV – + R3

4 k + ∠ ° I = 2 mA 0 R1 1 k V R2 3 k R4 2 k –

FIG. 18.125 Problem 37.

SECTION 18.5 Maximum Power Transfer Theorem

39. Find the load impedance ZL for the networks in Fig. 18.126 for maximum power to the load, and find the maximum power to the load.

R1 XL XL

3 4 4 + ∠ ° E = 120 V 0 XC 6 ZL R1 3 R2 2 ZL – I = 2 A ∠ 30°

(a) (b)

FIG. 18.126 Problem 39.

*40. Find the load impedance ZL for the networks in Fig. 18.127 for maximum power to the load, and find the maximum power to the load.

X X L1 L2 XL Z L 4 9 4 R + 1 3 R2 12 ZL E = 60 V ∠ 60° X 5 X 6 + + C1 C2 ∠ ° E = 200 V ∠ 90° – E1 = 100 V 0 2 X 8 R – C – 10

(a) (b)

FIG. 18.127 Problem 40. boy30444_ch18.qxd 3/24/06 3:00 PM Page 832

832 ⏐⏐⏐ NETWORK THEOREMS (ac) Th

41. Find the load impedance RL for the network in Fig. 18.128 45. a. For the network in Fig. 18.129, determine the value of for maximum power to the load, and find the maximum RL that will result in maximum power to the load. power to the load. b. Using the results of part (a), determine the maximum power delivered. I R

+ 50I 2 k LOAD ∠ ° E = 1 V 0 R1 1 k R2 40 k RL XC – + RL 2 k E = 50 V ∠ 0°

– XL 2 k FIG. 18.128 Problem 41.

*42. a. Determine the load impedance to replace the inductor FIG. 18.129 XL in Fig. 18.109 to ensure maximum power to the Problem 45. load. b. Using the results of part (a), determine the maximum power to the load. *46. a. For the network in Fig. 18.130, determine the level of *43. a. Determine the load impedance to replace the capacitor capacitance that will ensure maximum power to the XC in Fig. 18.110 to ensure maximum power to the load. load if the range of capacitance is limited to 1 nF to b. Using the results of part (a), determine the maximum 10 nF. power to the load. b. Using the results of part (a), determine the value of RL *44. a. Determine the load impedance to replace the inductor XL that will ensure maximum power to the load. in Fig. 18.111 to ensure maximum power to the load. c. Using the results of parts (a) and (b), determine the max- b. Using the results of part (a), determine the maximum imum power to the load. power to the load.

R L LOAD

+ 1 k 30 mH RL E = 2 V ∠ 0° C 4 nF

– C (1 – 10 nF)

f = 10 kHz

FIG. 18.130 Problem 46.

SECTION 18.6 Substitution, Reciprocity, and Millman’s Theorems a 47. For the network in Fig. 18.131, determine two equivalent ∠ ° branches through the substitution theorem for the branch a-b. I = 4 mA 0 R1 4 k R2 8 k

b

FIG. 18.131 Problem 47. boy30444_ch18.qxd 3/24/06 3:00 PM Page 833

Th GLOSSARY ⏐⏐⏐ 833

48. a. For the network in Fig. 18.132(a), find the current I. b. Repeat part (a) for the network in Fig. 18.132(b). c. Do the results of parts (a) and (b) compare?

R1 R1 I – + 4 k 4 k E = 20 V ∠ 0° ∠ ° 6 k + E = 20 V 0 1 k R3 I R 1 k R3 5 R5 = 6 k – R4 R4

11 k 11 k R2 R2

8 k 8 k (a) (b)

FIG. 18.132 Problem 48.

49. Using Millman’s theorem, determine the current through the Millman’s theorem A method using voltage-to-current source 4 k capacitive reactance of Fig. 18.133. conversions that will permit the determination of unknown variables in a multiloop network. Norton’s theorem A theorem that permits the reduction of any I two-terminal linear ac network to one having a single current R 2 k X 4 k C

1 L source and parallel impedance. The resulting configuration + – XC 4 k can then be used to determine a particular current or voltage in ∠ ° E = 50 V ∠ 360°

E1 = 100 V 0 2 the original network or to examine the effects of a specific por- – + tion of the network on a particular variable. Reciprocity theorem A theorem stating that for single-source networks, the magnitude of the current in any branch of a net- FIG. 18.133 work, due to a single voltage source anywhere else in the net- Problem 49. work, will equal the magnitude of the current through the branch in which the source was originally located if the source is placed in the branch in which the current was originally SECTION 18.8 Computer Analysis measured. PSpice or Multisim Substitution theorem A theorem stating that if the voltage across and current through any branch of an ac bilateral net- 50. Apply superposition to the network in Fig. 18.6. That is, de- work are known, the branch can be replaced by any combina- termine the current I due to each source, and then find the tion of elements that will maintain the same voltage across and resultant current. current through the chosen branch. *51. Determine the current IL for the network in Fig. 18.22 using Superposition theorem A method of network analysis that per- schematics. mits considering the effects of each source independently. The

*52. Using schematics, determine V2 for the network in Fig. resulting current and/or voltage is the phasor sum of the cur- 4 18.57 if Vi 1 V 0°, R1 0.5 k , k1 3 10 , k2 50, rents and/or voltages developed by each source independently. and R2 20 k . Thévenin’s theorem A theorem that permits the reduction of any *53. Find the Norton equivalent circuit for the network in Fig. two-terminal linear ac network to one having a single voltage 18.77 using schematics. source and series impedance. The resulting configuration can then be employed to determine a particular current or voltage *54. Using schematics, plot the power to the R-C load in Fig. in the original network or to examine the effects of a specific 18.90 for values of R from 1 to 10 . L portion of the network on a particular variable. Voltage-controlled voltage source (VCVS) A voltage source whose parameters are controlled by a voltage elsewhere in the GLOSSARY system. Maximum power transfer theorem A theorem used to deter- mine the load impedance necessary to ensure maximum power to the load.