Improved Generation Operation of an Induction Machine Based Automotive Integrated Starter Alternator
Chathura Prasanna Mudannayake
A Thesis Submitted to The University of New South Wales for the Degree of Doctor of Philosophy
School of Electrical Engineering and Telecommunications
July 2009 CERTIFICATE OF ORIGINALITY
I hereby declare that this submission is my own work and to the best of my knowledge it contains no material previously published or written by another person, nor material which to a substantial extent has been accepted for the award of any other degree or diploma at UNSW or any other educational institution, except where due acknowledgement is made in the thesis. Any contribution made to the research by others, with whom I have worked at UNSW or elsewhere, is explicitly acknowledged in the thesis. I also declare that the intellectual content of this work is the product of my own work, except to the extent that assistance from others in the project’s design and conception or in style, presentation and linguistic expression is acknowledged.
(Signed)…………………………..
Chathura Prasanna Mudannayake
ii ABSTRACT
This thesis proposes improved techniques for controlling of induction machine based integrated starter alternator in order to achieve the challenging requirements of future automotive on-board power system. The proposed techniques are focused on improving DC voltage regulation, maximum power extraction and efficiency improvements.
A new sophisticated DC voltage controller that provides tight voltage regulation is proposed. The proposed controller is based on a linearized model for the combined inverter and induction machine. The proposed DC controller is included with speed and flux decoupling and an anti- windup technique. Extensive simulation and experimental results demonstrate the excellent DC voltage control performance of the proposed DC controller over a wide speed range and under various operating conditions.
This thesis proposes an improved field weakening implementation which is based on stator voltage control for maximum power extraction in generation mode. The controllers in proposed implementation included use a non-linear dynamic compensator (NDC), stator frequency decoupling and an anti-windup technique. This thesis investigates and eliminates the oscillation in high speed field weakening region caused when large loads are applied. The proposed implementation also provides better regulation for the stator voltage and stable operation over a wide speed range in the generation mode of the ISA. The proposed implementation allows extracting significant amount of additional power compared to conventional field weakening technique. The simulation and experimental results clearly demonstrate the performance of the proposed implementation for ISA.
A new loss minimised control method for the integrated starter alternator based on a loss model of the system was developed. The loss model operates in dq axes and takes into account the inverter and the machine losses. The experimental results demonstrates that proposed loss minimised control provides significant efficiency improvements under light load condition of the ISA.
This thesis also presents complete modeling of ISA, an extensive study on induction machine parameter variations and overall control design of the ISA system. The extensive simulation and experimental studies presented in this thesis clearly demonstrates the development of a new ISA which is low-cost, optimised for high efficiency and maximised power over a wide speed range and excellent DC bus voltage regulation under all conditions of generation.
iii
To my mother and father
iv ACKNOWLEDGEMENTS
This thesis was possible only due to the immense support I received from many people. First and foremost, I would like to express my gratitude to my supervisor, Professor Faz Rahman for giving me this opportunity; stimulating suggestions and countless encouragements during this research project. I would not have been able to complete this work without his immense support.
I am grateful to Dr. Baburaj Karanayil for kind support, technical suggestions and many interesting discussions during this thesis project. Thanks to my colleagues in Power Electronics and Drives Systems research group during the time of this candidature for their sincere friendship and interesting discussions.
I am thankful to my colleagues at Toshiba International Corporation for their continuous encouragements and support during last two years of my part-time candidature.
From my heart, great thanks to my wife, May, for her immense support, unswerving love and encouragements. This thesis would not have been possible without her help. I thank my sisters Manoja and Deshani; brother Ruchitha, brother-in-law Keshara and sister-in-law Suphunnika for their never ending love, help and countless encouragements.
My deepest gratitude to my mother and father, for giving me the best of the best throughout my life. I certainly owe all my success to them for their never ending love, guidance and immense encouragements.
Chathura Prasanna Mudannayake
Kingsford, July 2009
v CONTENTS
ABSTRACT...... iii
ACKNOWLEDGEMENTS ...... v
CONTENTS...... vi
LIST OF FIGURES ...... xii
LIST OF TABLES ...... xxi
LIST OF SYMBOLS AND ABBREVIATIONS ...... xxii
CHAPTER 1 ...... 1
INTRODUCTION ...... 1
1.1 Trends in Automotive On-board Electric Power Demand...... 1
1.2 Conventional On-board Electric Power Generation and Associated Problems...... 4
1.2.1 System Description 4 1.2.2 Generation Efficiency and Capacity Limitations 5 1.2.3 Voltage Regulation and Power Semiconductors 6 1.3 Trends in On-board Voltage Level ...... 8
1.4 New Developments in On-board Electrical Power Generators...... 11
1.4.1 Improvements to Lundell Alternator 11 1.4.2 Power Generation based on other Machine Topologies 12 1.5 Integrated Starter Alternator (ISA) ...... 12
1.5.1 Introduction 12 1.5.2 Specification and Requirements for ISA 16 1.6 Problem Statement ...... 21
1.7 Scope of this Thesis ...... 22
CHAPTER 2 ...... 24
vi REVIEW OF IMPROVED AUTOMOTIVE POWER GENERATION AND
INTEGRATED STARTER ALTERNATOR TOPOLOGIES ...... 24
2.1 Overview...... 24
2.2 Improved Automotive Power Generation ...... 25
2.2.1 Application of More Power Electronics to Lundell Alternator 25 2.2.2 Application of Other Electric Machine Topologies 27 2.3 Integrated Starter Alternator Topologies ...... 30
2.4 Comparison of Various Electric Machine Topologies Proposed for Integrated
Starter Alternator...... 36
2.4.1 Permanent Magnet Machine 36 2.4.2 Induction Machine 38 2.4.3 Switched Reluctance Machine 38 2.5 Summary ...... 39
CHAPTER 3 ...... 40
INTEGRATED STARTER ALTERNATOR SYSTEM MODELLING ...... 40
3.1 Overview...... 40
3.2 Induction Machine Modelling...... 41
3.2.1 Dynamic Model in Stationary Reference Frame 41 3.2.2 Dynamic Model in Synchronous Reference Frame 44 3.2.3 Steady-state Equivalent Circuit 47 3.3 Three-Phase Inverter Modelling ...... 49
3.3.1 Dynamic model for three-phase inverter 50 3.3.2 Loss Model for Three-phase Inverter 52 3.4 Battery Modelling...... 59
3.5 Internal Combustion Engine Modelling...... 61
3.5.1 Cylinder Pressure Torque Modelling 63 3.5.2 Engine Friction Torque Modelling 67 3.6 Summary ...... 67
vii CHAPTER 4 ...... 69
PARAMETER VARIATIONS OF INDUCTION MACHINE ISA...... 69
4.1 Overview...... 69
4.2 Variable Parameter Equivalent Circuit ...... 71
4.3 Determination of Parameter Variations ...... 73
4.3.1 Variable-frequency Stand-still Test 73 4.3.2 Variable-frequency Variable-voltage No-load Test 77 4.3.3 Impressed Stator Current Test 80 4.4 Variable Parameter MATLAB/Smiulink Model for IM ISA...... 85
4.5 Summary ...... 87
CHAPTER 5 ...... 88
CONTROL OF PROPOSED INTEGRATED STARTER ALTERNATOR...... 88
5.1 Overview...... 88
5.2 Rotor Flux Oriented Control (RFOC)...... 90
5.2.1 Rotor Flux Estimation during Starting 91 5.2.2 Rotor Flux Estimation at Higher Speeds 92 5.3 Computation of Stator Voltages...... 94
5.4 Inverter Nonlinearity Compensation...... 97
5.5 Designing of Current Controllers...... 102
5.5.1 Current, Flux and Torque Dynamics in RFORF 102 5.5.2 Current Controllers 105 5.6 Modelling and Experimental Results for Current Control Design ...... 108
5.7 Cranking Simulation...... 111
5.8 Summary ...... 115
CHAPTER 6...... 117
DC BUS VOLTAGE CONTROLLER ...... 117
6.1 Overview...... 117
viii 6.2 DC Voltage Dynamics of ISA ...... 118
6.3 Proposed DC Voltage Controller ...... 120
6.3.1 Controller Design 120 6.3.2 Simulation Results 123 6.3.3 Anti-windup Control Loop 126 6.4 Sizing DC bus Capacitor for ISA inverter ...... 131
6.5 Experimental Results...... 135
6.6 Summary ...... 137
CHAPTER 7 ...... 138
FIELD WEAKENING OPERATION OF INTEGRATED STARTER
ALTERNATOR ...... 138
7.1 Overview...... 138
7.2 Stator Voltage Control Field Weakening in Generation Mode...... 139
7.2.1 Principle of Operation 139 7.2.2 Effect of Stator Resistance and Inverter Losses on Field Weakening Optimization 145 7.3 Implementation of Field Weakening Optimization...... 148
7.3.1 Overview of the Proposed Implementation 148 7.3.2 Field Weakening Controller Designing 149 7.3.3 Simulation Results 153 7.3.4 Anti-windup Loops 159 7.3.5 Oscillation Caused by Large Load Disturbances and its Mitigation 163 7.3.6 Experimental Results for Stator Voltage Regulation and Power Capability in Generation Mode 167 7.4 Summary ...... 170
CHAPTER 8 ...... 171
LOSS MINIMISED CONTROL OF INTEGRATED STARTER ALTERNATOR... 171
8.1 Overview...... 171
ix 8.2 Loss Model of the System...... 173
8.3 Loss Minimization ...... 175
8.4 Experimental Results...... 178
8.5 Summary ...... 180
CHAPTER 9 ...... 182
CONCLUSIONS...... 182
9.1 Conclusion of this Thesis...... 182
9.2 Suggestions for Future Work ...... 185
REFERENCES...... 188
APPENDIX A ...... 195
A.1 Space-Phasor Representation of Three-phase Quantities...... 195
A.2 Stator and Rotor Flux Linkage of an Induction Machine...... 196
A.3 Induction Machine Model in Arbitrary Reference Frame ...... 202
A.4 State-space Model ...... 208
A.5 Power Flow and Electromagnetic Torque Production of Induction Machine...... 211
A.6 Matrix Representation of Reference Frame Transformation ...... 215
A.7 Induction Machine Loss Modelling ...... 215
APPENDIX B ...... 221
APPENDIX C ...... 236
EXPERIMENTAL SETUP...... 236
C.1 Overview of Experimental Setup...... 236
C.2 Equipment Details...... 238
C.3 Photographs of the Experimental Setup...... 239
APPENDIX D ...... 243
Refereed journal and magazine publications 243
x Refereed conference publications 243
xi LIST OF FIGURES
Figure 1.1 Power demand trend in automobiles equipped with above discussed new features [3] ...... 3
Figure 1.2 Schematic Diagram of Conventional Lundell Alternator ...... 4
Figure 1.3 Efficiency Vs. Speed characteristics of typical Lundell Alternator [5] ....5
Figure 1.4 Load dump transient of conventional system with Lundell alternator without avalanche diode protection [9]...... 6
Figure 1.5 DC voltage transient of conventional system fed by Lundell alternator during rapid engine speed increase [9]...... 7
Figure 1.6 Conventional 14V point-to-point automotive power distribution system.8
Figure 1.7 Voltage specification of proposed 42V PowerNet standards...... 9
Figure 1.8 Dynamic over voltage in proposed 42V PowerNet...... 10
Figure 1.9 A schematic diagram of three-phase AC electric machine based ISA....13
Figure 1.10 A schematic diagram of crankshaft mounted ISA ...... 15
Figure 1.11 A schematic diagram of a belt driven ISA...... 16
Figure 1.12 Motoring torque requirement for the ISA [22] ...... 18
Figure 1.13 Typical likelihood of starting: battery SOC Vs temperature [24]...... 19
Figure 1.14 An example motoring torque and power requirement of an ISA...... 19
Figure 1.15 An example DC power generating requirement of an ISA (directly coupled) ...... 20
Figure 2.1 Lundell power generation system with proposed fast field control ...... 25
Figure 2.2 Lundell power generation system with semi-boost converter...... 26
Figure 2.3 Power generation system with split winding PM machine ...... 27
xii Figure 2.4 Power generation system based on PM machine with switch mode rectifier ...... 28
Figure 2.5 Induction machine based power generation system with diode-bridge and
PWM VSI ...... 29
Figure 2.6 Engine stopping characteristics (a) conventional stop by merely cutting off fuel supply (b) controlled stop for reducing vibrations [8] ...... 31
Figure 2.7 Battery voltage and current during regenerative braking [8]...... 31
Figure 2.8 Power assist performance of IMA system in Honda Insight coupe [9] ..32
Figure 2.9 Cranking performance of IMA system in Honda Insight coupe [9] ...... 32
Figure 2.10 A comparison of torque capability and efficiency (a) Insight IMA electric machine (b) Civic IMA electric machine [10] ...... 33
Figure 3.1 qd axis equivalent circuit representation of dynamic model for induction machine in stationary reference frame...... 44
Figure 3.2 qd axis equivalent circuit representation of dynamic model for induction machine in synchronous reference frame...... 47
Figure 3.3 Steady state per phase equivalent of induction machine...... 49
Figure 3.4 Schematic diagram of three phase inverter ...... 50
Figure 3.5 Voltage and current during (a) switching ON (b) switching OFF ...... 52
Figure 3.6 Total switching energy loss per one cycle Vs (a) current at constant voltage (b) voltage at constant current...... 53
Figure 3.7 A leg of a three-phase PWM inverter with sinusoidal output current.....54
Figure 3.8 Conduction of active switch T1 and diode D2 in positive half cycle of current ...... 55
xiii Figure 3.9 Sketch of typical switching waveforms and associated energy losses in positive half cycle of current...... 55
Figure 3.10 Conduction pattern of active switch T1 and diode D2 during positive current half cycle...... 57
Figure 3.11 RC Battery Model ...... 60
Figure 3.12 An overview of a cylinder of a four-stroke engine ...... 62
Figure 3.13 Pressure Vs cranking angle for a single cylinder...... 65
Figure 3.14 Cylinder pressure torque vs. cranking angle for a single cylinder without ignition ...... 66
Figure 4.1 The steady-state variable parameter equivalent circuit model for induction machine ...... 72 Figure 4.2 An overview of variable-frequency stand-still test...... 74 Figure 4.3 Variation of rotor resistance and referred value of rotor resistance with slip frequency...... 76 Figure 4.4 Rotor leakage inductance variation with slip frequency...... 76 Figure 4.5 An overview of variable-frequency variable voltage no-load test...... 77 Figure 4.6 Variation of magnetising inductance with magnetising current at various frequencies...... 79 Figure 4.7 Stator core loss variation with magnetising current at various stator frequencies...... 79 Figure 4.8 An overview of impressed stator current test...... 80 Figure 4.9 Stator current (a) and stator voltage (b) during impressed stator current test...... 81 Figure 4.10 Stator current and voltage during impressed stator current test...... 85 Figure 4.11 Overview of the dynamic model to including parameter variation obtained in section 4.3 ...... 86
Figure 5.1 Overview block diagram of proposed ISA...... 89 Figure 5.2 Schematic diagram for flux estimation at starting (i.e. low speeds) ...... 91
xiv Figure 5.3 Schematic diagram for flux estimation in generation mode ...... 93 Figure 5.4 (a) Three-phase inverter, (b) Space vectors of three-phase inverter...... 94 Figure 5.5 Phase voltage pulse pattern (or switching pulse patterns) for six sectors...... 95 Figure 5.6 Approximated forward characteristics of (a) IGBT (b) diode ...... 98 Figure 5.7 One leg of a three phase inverter with (a) positive current (b) negative current ...... 99 Figure 5.8 Voltage pulses correspond to voltage vector in sector-1 for a practical inverter ...... 100 Figure 5.9 Overview block diagram for inverter nonlinearity compensation ...... 102 Figure 5.10 q and d axis current dynamics in RFORF ...... 104 Figure 5.11 Current, rotor flux and torque dynamics in RFORF ...... 105 Figure 5.12 Cross-coupling and back EMF voltages variation with the speed in generation mode...... 106 Figure 5.13 Closed qd axes current loops with decoupling...... 107 Figure 5.14 Phase currents, q and d axis current references and feedbacks in generation mode 120A load application at 2000rev/min ...... 109 Figure 5.15 Phase currents, q and d axis current references and feedbacks in generation mode 120A load dump at 2000rev/min...... 109 Figure 5.16 d axes current references and feedbacks in generation mode for 100A load dump at 3500rev/min (i) without decoupling (ii) with decoupling...... 110 Figure 5.17 Cold cranking (-20°C) specification and ISA starting torque...... 112 Figure 5.18 Torque, battery current and q axis current during cranking...... 112 Figure 5.19 Cold cranking performance (i) speed, (ii) torque, (iii)battery current, (iv) stator current, (v) stator voltage and (vi) q axis stator current...... 114 Figure 5.20 Speed and battery discharging current during cranking (i) 20°C at 36V (ii) -20°C at 36V and (ii) -20°C at 18V...... 115
Figure 6.1 Combined Dynamic Model for Induction machine and Inverter...... 120 2 Figure 6.2 vDC controller with decoupling the stator frequency and rotor flux...121 Figure 6.3 Closed DC bus voltage control loop...... 121
xv Figure 6.4 Bode plot for closed current and DC voltage loops (closed current loop natural frequency ( nc ) 100Hz, closed voltage loop natural frequency ( nvd ) 15Hz) ………………………………………………………………………………………...122 Figure 6.5 Load current, DC bus voltage, torque, q axis stator current, peak phase voltage, rotor flux and d axis stator current during application and dump of 133A load ...... 123 Figure 6.6 Speed, DC bus voltage, torque, q axis stator current, peak phase voltage, rotor flux and d axis stator current during engine acceleration and deceleration...... 124 Figure 6.7 Speed, DC bus voltage and q axis stator current during engine acceleration with and without rotor flux decoupling...... 125 Figure 6.8 DC bus voltage and q axis stator current during engine acceleration with and without rotor flux decoupling, zoomed in view of Figure 5.7 ...... 126 2 Figure 6.9 vDC controller including anti-windup control loop (The dotted line box is the “DC Voltage Control Block (VCB)” of Figure 5.1)...... 127 Figure 6.10 Changeover to generation mode at 600 rev/min (i) DC voltage (ii) q current ...... 128 Figure 6.11 DC load current, DC voltage and q axis current reference (i) with anti- windup loop (AWL) and (ii) without anti-windup loop...... 129 Figure 6.12 DC load current, DC bus current and battery current (i) with anti-windup loop (AWL) and (ii) without anti-windup loop...... 129 e*1 e* Figure 6.13 iqs and iqs current (i) with anti-windup loop (AWL) and (ii) without anti-windup loop...... 130 Figure 6.14 A schematic diagram of three phase inverter...... 131 Figure 6.15 DC bus voltage transients for 133A load dump at 2000rev/min for DC bus capacitance of (i) 20mF (ii) 50mF (iii) 80mF...... 133 Figure 6.16 DC bus voltage transients for 133A load application at 2000rev/min for DC bus capacitance of (i) 20mF (ii) 50mF (iii) 80mF...... 134 Figure 6.17 DC bus voltage ripples for (i) 50mF (ii) 80mF (iii) 20mF...... 134 Figure 6.18 Experimental results for 120A load application / dump at 2000rev/min (i) DC load current (ii) DC voltage (iii) d current (iv) q current ...... 135
xvi Figure 6.19 Experimental results: DC bus voltage transients for full load dumps at different speeds...... 136 Figure 6.20 Experimental results for transient during acceleration and deceleration of engine speed at no-load (i) speed (ii) DC voltage (iii) d current (iv) q current ...... 136
Figure 7.1 Voltage limit circle, current limit ellipses and DC power hyperbolas in ee vvqs ds plane ...... 142 Figure 7.2 Three sample MPOPs at three different stator frequencies in generating operation (i.e. at three different frequencies)...... 143 ee Figure 7.3 Operation trajectory over complete speed range in vvqs ds plane...... 144
ee Figure 7.4 DC power contours on vvqs ds plane with current ellipses and voltage limit circles ...... 146 Figure 7.5 Calculated maximum DC power output with and without assumptions...... 147 Figure 7.6 Overview of proposed implementation of stator voltage controlled field weakening method ...... 149 Figure 7.7 Rotor flux transfer function and controller in a closed loop ...... 149 e Figure 7.8 vqs controller and approximated transfer function in a closed loop. ..151
e Figure 7.9 vds closed control loop ...... 153
Figure 7.10 Speed, peak stator voltage and qd axis voltages during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus...... 154 Figure 7.11 DC voltage, DC bus current and battery current during engine speed accelerates from 1000rev/min to 5000rev/min with 100A load on DC bus ...... 155 Figure 7.12 Torque, q axis current, rotor flux and d axis current during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus...... 155 Figure 7.13 Trajectory of q axis voltage and d axis voltage during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus...... 156 Figure 7.14 Speed, peak stator voltage and qd axis voltages during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus (i) with and (ii) without stator frequency decoupling...... 156
xvii Figure 7.15 Load current, DC bus current, battery current voltage, peak stator voltage during sudden reduction of load from 100A to 40A at 5000rev/min...... 158 Figure 7.16 DC bus voltage, torque, q axis stator current, rotor flux and d axis stator current during sudden reduction of load from 100A to 40A at 5000rev/min. ....158 Figure 7.17 Trajectory of q axis voltage and d axis voltage during sudden reduction of load from 100A to 40A at 5000rev/min...... 159 e Figure 7.18 With and without anti-windup loop for vqs control when speed ramp form 800rev/min to 5000rev/min in 1 sec (i) speed (ii) input of flux saturation function e1* e* ( dr ) in Figure 7.6 (iii) output of flux saturation function ( dr ) in Figure 7.6...... 160 Figure 7.19 DC voltage, q axis stator current, d axis stator current and flux when speed is ramped up form 800rev/min to 5000rev/min in 1 sec (a) with anti-windup for e e loop vqs control (b) without anti-windup loop for vqs control ...... 161
e Figure 7.20 With and without anti-windup loop for vds control when load of 100A is applied at 5000rev/min (i) load current (ii) input of the q axis current saturation function (ip ) indicated in Figure 7.6 (iii) output of the q axis current saturation
max function ( Iqs ) indicated in Figure 7.6 ...... 162 Figure 7.21 DC voltage, q axis stator current, d axis stator current and DC bus
e current when load of 100A is applied (a) with anti-windup loop for vds control (b)
e without anti-windup loop for vds control ...... 163
e Figure 7.22 vds controller and the system that needs to be controlled by
e vds controller (i.e. plant) in closed loop ...... 164 Figure 7.23 An example approximated characteristics of the system that needs to be e controlled by the vds controller. This example corresponds to stator frequency of e =1570 rad/s...... 165
e Figure 7.24 Transients during full load disturbance at 5000rev/min with a PI for vds - controller (i) DC bus voltage (ii) d axis stator voltage (iii) q axis stator current...166
xviii Figure 7.25 Transients during full load disturbance at 5000rev/min with NDC for e vds -controller (i) DC bus voltage (ii) d axis stator voltage (iii) q axis stator current...... 167 Figure 7.26 Experimental results: maximum power output for stator voltage control and 1/ r method...... 169 Figure 8.1 Power flow of the ISA system in generation mode ...... 173 Figure 8.2 Total estimated inverter losses with and without the simplification of the e model (at iAds 150 )...... 175
e* Figure 8.3 Loss minimised trajectories of im and iqs for various speeds...... 177 Figure 8.4 Overview of flux reference generation of the ISA based on loss minimisation and field weakening (This is the “Flux Reference Block (FRB)” of Figure 5.1) ...... 178 Figure 8.5 Efficiency Vs DC power output at 2000 rev/min for conventional and loss minimized (LM) control (i) induction machine (ii) three phase inverter...... 179 Figure 8.6 Induction machine efficiency Vs speed for conventional and loss minimized (LM) control (i) for 250W DC load (ii) 500W DC load and (iii) 2000W DC load...... 179 Figure 8.7 Rotor flux, stator current and torque variation with speed at 500W output power for conventional and loss minimized (LM) control ...... 180 Figure A.1Stator of an elementary three phase symmetrical induction machine ...... 195 Figure A.2 Magnetic axes of a three phase induction machine...... 197 Figure A.3 Space-phasor representation using stationary, rotor and arbitrary reference frames...... 202 Figure A.4 Complex variable equivalent circuit representation for dynamic model of induction machine in arbitrary reference frame...... 208 Figure A.5 Synchronous reference frame IM model with core loss resistances .....216 Figure A.6 Simplified synchronous reference frame IM model with core loss resistances ...... 217 Figure A.7 Simplified IM model with core loss resistances in rotor flux oriented reference frame...... 218 Figure C.1 Experimental setup with proposed ISA...... 237
xix Figure C.2 Overview of experimental setup ...... 239
Figure C.3 ISA induction machine and drive motor ...... 240
Figure C.4 PC which hosts dSPACE1104 and 1102 control boards...... 240
Figure C.5 PC which hosts dSPACE1104 and 1102 control boards...... 241
Figure C.6 36V battery...... 241
Figure C.7 Prototype ISA inverter, load controller and drive inverter ...... 242
Figure C.8 Some control and power cable connections...... 242
xx LIST OF TABLES
Table 1.1 Estimated power requirement for potential future automobile loads [3] ...... 3
Table 1.2 Automotive Power Systems [11] ...... 10
Table 4.1 Contstant parameters obtained from conventional stand-still and no-load tests...... 73
Table 4.2 Impressed stator current test results...... 85
xxi LIST OF SYMBOLS AND ABBREVIATIONS
or e Electrical angle of synchronous reference frame i Cranking angle in degrees
Phase angle between voltage and current
s qs q axis stator flux in stationary reference frame
s qr q axis rotor flux in stationary reference frame
s ds d axis stator flux in stationary reference frame
s dr d axis rotor flux in stationary reference frame
e qs q axis stator flux in synchronous reference frame
e qr q axis rotor flux in synchronous reference frame
e ds d axis stator flux in synchronous reference frame
e dr d axis rotor flux in synchronous reference frame
s r Complex dynamic rotor flux vector in stationary reference frame
Polytrophic index of gas mixture e Electrical angular frequency of synchronous reference frame r Electrical angular frequency of rotor g Electrical angular frequency of arbitrary reference frame
nc Natural frequency (angular) of current control loop nvd Natural frequency of the closed-loop DC bus voltage transfer function nf Natural frequency (angular) of flux control loop
nfvd Natural frequency of the closed control loop.
e nlvd Natural frequency of the vqs closed loop
xxii A Piston area
Cb Capacitance that represents storage capacity of battery model
Cc Capacitance that represents surface effects of battery model
Cdc DC bus capacitor value
CR Compression ratio
D Duty cycle
Em Steady state peak backEMF
eon Energy loss during power semiconductor switching-on
eoff Energy loss during power semiconductor switching-off
eon/ off Total switching energy loss of power semiconductor per switching cycle. f2 Slip frequency in Hz
fe Stator electrical frequency in Hz
fs Sampling frequency
fsw Switching frequency f s s Generic representation of complex dynamic variable
Fs Generic representation of steady state variable
ias ,ibs ,ibs a , b and c phase stator voltages s iqs q axis stator current in stationary reference frame s ids d axis stator current in stationary reference frame s idr d axis rotor current in stationary reference frame s iqr q axis rotor current in stationary reference frame e iqs q axis stator current in synchronous reference frame e iqr q axis rotor current in synchronous reference frame
xxiii e ids d axis stator current in synchronous reference frame e idr d axis rotor current in synchronous reference frame s is Complex dynamic stator current vector in stationary reference frame s ir Complex dynamic rotor current vector in stationary reference frame
Is Steady state peak stator current
Ir Steady state peak rotor current
Im Steady state magnetising current
Imax Maximum current capability
iI Inverter current
iCAP Current through DC bus capacitor
iDC DC bus current
iBAT Battery current
iL Load current
iR DC capacitor leakage current
I peak Peak sinusoidal current j Imaginary
KPc Proportional gain of current controller
KIc Integral gain of current controller
KPVd Proportional gain of DC bus voltage controller
KIVd Integral gain of DC bus voltage controller
KPf Proportional gain of flux controller
KIf Integral gain of flux controller
e KIlvq Integral gin of vqs controller.
K e Ifvd Integral gain of the vds controller
xxiv Lm Magnetising inductance
Lsl Stator leakage inductance
Ls Stator inductance
' Ls Stator transient inductance
' Lrl Rotor leakage inductance referred to the stator
Lr Rotor leakage inductance
L Length of connecting rod m Modulation function ma , mb , mc a , b and c phase modulation function errors
M Peak modulation index
e mqs q axis modulation function in synchronous reference frame
e mds d axis modulation function in synchronous reference frame ncyl Number of cylinders.
P No of poles
PLR Lock rotor power
PNL No load power
PDC Power flow into the inverter
pswi_ Switching power loss (averaged within a switching period)
ptot_ sw Total switching losses of inverter
Ptot_ con Total conduction losses of inverter
Pinvtot Total inverter losses
Pmech Mechanical power of the induction machine
PAC Electrical power of the stator terminal of the induction machine
Pmtot Total induction machine losses
xxv Psysloss Total system losses
Pfe Core losses
Rdc Leakage resistance of capacitors
Rm Core-loss resistance
' Rr Rotor resistance referred to the stator
ref Rr Referred rotor resistance
Rs Stator resistance
Rt Terminal resistance of battery model
Rc Surface resistance of battery model
Re End resistance of battery model
rce Dynamic forward resistance of active switch
rf Dynamic forward resistance of diode
R Radius of crankshaft s Slip
TL Time constant of the low pass filter
tleft , tright , t0 , Parameters for SVM switching sector .
Tr Rotor time constant
td Dead time of inverter leg
toff Switching off time of the active switch
ton Switching on time of the active switch
Tsw Periodic time of switching cycle
Tem Electromagnetic torque vag ,vbg vcg a , b and c phase stator voltages with respect to the negative rail of the inverter
xxvi van , vbn , vcn a , b and c phase stator voltages with respect to the motor star point vas , vbs , vbs a , b and c phase instantaneous stator voltages
s vds d axis stator voltage in stationary reference frame
s vqs q axis stator voltage in stationary reference frame
e vqs q axis stator voltage in synchronous reference frame
e vds d axis stator voltage in synchronous reference frame
s vs Complex dynamic stator voltage in stationary reference frame
Vs Steady state peak phase voltage
Vmax Maximum voltage capability
Vpeak Peak sinusoidal voltage
vDC DC bus voltage
vripple Ripple component of DC bus voltage
v DC con DC component of DC bus voltage
vce Forward conduction drops of active switch
v f Forward conduction drops of diode
Vceo Saturation voltage drops of active switch
Vfo Saturation voltage drops of diode
V ()i Volume of i th cylinder at crank angle i
Zeq Equivalent impedance
Superscripts e Synchronous reference frame s Stationary reference frame g Arbitrary reference frame
xxvii * Reference or command value Subscripts
d Direct Axis q Quadrature Axis s Stator
r Rotor
Abbreviation AC Alternating Current AWL Anti Windup Loop BDC Bottom Dead Centre CLE Current Limit Ellipse DC Direct Current EMF Electro Motive Force fmep Frictional Mean Effective Pressure FRB Flux Reference Block FW Field Weakening FW-1 Field Weakening region-1 FW-2 Field Weakening region-2 ICE Internal Combustion Engine IGBT Insulated Gate Bipolar Transistor IM Induction Machine IMA Integrated Motor Assist IPM Interior Permanent Magnet IPU Integrated Power Unit ISA Integrated Starter Alternator ISG Integrated Starter Generator
xxviii ITAE Integral of time multiplied by absolute magnitude of the error criterion KCL Kirchoff’s current law KVL Kirchoff’s voltage law LM Loss Minimized LPF Low Pass Filter M/G Motor/Generator MOSFET Metal Oxide Semiconductor Field Effect Transistor MPOP Maximum Power Operating Point NDC Nonlinear Dynamic Compensator OEM Original Equipment Manufacturers PCU Power Control Unit PH Power Hyperbola PI Proportional-Integral PM Permanent Magnet PWM Pulse Width Modulation RC Resistive-Capacitive RFOC Rotor Flux Oriented Control RFORF Rotor Flux Oriented Reference Frame RMS Root Mean Square SCR Silicone Controlled Rectifier SMR Switch Mode Rectifier SOC State of Charge SPM Surface Permanent Magnet SPWM Sinusoidal pulse width modulation STB Starting Torque Profile Block SVM Space Vector Modulation TDC Top Dead Centre TEB Torque Estimation Block
xxix TF Transfer function VA Volt-Ampere VCB Voltage Control Block VSI Voltage Source Inverter
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CHAPTER 1
INTRODUCTION
1.1 Trends in Automotive On-board Electric Power Demand
The electrical system of the automobiles manufactured in early 20th century was very simple. They consisted of a battery, battery charger, a starter, an ignition device, a horn and few lamps. The battery voltage was 6V and the electrical power distribution was point-to-point trough the dashboard control switches. Complexity of on-board power system started to grow rapidly after the word war-II, adding new loads such as radio, multi speed windshield wipers, window lifts and etc [1]. In 1950s, the automotive battery voltage was increased to 12V in order to cater this increased on-board electric power demand. Since then, the on-board power demand increases at even faster rate due to addition of large number of electric powered on-board devices and substitution of mechanical actuators with electric actuators. The power demand of present mid size car is about 1.5kW, compared to 500W in 1950s automobile. Current trends in automotive on-board electrical system focused on three broad objectives. They are; (1) increased passenger comfort, convenience & safety; (2) increased fuel economy; (3) reduced emission. These objectives are briefly discussed in following section.
Comfort, Convenience & Safety: these have been the most dominant reason for rapid increase in on-board electric power demand in the past. This will continue to be the case in the future due to high competition in automotive markets and stringent standards for passenger safety. Some of the potential future loads in automobile for increased comfort, convenience and safety are: air conditioning while engine is stopped; electrically heated seats and electric steering wheel; computers, business machines, communication devices, navigators, games and high-end sound; active suspensions; electric power steering; electric braking; and electric de-icing (i.e. Windshields) [2].
Fuel Economy: the cost of fuel has increased significantly during last decade due to the high demand and new geopolitical environment in the world. It is a well known fact that the fossil fuel is not a renewable energy source and rapidly diminishing due to extensive exploitation. In order to reduce the diminishing rate and high dependency of foreign oil
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sources (i.e. energy security), various initiatives and regulations are being introduced by the authorities of countries to improve the fuel economy in transportation and other sectors. For example, in the U.S, Partnership for New Generations of Vehicle (PNGV) is a program co-funded by the US government and auto industry to develop an 80mpg (2.94 litres/100km) “green” automobile. Europe is also pursuing an equivalent program to develop a green car [2].
More electrification has been identified as one of the best way to improve the fuel economy of automobiles. Substitutions of various mechanical actuators and mechanically driven systems by electric motor drives have been considered by automotive engineers. The better controllability of electrically driven system leads to save energy while delivering same or better performance. Some of the reported potential new electric loads require for improved fuel economy are: electrically driven engine accessories; electromagnetic valves; electric power steering; electric air conditioning; and electric turbo boost.
“Soft Hybrid” or “Mild Hybrid” functions such as idle engine stop, electric launch assist and regenerative braking also allow improving the fuel economy greatly and will be discussed in later section of this chapter.
Reduced Emissions: increased public concerns over air/water pollution and global warming, emission targets imposed by Kyoto protocol and other emission reduction ambitions are pushing automotive manufactures to investigate emission reduction techniques. Obviously, improvement of fuel economy results in great reduction in emissions. In addition various approaches are being considered to improve the combustion process and to condition the exhaust so that emissions can be reduced. Some of these approaches require using electrical apparatus such as eclectically heated catalytic converter, plasma exhaust processor, electromagnetic valves and etc.
Estimated power requirement for some of the above discussed potential electrical loads are given in Table 1.1[3]. As can be seen in Table 1.1, above features require significant amount of electric power and will greatly increase the on-board power demand. If these features are adopted, the estimated on-board electric power demand for large car can be as high as 5.5kW and 3kW for a small car. This is shown in Figure 1.1.
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Table 1.1 Estimated power requirement for potential automobile loads [3, 4]
Load Peak Power Summer Winter average (W) average Power Power (W) (W) Electromechanical engine valve 3200 1000 1000 actuator Engine cooling fan 800 500 40 Engine coolant pump 500 400 80 Active suspension 3000 100 100 Electrically assist power steering 1000 100 100 Electrically heated catalytic 3000 60 120 converter AC mobile power 2400 600 600 Electric brake by wire 2000 40 40 Heated windscreen 700 - 100 Telimatic, Navigation, etc 100 100 100 Electric A/C compressor 3000 - 4000 -
6000
5000
4000
3000
2000
1000
Estimated Power Demand (W) Demand Estimated Power 0 1990 1995 2000 2005 2010 Year Small Cars Medium Cars Large Cars
Figure 1.1 Power demand trend in automobiles equipped with above discussed new features [3]
Next Section discusses the conventional automotive power generation and associated problems in applying it for future automobiles.
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1.2 Conventional On-board Electric Power Generation and Associated Problems
1.2.1 System Description
The conventional 12V DC system is supplied by a Lundell alternator. Lundell, or Claw- Pole, alternator is a wound field synchronous machine. The rotor of the machine comprises pair of claw pole pieces and field winding which is wound around the claw poles. The field winding is supplied by controlled DC current through a pair of slip rings. The power winding, or stator winding, is wound in a three-phase configuration and a three-phase uncontrolled rectifier (i.e. three-phase Diode Bridge) is used at the machine output to obtain DC voltage. The output DC voltage of the alternator is regulated by controlling the field current using pulse with modulation. The field winding typically has a long time constant, for example, 100ms or more. The stator of the Lundell alternator has a high synchronous reactance value. These characteristics determine the electrical performance, such as transient behaviour of the Lundell alternator [5]. Figure 1.2 shows the schematic diagram for conventional Lundell alternator.
vDC
vref
Figure 1.2 Schematic Diagram of Conventional Lundell Alternator
The conventional Lundell alternator associated with many problems that make this topology less attractive to the future automotive power generation. These problems are; low efficiency, capacity limitations and poor voltage regulation of the DC bus. These problems are described in next sections.
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1.2.2 Generation Efficiency and Capacity Limitations
Lundell alternator has poorest efficiency characteristics of any component in the on board power system. Figure 1.3 shows efficiency vs. speed of a typical Lundell alternator. As can be seen in this figure, the efficiency is about 61% at alternator speed of 1800 rev/min, which corresponds to idle engine speed (for pulley ratio of approximately 1 : 3 ). At alternator speed of 6000 rev/min, which corresponds to normal cursing speed range of engine, the efficiency is about 45% [5].
100 90 80 70 60 50 40
Efficiency (%) 30 20 10 0 1800 3000 4000 5000 6000 Alternator Speed (rev/min)
Figure 1.3 Efficiency Vs. Speed characteristics of typical Lundell Alternator [5]
This is a poor choice for future system that requires higher efficiency. The low efficiency means not only that considerable amount of energy is wasted, but that the alternator has to be designed to withstand the thermal stress of dissipating large amount of heat. Lundell alternator has high synchronous reactance. The high synchronous reactance means that the stator core must be designed to support unnecessary high flux, adding more cost and weight. Due to these reasons, inexpensive Lundell alternators do not scale well to rating higher than the present maximum [6]. The unique structure of this alternator topology inherently limits output power capacity for given rotor diameter. The Lundell alternator-based system has already reached its maximum output power capability in some vehicles, even after incorporating several improvements to the machine [7].
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1.2.3 Voltage Regulation and Power Semiconductors
Conventional system supplied by Lundell alternator has poor voltage regulation characteristics. Voltage of conventional 12V on-board power system varies between 9V and 16V depending on output current, state of charge, age of the battery, and other factors. When system voltage is higher, loads draw higher current than necessary and thus the power electronics need to be rated for continuos operation at highest current.
Moreover, load-dump transient, a voltage spike that appears when fully loaded alternator suddenly loses its load, may occur in the automotive power system. For example, when a charging battery is inadvently disconnected, the voltage behind armature reactance can suddenly show up on the system. This can be as high as 40V, 100ms transient if the alternator is protected with avalanche diodes, and much higher spike if not [8].
Figure 1.4 Load dump transient of conventional system with Lundell alternator without avalanche diode protection [9]
Figure 1.4 shows an example load dump transient of conventional 12V system without avalanche diode protection. As can be seen in this figure, a large voltage spike occurs momentarily due to the energy in the stator winding leakage inductance at the time of
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load transient. However, the energy associated with this momentary spike is relatively small and can be dissipated using an avalanche diode. The long ramp down transient voltage from 40V to 14.5V over 100ms is more troublesome, the long time constant of field winding of Lundell alternator causes continued conversion of shaft mechanical energy into the electrical energy, even though it is no longer needed [9]. The poor voltage regulation due to long field winding time constant of Lundell alternator can also be witnessed in Figure 1.5, which illustrates the voltage transient occurring during rapid engine speed increase. Initial voltage overshoot followed by low frequency ripples due to the regulator hunting can be seen in this figure.
Figure 1.5 DC voltage transient of conventional system fed by Lundell alternator during rapid engine speed increase [9].
The poor voltage regulation is very problematic for future automotive power system as utilization of power electronics is expected to grow rapidly due to application of more advanced loads. More power electronic switches would be used to replace relays (i.e. driving ON/OFF type loads) and controlling loads such as inverter for motor control, DC/DC converters and etc. The power electronics switches would have to be rated for transient voltage at least four times the nominal system voltage to withstand the transient caused by the poor voltage regulation of Lundell alternator. This is an expensive requirement for semiconductors because load dump may never occur during
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the lifetime of a car, yet components have to be ready to handle it. The result is that power semiconductors have to be grossly overrated both for highest current and voltage transient [8]. Given the fact that the cost is an overarching concern in automotive industry, this issue somewhat discourages the idea of using more power electronic devices in automotive electrical system.
1.3 Trends in On-board Voltage Level
The present 14V automobile power system uses point-to-point wiring to power the majority of on board loads. Consequently, the wiring harness of automobiles is heavy and complex [10]. For example, wiring harness of a mid-range car has over 2km of wires, weighing over 35kg. This complex wiring harness causes many difficulties during assembly process and leading to higher cost. Also retrofitting, fault tracing and repairing are more time consuming. The bulky wiring harness also constraints the vehicle body design. Figure 1.6 shows the conventional 14V automotive power distribution system.
Figure 1.6 Conventional 14V point-to-point automotive power distribution system
Ever increasing new loads and consequent higher on-board power consumption increases the complexity and associated difficulties in wiring harness of modern
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automobiles. In view of addressing these difficulties, several changes to the conventional automotive power distribution system have been proposed. Adaptation of a higher distribution voltage is one of the most radical changes. A higher voltage system will reduce the weight and volume of the wiring harness since higher voltage can deliver same amount of power at lower current. Furthermore, some automotive loads such as motors, will be able to operate more efficiently at higher voltage compared to conventional 14V. The MIT Consortium on Advanced Automotive Electrical/Electronic System and Component which constitutes some automakers and automotive OEMs, has proposed voltage level of 42V automotive power system. This automotive power system is called “42V PowerNet” and ISO standard for 42V PowerNet has been proposed (ISO/WD 21848-2). The proposed standard requires much tighter voltage regulation than the conventional 14V electrical system. The operating voltage criterion of the draft standard is shown in Figure 1.7.
Figure 1.7 Voltage specification of proposed 42V PowerNet standards
As can be seen in Figure 1.7, the maximum load dump transient voltage of 42V PowerNet is 58V. The proposed standard also defines the time duration for which the load dump voltage transient is allowed. The allowable time duration is 400ms with rise time and falling time of 10ms and 20ms respectively. Illustration of allowable limits to over voltage transient specified in 42V PowerNet is shown in Figure 1.8.
The changes in the automotive power system voltage and over voltage specification have a direct impact on the vehicle electronics in terms of cost, reliability, and packaging [11]. The tight over-voltage transient specifications encourage the move towards using more on-board power electronics because of comparatively higher utilisation of voltage rating of power electronic switches. Table 1.2 compares present automotive power systems and proposed 42V PowerNet voltages and the voltage rating requirement for power electronics [12].
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tms 10 tms 400 tms 20 R S F
Figure 1.8 Dynamic over voltage in proposed 42V PowerNet
Table 1.2 Automotive Power Systems [12]
Automotive Battery Nominal Maximum Maximum Power Power System voltage Operating Operating Dynamic Electronic voltage Voltage Over voltage Voltage Rating 14V Car/Light truck 12V 14V 24V - 60 - 40V 28V Heavy Truck 24V 28V 34V - 80 – 60V 42V PowerNet 36V 42V 50V 58V 100 – 80V
A concept of dual 42/14V (i.e. two voltage levels) on-board power system has also been proposed in order to facilitate smooth transition from convention 14V to 42V system [10][12, 13]. 42V Power system is one of the many voltage levels proposed or applied for automobile. Much higher voltage levels such as 144V/12V have also been proposed and applied [13]. At this moment of time it is difficult to predict if automobile manufacturers will adopt a universal standard voltage level or different voltage levels that suit to individual automotive manufacturer.
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1.4 New Developments in On-board Electrical Power Generators
1.4.1 Improvements to Lundell Alternator
Various attempts have been made to enhance the performance of the conventional Lundell alternator. The proposed improvements are focused on three broad areas; increasing power capability, increasing efficiency and reducing load dump transients. The efficiency improvements to the alternator not only save the energy wasted in the electric machine but also improve the fuel economy. The alternator manufacturer, Robert Bosch uses rule of thumb that says for every 5% improvement in alternator efficiency results in about 1% improvement in fuel economy [14]. The power capability improvements attempt to address the power demand forecast of the future automobiles. The most of the proposed load dump transient improvements are aimed at achieving 42V PowerNet specification.
It has been reported that the major automotive alternator manufactures such as Delco Remy, Visteon, Delphi, Denso, Bosch, Valeo, Mitsubishi and Ecoair have enhanced the claw pole machine design in their new high output alternators. One solution for low output problem (i.e. low efficiency and low power output) is to reduce the inherent leakage flux in the claw pole machine design. In some of the new high output alternators, lower leakage flux is achieved by mounting permanent magnets between claws so that more flux is coupled to the stator winding. An efficiency improvement of 20% (i.e. increase efficiency from 50% to 60%) has been reported for alternator with this improvement [14].
The stator of the conventional Lundell alternator is wound with round-wire since it is the well established technology with a lower cost. However, in new Lundell alternators, the rectangular conductors used for the stator winding in order to achieve high Slot Fill Ratio (SFR), hence reduce the resistance loss and increase the output. The thickness of the rectangular conductors must also be limited to proper thickness level in order to reduce the resistance caused by the skin effect. In addition many patents and manufacturing reports have been presented by major alternator manufacturers in regard to design and manufacturing of the stator winding with segmented conductors [15].
Some high power alternators use water jacket to cool the machine instead of the conventional forced air cool. The heated water caused by waste heat in this arrangement
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can be used for various heating requirement such as defroster used to clear the windshield and heaters that heat the passenger compartment more rapidly. Visteon claims that their alternator can deliver up to 1250 Watts of waste heat to its cooling system [14]. With this type of alternator system, the overall efficiency of the alternator can be somewhat increased during the winter periods.
In addition to the improvements in Lundell machine design, improvements or alternative to the power electronics of the Lundell alternator have also been proposed by automotive researchers. Some of these will be discussed in Chapter 2.
1.4.2 Power Generation based on other Machine Topologies
Adopting and improving conventional Lundell alternator to suit future automotive power systems seems very logical due to the large investment made in manufacturing infrastructure of this type of alternator and relatively low cost of this technology. However, the power capability and efficiency figures still remain inadequate in spite of the various attempts to improve this technology due to the unique structure of this machine topology. As a solution to this problem, power generation systems that utilize various other machine topologies have been proposed. Some of these proposals will be discussed in Chapter 2. The motivation for replacing conventional claw pole Lundell generator with another electric machine topology such as induction machine or permanent magnet machine opens up a new opportunity. These machines can be designed to have a torque profile that suits both engine cranking and power generation so that there is a prospect of elimination of starter motor. This combined approach is called “Integrated Starter Alternator (ISA)” or Integrated Starter Generator (ISG) and will be discussed in next Section.
1.5 Integrated Starter Alternator (ISA)
1.5.1 Introduction
In traditional automobiles, the power generation and starting of the engine perform by two different machines. As discussed in the previous sections, Lundell alternator is used for the power generation. The starting or cranking of the engine is performed conventionally by a seperate DC series motor so called starter motor. A typical starter motor consists of a solenoid relay and a pinion gear mechanism. The solenoid is
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energised by the engine starting switch on the dashboard. When the solenoid is energised, the starter motor is connected to the battery via relay contacts and the pinion gear mechanism moves towards the flywheel ring and engages with the teeth on flywheel ring to initiate cranking of the engine. The starter motor draws large current and develops characteristically high starting torque.
Integrated Starter Alternator is an automotive electric subsystem which fulfils the functions of engine starting and power generation by one on-board electric machine. In fact, the concept of using one machine for cranking and on-board power generation is not a new idea and the early attempts are dated back to 1930s [15]. Automotive engineers were well aware of the fact that the DC machine can be used as a motor and generator. However, the concept of combined starter generator was not evolved to a practical reality, because of the great difference in the torque specification of starter motor and generator; and the complexity of the system control in the age without sophisticated power electronics.
Figure 1.9 shows a schematic diagram of an integrated starter alternator system based on a three-phase AC machine. The system consists of electric machine, three phase inverter and controller that controls the electric machine in both motoring and generation modes. The DC bus of the inverter is connected to the battery and various on-board loads.
Figure 1.9 A schematic diagram of three-phase AC electric machine based ISA
ISA allows deleting one electric machine from the conventional onboard power system. In addition, the controllability and better performance of the ISA would allow successful implementation of idle stop/restart function. This function could allow the
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engine to be automatically stopped when it would otherwise be idling, and then “instantly” restart, thus saving the fuel consumption during idling; for example, stop at a traffic light in an urban drive. The successful implementation of this function would require not only instant restart but also controlled stop of the engine for reduced stop- vibrations. Given the frequent occurrence of engine stops of an automobile equipped with stop/restart feature, reduced engine stop-vibrations are important for passenger comfort as well as life-time of various automotive equipments.
The ISA-equipped car is also of interest because it represents a halfway point between conventional vehicle and true hybrid electric vehicle which has higher capacity power electric drive and greater battery energy storage capacity. As a result, vehicles with ISA systems rated in vicinity of 10kW such as Honda Civic Hybrid are often referred to as “soft” or “mild” hybrids [16]. ISA can be used to achieve optional soft hybrid functions such as launch assist (or torque boost) and regenerative braking. With launch assist or torque boost function; an extra torque is produced by the electric machine which is added to the engine torque to improve the acceleration performance. With regenerative braking function, part of the braking energy is extracted to charge the battery by applying a negative torque from the ISA. Also, charging of the battery can be achieved when driving down-hill by applying a negative torque. This is called “opportunity charging”.
In addition to the above benefits, ISA can be used for active cancellation of engine torque ripple [17, 18]. For this function, the torque controller of ISA should have high bandwidth so that the ISA can generate torque ripple with same frequency and 180o phase shift to engine torque ripple caused by cylinder firing action. This is sometimes referred to as “active flywheel” since its action helps smoothing of the engine speed.
Two main mounting configurations for the ISA machine have been considered in the literature, namely, (i) crankshaft mounting and (ii) mounting the ISA offsetting the crankshaft.
Crankshaft mounting: the electrical machine is mounted directly on the crankshaft, usually between engine and the clutch. The flywheel can be eliminated as the rotor of the machine can provide the inertia required for smoothing the engine torque ripples. Crankshaft mounted ISA has high capability for mechanical power transmission between engine and the electrical machine as it is directly mounted on the crankshaft.
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For example, Honda Civic Hybrid utilizes the directly mounted electric machine to realize the power transmission capability required for soft hybrid operations. However, the space between engine and clutch is limited and the ISA machine should be designed to fit the available space in order to avoid major changes to the conventional engine structure. Figure 1.10 shows a schematic diagram of crankshaft mounted ISA.
Figure 1.10 A schematic diagram of crankshaft mounted ISA
Offset Coupled: the ISA is mounted offsetting the crankshaft. There are several options for mechanical power transmission between ISA machine and the crankshaft; namely, belt drive, chain drive and gear drive. The belt drive is the most attractive among them because it requires least modification to the existing engine structure. Also, a lager machine can be packed in the vicinity of the existing alternator. The ISA can be driven by either a separate belt or included within the existing belt system. Even though the belt drive can satisfactory transmit the torque stop/restart cranking, the conventional starter motor may still be required for initial cold engine cranking due to the belt tension limitations, wearing issues, power rating and etc.[15, 19]. The much higher power requirement associate with ISA system may also place heavier demands on the bearing compared to conventional alternator system. Figure 1.11 shows a schematic diagram of belt driven ISA.
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Figure 1.11 A schematic diagram of a belt driven ISA
1.5.2 Specification and Requirements for ISA
Many challenging requirements and specifications need to be considered in ISA system design. These include withstand capabilities, features and various performance requirements. Some of those specifications and requirements are discussed in this section. Withstand capabilities and General requirements
Coolant and Ambient Temperatures: Air and liquid are the main possible coolants for the ISA electric machine. The ambient temperature of the engine bay ranges from -40oC to 125oC which is a typical requirement for air cooled ISA machine. There are usually two cooling options for liquid cooling, namely, engine coolant and transmission oil. The temperature of engine coolant can be up to 120oC ~ 130oC while the available transmission oil temperature can be up to 135oC ~ 150oC [15]. Thermal design of the ISA machine is quite challenging due to the high temperature of the available coolants. A separate liquid cooling loop may be required if the temperature of cooling medium needs to be lower than the above temperatures.
Same coolant media (i.e. air or liquid) can also be used for cooling the power electronics box of the ISA. The power electronics module also experiences the harsh ambient conditions similar to the electric machine. The power devices may require
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higher junction temperature rating about 175oC and higher temperature rating of about 150oC for packaging and other electronic components [20]. In addition to the high temperature of the coolant, the power electronics converter arrangement is subjected to frequent temperature cycles. For example, assuming a lifetime of 15 years and two cold starts per day, the coolant is heated up twice from 5oC to 105oC and cooled down again. This equals to 10,000 temperature cycles with temperature shift of 100oC. This passive temperature cycling together with active temperature cycling due to the loading and unloading of the ISA, stresses the power semiconductors and their interface material like bond wires, substrates, solder, etc [21]. These issues also need to be taken into consideration when designing power electronic module.
Mechanical Stresses in High Speeds: The operational speed of the electric machine is typically in the range from 0 to 6000rev/min for crankshaft mounted ISA while the maximum speed of belt driven ISA can be as high as 13800 ~ 19200rev/min for pulley ratio of 2.3 ~3.2. The mechanical stresses at high speed operation should be taken into account when designing the electric machine.
Geometry: Designing the geometry of ISA system to suit existing engine structure is vital in order to maintain low overall manufacturing cost of the vehicle as any significant change to existing structure would come with large investments to manufacturing infrastructure.
Weight: Total vehicle weight reduction is one of the best ways to increase the overall fuel economy of the vehicle. Therefore, minimising the weight of the total ISA system should be considered during the design of the ISA hardware.
Noise Limits: The noise produced by the ISA system must be within the acceptable limits to ensure the passenger comfort.
Vibration and Shock Resistance: The ISA system should be design to withstand the vibrations and shocks presence in the engine compartment of the automobile. Vibration has a great impact on the life time of the electric machine as well as to power electronic module of the ISA. Resistance to these vibrations and shocks need to be taken into account when designing the ISA system.
Protection against Water and Dust Ingress: The electric machine and power module should have suitable protection rating to avoid harmful dust and water ingress.
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Performance Requirements
Torque Requirement in Motoring Mode: The ability to start the engine quietly and quickly is an important feature of ISA. This enables realising stop/restart function so that the fuel can be saved when the vehicle is not moving in heavy traffic conditions. The cold cranking is more onerous than staring the engine under hot conditions. The ISA needs to provide breakaway torque of about 1.5~1.8 times of normal cranking torque from 0 to 20~30rev/min to overcome static friction under cold conditions (i.e. - 29oC ~ -50oC) [15]. The quick starting and cold cranking requirements defines the peak motoring torque and maximum motoring torque curve for the ISA system in motoring mode. Figure 1.12 illustrates the shape of the desired motoring torque characteristics for the ISA.
Figure 1.12 Motoring torque requirement for the ISA [22]
The peak torque value is determined by the torque requirement for cold cranking operation of the engine. The maximum torque curve is determined by the desired starting time. The desired starting time is usually in the range of 250ms to 400ms [22]. With conventional starter motor, the initial cranking typically stops at about 80~180rev/min and it requires a burst of fuel to accelerate to idle speed. In automobiles equipped with ISA, it is preferred to crank the engine close to the idle speed and seamless transition to the fuelled operation in order to reduce the emissions and increase the fuel economy [23]. Even though time to reach the idle speed is small, achieving
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required cranking torque at higher speed is a challenge to machine designers due to the lower battery voltage available, especially at low temperatures with low state of charge (SOC) of the battery. Figure 1.13 shows battery state of charge (SOC) and temperature with typical likelihood of engine starting.
Figure 1.13 Typical likelihood of starting: battery SOC Vs temperature [24]
1.8 1.6 1.4 1.2 1 Power 0.8 0.6 Torque 0.4 Torque & Power(PU) Torque & 0.2 0 0 500 1000 1500 2000 2500 Speed (rev/min)
Figure 1.14 An example motoring torque and power requirement of an ISA
In addition to cranking, launch assist function may be needed depending on the configuration of ISA. The torque value for launch assist or torque boost is determined by the performance requirements of the vehicle. The launch assist torque is required typically in torque lacking speed range of the engine. For most of the gasoline engines, this range is usually from engine idle speed up to about 2000 ~ 2500 rev/min where the engine can develop higher torque output. However, the launch assist can be applied to increase the engine torque in wider speed range, for example, from idle speed up to red-
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line speed (i.e. 5000~6000 rev/min) if required. The time duration of launch assist could be from 3~4 seconds up to 15 seconds. However, this time period is varied depending on the vehicle configuration and design approach for the ISA. Some crankshaft mounted ISA may be required providing the torque assistance continuously, similar to hybrid electric vehicle[15]. An example motoring torque and power requirement is shown in Figure 1.14.
Requirements in Generating Mode:
DC power generation requirement for a crankshaft mounted ISA is shown in Figure 1.15 . This figure uses per unit values to describe the power requirement as the power demand is different for different automobiles. Generally, the output power at idle speed needs to be at least 35~ 60% of the maximum continuous output power of the generator. The typical short duration (i.e. about 1~3 seconds) maximum power requirement is about 1.3~1. The generating power requirement above 4000rev/min is significantly lower since it is beyond normal driving speed of the engine [15]. ISA system is expected to be operated with high efficiency than the conventional alternator throughout the entire generation speed range.
1.6 Short Duration 1.4 1.2 1 0.8 Continuous 0.6
DC Power (PU) 0.4 0.2 0 0 1000 2000 3000 4000 5000 6000 Speed (rev/min)
Figure 1.15 An example DC power generating requirement of an ISA (directly coupled)
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1.6 Problem Statement
The future automotive power generation is facing many challenges discussed in this chapter. Conventional automotive Lundell power generator topology hither to used fails to meet the stringent requirement of future on-board power system. These requirements still remain challenging for the generation mode operation of ISA despite various attempts reported in literature. These challenges are given follow. 1. Tight DC voltage regulation over wide speed range 2. Maximum power extraction from the generator 3. High efficiency operation 4. Lowest possible cost
The future generation system requires achieving above goals simultaneously which requires rigours analysis for strong understanding of all aspects of generation operation of the ISA. For this, the dynamics and steady state behaviour of the complete ISA system should be carefully investigated. The dynamic analysis will require utilizing coordinate reference frames in order to capture fast changes of the system including electric machine as well as the power converter. Various losses of the system and their variations with the operating point will need to be identified for the generation of power at high efficiency. The control design should be based on systematic analysis of the dynamics of the ISA system to enable fast response and stable operation over a wide speed range and fast acceleration and deceleration of the engine speed. The maximum power extraction of ISA in generation mode should be available throughout wide operational speed range and insensitive to the parameter changes of the system. Extensive testing and computer simulation studies will be required for the purpose of understanding and evaluating the behaviour of the system. Computer simulations using conventional constant lump parameter models will not be sufficient for accurate prediction of the system behaviour. Parameter variations of the electric machine will require incorporating into the computer models in order to closely mimic the actual system.
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1.7 Scope of this Thesis
The aim of this thesis is to achieve challenging requirements of future automotive power system by utilizing an induction machine based ISA system. The induction machine is selected in this thesis due to the lower cost and various other benefits which are discussed in Chapter 2. The prototype of the proposed ISA system was built and extensively tested for various functionality and performance discussed throughout this thesis. The prototype ISA system discussed in this thesis is designed for sample DC bus voltage of 42V. However, the analysis and control design proposed in this thesis is general enough so that it can be equally applied to ISA with any DC bus voltage level that may be used in future automobile.
This thesis proposes a new DC voltage control method for achieving tight voltage regulation over wide operational speed range of ISA. The proposed method also helps reducing the size of DC bus capacitor due to the fast operation of DC voltage controller. The control design, simulation and test results of the proposed method are presented in Chapter 6.
An improved field weakening control method which allows extraction of maximum power is presented in Chapter 7. The proposed method allows increased power capability and stable operation of ISA in generation mode over wide operational speed range. Simulation and experimental test results that demonstrates the performance of the proposed field weakening implementation are also presented in this chapter.
Chapter 8 proposes loss minimised operation of the ISA. The proposed model based loss minimisation algorithm is derived based on the loss model of the motor and the loss model of the inverter in rotor flux oriented reference frame which is discussed in Chapter 3. Experimental results for efficiency improvements are presented in this chapter.
As described above, Chapter 6, 7 and 8 of this thesis address the main challenges of future automotive power system, namely, tight DC voltage control, high power extraction and efficient generation. The work leading up to these topics are included in Chapter 2 to Chapter 5.
Chapter 2 presents a review of various improved automotive power generation and integrated starter alternator topologies.
22 Chapter 1: Introduction
Chapter 3 discusses the complete modelling of the ISA system which forms the basis for developments and analysis proposed in this thesis. This chapter includes detailed discussion on modelling of induction machine, three phase inverter, internal combustion engine and battery. The modelling approaches presented in this chapter are used throughout the thesis for control design as well as computer simulations.
Extensive study on parameter variations of induction machine and its determination is presented in Chapter 4. In this chapter, the induction machine of the ISA is modelled with variable parameters instead of conventional constant parameters. Parameter variations are determined through experiment studies. A computer model which incorporates parameters variations is obtained. This model is used for computer simulation studies presented in this thesis which mimic real induction machine more accurately.
Chapter 5 of this thesis presents the overall control design which incorporates the above discussed developments to the proposed ISA. This chapter also discusses field oriented control, inverter nonlinearity/dead band compensation and decoupled current control design. Brief discussions on cranking (i.e. starting) operation of the ISA are presented for the sake of completeness even though the main focus of this thesis is on generation operation of ISA. Experimental and simulation results for current control; and cranking simulation results are also presented in this chapter.
In Chapter 9, conclusions of this thesis and recommendation for further work are presented.
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CHAPTER 2
REVIEW OF IMPROVED AUTOMOTIVE POWER GENERATION AND INTEGRATED STARTER ALTERNATOR TOPOLOGIES
2.1 Overview
This chapter reviews various topologies reported in literature for improved automotive power generation; and integrated starter alternator systems. The improvements to the automotive power generation focus on three main areas; efficiency improvement, capacity improvements and load dump transients. Two broad approaches have been considered in the literature for realising above main areas of focus. They are; (1) application of power electronics in conventional Lundell alternator and, (2) application of new machine topology and power electronics.
The integrated starter alternator (ISA) systems presented in literature utilise various electric machine, converter and control topologies. The electric machine topologies such as induction machine, permanent magnet machines, AC synchronous machine, and switched reluctance machine have been considered. Some of these ISA systems utilise high DC bus voltages (for example 300V) whereas others utilise low DC bus voltage (i.e. 42V). Control methodology is closely tied-up with the machine topology of a particular system. The mechanical drive methodology of the reported ISA is either crankshaft mounted or offset coupled as discussed in the previous chapter.
Some of the proposals presented in literature will be briefly discussed in the rest of this chapter. In addition, a comparison of various electric machine topologies for integrated starter alternator will also be discussed.
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2.2 Improved Automotive Power Generation
2.2.1 Application of More Power Electronics to Lundell Alternator
This section discusses the application of more power electronics to conventional Lundell machine proposed in literature to achieve reduced load dump transients, increased efficiency and higher power output.
Namuduri et al [11] proposed an improvement to the field regulator of the Lundell alternator in order to reduce load dump transient of the 42V automotive power system. As shown in Figure 1.2 of Chapter 1, the conventional field regulator is a single quadrant PWM chopper. The proposed field regulator consists of a modified single quadrant chopper that decays field current quickly as the case with two quadrant chopper.
vDC
vref
Figure 2.1 Lundell power generation system with proposed fast field control
A schematic diagram of the system proposed in [11] is shown in Figure 2.1. Under normal operation Q2 is left ON and Q1 is switched to regulate the voltage. During over voltage, gate signal to both Q1 and Q2 are turned off. There is a small signal Zener diode connected between Drain and Gate of the MOSFET Q2. This Zener diode sets the clamp voltage of the MOSFET for quick inductive energy dissipation. MOSFET Q2 needs to have an adequate single pulse energy rating for this operation. Since the load dump transient is rather infrequent, a MOSFET can usually handle such single pulse events. The advantages of this scheme are low parts count, simplicity and low cost. Experimental results given in the paper indicates that it achieves over voltage requirements of the 42V PowerNet specification.
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Perreault et al [5] proposed an application of simple switch mode rectifier to Lundell alternator to achieve substantially higher level of power output compared to the conventional alternator. The proposed topology utilises duty cycle control of switch mode rectifier (SMR) as a second control in addition to the field regulator. The circuit arrangement of this method is shown in Figure 2.2.
v0
Ls
Figure 2.2 Lundell power generation system with semi-boost converter
As can be seen in Figure 2.2, Lundell machine stator is connected to semi-boost converter. The application of semi-boost converter allows varying the terminal voltage seen by the alternator while the DC voltage is regulated to desired value by changing the duty ratio of the three switches. In the proposed approach, the maximum power of the Lundell alternator is extracted by properly selecting the alternator field current (i.e. field regulator duty ratio) and SMR duty ratio as a function of output voltage and speed. A unique combination exist which produces maximum power at given speed. When maximum power is not required (i.e. light load operation), there is some degree of freedom for control of the alternator. Authors suggested that this degree of freedom can be used to maximise the efficiency under light load conditions. The experimental results presented in the paper demonstrate significant increase in power and efficiency of the proposed SMR based Lundell alternator system compared to the conventional system.
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Liang et al [25] presented a scheme that increases the output of the alternator at low speed by using a controllable rectifier. In the proposed method number of turns of the alternator winding is changed electronically in according to the operational speed. The paper reported 21% efficiency improvement (i.e. from 47% to 57%) in higher speed operation with this technique.
Hassan at el [26] proposed a dual output alternator system for power generation transition 42V/14V power system. The proposed system utilizes a conventional Lundell machine with dual output switch mode rectifier to obtain required 14V and 42V regulated outputs. The paper presented simulation and experimental results to demonstrate the feasibility of the system.
2.2.2 Application of Other Electric Machine Topologies
Despite various attempts to improve the Lundell alternator, it is difficult to achieve demanding requirement of future on-board power system due to the inherent limitation of this machine topology. As a solution to this problem, power generation systems that utilize various other machine topologies have been proposed.
v0
Figure 2.3 Power generation system with split winding PM machine
Naidu at el [27] proposed an application of Permanent Magnet (PM) machine with split stator winding as a generator for 14V system in order to achieve high power and efficiency. The proposed system consists of PM machine with split stator winding with a tap at 1/3 the turns. Each of two three-phase winding is connected to a separate phase
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Topologies controlled three-phase thyristor (SCR) bridge rectifier to regulate the alternator variable voltage to 14V. This scheme is shown in Figure 2.3.
Since outputs of the two SCR bridges are tied together, only one bridge needs to operate at any given time. At lower speed, the SCR bridge connected to the winding with higher number of turns operates and regulates the DC voltage. At higher speeds, the SCR bridge connected to the winding with lower number of turns regulates the DC voltage. Purpose for switching from higher turns to lower turns number at higher speed is to reduce the pulse current and hence the ohmic losses in the windings and thyristors in the bridge. Authors claimed that this switching between SCR bridges does not have any impact to the electrical system. The experimental results given in the paper demonstrate significantly high efficiency compared to the conventional Lundell system. Also, the paper claims 27% improvement in the power density compared the conventional Lundell technology.
SWITCH MODE RECTIFIER
+
PM Machine v0
Qz
-
Figure 2.4 Power generation system based on PM machine with switch mode rectifier
Soong et al [28] proposed an application of PM motor with Switch Mode Rectifier in order to achieve high power generation in the automobile. The switch mode rectifier allows regulating DC voltage in spite of the varying terminal voltage due to the speed variation. The paper discusses the optimal design of the interior permanent magnet machine in order to achieve the required constant power speed range. The machine design parameters that influence the constant power speed range of this alternator are:
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Topologies the back-EMF ratio (the ratio of the back-EMF at maximum speed to the rated output voltage), and the saliency ratio. A diagram of proposed scheme is shown in Figure 2.4. The experimental results presented in the paper demonstrate high efficiency in a wide range of speed.
Naidu at el [7] proposed an induction machine based automotive power generation system. The system consists of diode-bridge to rectify active power component; and PWM Voltage Source Inverter (VSI) to supply excitation (i.e. excitation reactive power) to the induction machine. A diagram of this scheme is shown in Figure 2.5.
v0
Figure 2.5 Induction machine based power generation system with diode-bridge and PWM VSI
Since the PWM VSI needs to supply only the excitation current, the VA rating of the inverter required is lower compared to the induction machine rating. The capacitor on the DC bus of the VSI is charged to the battery voltage before starting the engine. When in operation, the voltage of the capacitor is regulated to a preset value. The output voltage of the diode bridge is regulated to the desired value (42V in this case) by controlling the excitation of the induction motor. The simulation results presented in the paper indicates substantially high power capability and efficiency compared to the conventional Lundell alternator topology.
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2.3 Integrated Starter Alternator Topologies
This section briefly reviews design and analysis of various integrated starter alternator system topologies reported in literature.
Toyota launched the first commercially available vehicle with 42V integrated starter alternator system to the Japanese market in August 2001. Teratani at el presented a paper [29] on the development of this vehicle including Motor/Generator (M/G) system. The electric machine used for the M/G is an AC Synchronous machine with rating of 36V, motoring 3.0kW, and generating 3.5kW. The electric machine is side mounted in the place of conventional alternator and connected to the crankshaft pulley of engine using ribbed V-belt. A solenoid clutch is employed between the crankshaft pulley and engine to disconnect on demand. This is required to run the auxiliaries such as air conditioner when engine is stopped during stop/restart function. The system consists of two batteries; 36V battery which is connected to the 42V DC bus directly, 12V battery which is connected to the 42V DC bus via a DC/DC converter. The conventional 12V starter motor exists in the system to provide initial cranking torque (i.e. cold cranking torque) for the engine starting. M/G is used for starting the engine during stop/restart function. In this system, the M/G cranks the engine virtually up to idle speed before firing initiates to allow smooth start of the engine. The motor generator system is also used for reducing the vibration during stopping by controlled stop instead of conventional way of stopping by merely cutting the fuel supply. In the applied controlled stop method, the engine is driven by the electric machine at idle speed for predetermined time after fuel supply is cut off until the sufficient negative cylinder pressure is achieved. When the sufficient negative pressure is achieved, the speed of the electric machine is gradually reduced under operation of electric machine. When the speed is dropped below 300rev/min electric machine switch to generation mode and negative torque (i.e. braking torque) is applied during cylinder expansion cycle immediately before the rotation stops. This low speed braking action prevents vibration due to low speed engine resonance. The results shown in the paper claims significant reduction in vibration compared to the conventional stop (i.e. stop merely by cutting off fuel supply). Figure 2.6 shows speed profile during conventional and controlled stopping.
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Figure 2.6 Engine stopping characteristics (a) conventional stop by merely cutting off fuel supply (b) controlled stop for reducing vibrations [29]
Toyota M/G system also facilitates the regenerative braking as well as regeneration during travelling down-hills (i.e. opportunity charging). Figure 2.7 shows battery terminal voltage and charging current during regenerative braking using M/G system.
Figure 2.7 Battery voltage and current during regenerative braking [29]
Aoki et at [13] presented development of mild hybrid system used in Honda Insight Hybrid coupe. The paper called the system as “Integrated Motor Assist” or IMA. The system consists of 10kW three-phase Permanent Magnet (PM) synchronous machine, air cooled power electronics arrangement and 144V Ni-MH battery pack. The power electronics arrangement so called “Power Control Unit (PCU)” has an inverter for controlling the electric machine as well as DC/DC converter to supply power to 12V loads available in the automobile. The PM machine of the IMA system is mounted on
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Topologies the crankshaft. The width of the PM machine is only 60mm which allows mounting of the machine with minimum modification to the existing engine structure. The magnet type used for the PM machine rotor is neodymium-sintered. The PM machine uses a split stator structure with salient pole centralized windings to enable reducing motor axial width and losses compared to a stator with usual wave winding. The system facilitates recovering of deceleration energy, idle stop function and power assist to the engine.
Figure 2.8 Power assist performance of IMA system in Honda Insight coupe [13]
Figure 2.9 Cranking performance of IMA system in Honda Insight coupe [13]
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Figure 2.8 shows the published performance for the power and torque capability of the IMA. Figure 2.9 shows the cranking performance of cold (engine water temperature at - 15oC) and warm (engine water temperature at 25oC) conditions. The cranking performance of conventional starter motor is also given in the figure for comparison.
Shibutani et al [30] presented further development made to Honda IMA system for adoption to Honda Civic mild hybrid car. In the improved PM machine, the stator was wound with lager diameter conductor in asymmetrical manner to increase the space factor by 27% and thus reduces the copper losses by 30% compared to the Insight IMA without changing the diameter of the motor. The paper claims that this improvement is achieved by applying new material and design. In addition to the improvement to the electric machine, the other components of the system such as IPU (i.e. the power electronics) and battery pack are also improved to achieve more compact and low weight system. A comparison of torque speed characteristics and efficiency of the Insight IMA machine and the improved Civic IMA machine is shown in Figure 2.10.
Figure 2.10 A comparison of torque capability and efficiency (a) Insight IMA electric machine (b) Civic IMA electric machine [30]
Soong at el [16] presented designing of an interior permanent magnet (IPM) machine for offset coupled (belt driven) ISA. The paper presents a conceptual design of a machine that provides wide field weakening range greater than 10:1. The test machine with proposed rotor design has unsaturated saliency ratio (Lq/Ld ) of 6. The paper presents estimated characteristics of torque, output power and input power factor Vs speed for both motoring and generation. These estimations were performed based on the 33 Chapter2: Reviews of Improved Automotive Power Generation and Integrated Starter Alternator
Topologies experimental results obtained by fixed speed variable-voltage dynamometer test. Estimated results for variable-speed constant voltage indicate wide constant speed range greater than 10:1. A comparison of estimated results and calculated performance obtained from the equivalent circuit parameters was also presented in this paper.
Lovelace at el [31] discussed the mechanical design issue of radially laminated IPM for possible use for integrated starter alternator. This paper mainly examines and mitigates mechanical stress of the rotor due to the centrifugal forces at higher speeds. The paper compares the various parameters including cost of the optimised IPM design for integrated starter generator application with and without mechanical consideration. de Varies at el [32] proposed a Switched Reluctance Machine (SRM) based integrated starter alternator. The proposed system utilizes a 24/16 three-phase SRM with a three leg inverter. The locked rotor test data shown in the paper indicates that it can produce a torque above 160Nm instantaneously which is sufficient for starting the engine. It is claimed in the paper that at 100rev/min with firing angle of -23o and 160o voltage pulse duration, average torque of 130Nm can be achieved in motoring mode. The paper does not provide the experimental results for 24/16 SRM. The experiments in generation mode were conducted with small 6/4 SRM. A constant output current control scheme based on simple empirical equations was used in generation mode.
Rehman at el [33] presented induction machine based integrated starter alternator that uses an ultra capacitor as main power source so that the need for solely dependence on the battery is eliminated. Ultra capacitors have benefits of light weight, longer life than conventional batteries, high current capability even close to voltage limit and possibility of judging existing charge precisely by measuring the voltage. The proposed system has DC bus of nominal 250V with three 1F, 100V ultra capacitors connected in series. The DC bus is connected to 36V and 12V batteries using two separate DC/DC converters. The induction machine used for the ISA is rated for 8kW. The designed starting torque is 280Nm. The controller utilises indirect field oriented control with gain scheduling for slip gain and torque scaling factor in order to compensate the parameter variations.
Leonardi at el [34] presented a study on comparing performance Low and High voltage integrated starter generator systems. In the study, three prototype induction machines with different stator winding construction are compared. Two machines are with stator
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Topologies winding designed to suit 42V DC bus voltage and driven by a MOSFET inverter. The stator of the one machine is wound with lower turns per coil with bigger cross-sectional area conductors. The other one is wound with higher number of turns with smaller cross-sectional area conductors. The stator winding of the third machine is designed for 300V DC bus voltage and driven by an IGBT inverter. The amount of copper in each stator windings is maintained to be the same. Various performance figures such as efficiency, cranking torque and power assist is compared. It can be seen from the study that the system voltage for ISA system has various trade-offs and the final choice will probably depend on the cost and safety considerations for a particular application.
Henry at el [35] presented a development of belt driven starter alternator system for future 42V PowerNet. The paper discusses the advantages of belt drive compared to other mechanical power transmission methods such as chain drive and gear drive. The advantages are lower cost, no requirement for lubrication, very low noise and more freedom for mounting location. The paper elaborates various design aspects of the belt drive for the ISA application. The ISA system was modelled for both motoring and generation. Simulation results for cranking at -20oC indicates that engine reaches 600rev/min in about 6 cranking cycles (i.e. 2160o) or approximately 800 milliseconds. The maximum cranking speed is limited to about 600rev/min as a result of low battery voltage at low temperature. The simulation assumed battery voltage of 18V at -20oC. At room temperature the engine speed reaches 600rev/min within less than 3 cycles. The cranking speed can be as high as 1200rev/min as a result of the availability of high battery voltage. The paper also presented simulation results for torque production at low battery voltage. The simulation result indicates that the induction machine can produce its maximum torque with slight delay at low battery voltage condition (i.e.18V).
Ly et al [36] presented optimal control of an induction machine based integrated starter alternator. The proposed ISA utilises a 4kW induction machine which is directly coupled to the engine crank shaft. The control of ISA is based on scalar control with lookup tables that provide optimum voltage and frequency reference to the inverter. The paper includes simulation and experiment results that demonstrate the improvements achieved by the optimum control.
Lui et al [37] presented design and control of high current 10kW inverter for integrated stater generator application for possible use in existing 14V automotive power system.
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The voltage range of the DC bus of the inverter is 12-30V. The proposed inverter utilizes parallelled MOSFET to achieve 1500A peak current which is required during starting. The inverter is cooled by air. The paper has presented simulation and experimental results to demonstrate the operation of the proposed inverter. The experimental results verify the inverter operation at 1500A peak current. In addition heat sink temperature rise curve for continuous operation is also presented.
Xu et al [38] presented a comparison of DC-DC-AC combined converter topologies for integrated starter generator application. This paper compares Z-source converter, Cuk- converter based DC-DC-AC combined converter and cascaded DC-DC and VSI topology. Simulation results for these topologies are presented in this paper. Advantages and disadvantages of each topology were discussed and it indicates that the simple cascaded DC-DC and DC-AC combination is an attractive solution for an ISA application.
2.4 Comparison of Various Electric Machine Topologies Proposed for Integrated Starter Alternator
As discussed in the previous section, there are a number of machine types have been considered for ISA application; namely, induction machine, permanent magnet machine, AC synchronous machine and switched reluctance machine. Pros and cons of each machine type will be discussed in following sections.
2.4.1 Permanent Magnet Machine
There are two types of PM machine of interest; surface mounted PM machine (SPM) and interior PM machine (IPM). Interior PM machine is more suitable for ISA application since wider field weakening speed range can be achieved with IPM machine compared to SPM machine. Although the SPM combining with concentrated and fractional winding has been developed for variable speed application, the speed ratio still remain relatively low compared to IPM machine [15]. This limitation can be overcome with an over-rated inverter (i.e. higher voltage power electronics) and an extra DC/DC converter [28]. However, this will not be economical since the lower utilization of power electronic devices. In IPM machine, there are reluctance torque component in addition to the magnet torque and in some cases the reluctance torque is
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Topologies higher than the magnet torque. These types of machines provide wide field weakening speed range and are suitable for application such as ISA. PM machines are usually high efficient due to absence of rotor resistance losses. Also, PM machines are usually compact and have high torque/volume and torque/weight ratio. The motoring and generator operation of the PM machine are achieved by controlling the power angle ( ) of the PM machine via vector control or other means. The positive power angle (i.e. 0 ) corresponds to the motoring whereas the negative power angle (i.e. 0) corresponds to generator operation of the PM machine.
The cost is one of the main concerns in automotive industry. PM machine is still not economical enough for automotive industry. PM machine designs that would satisfy the requirement of ISA are complex and still in evolving stage. PM machines have two troublesome fault modes; high latch-up torque in the event of machine or power electronics short circuit and the possibility of uncontrolled generator operation in the event of power electronics shut down in field weakening region [22, 23]. This uncontrolled generation causes over-voltages on the inverter and may damage the power electronics. One solution is to over-rate the inverter to highest voltage anticipated. However, this is a costly solution as the higher voltage power electronics are expensive. The rotor magnet flux of PM machine is strongly dependent on the temperature. The reduction of torque in constant torque region due to temperature may need to be considered. In addition, high temperature operation also presents risk of demagnetization of the rotor magnets. It is still challenging to achieve sufficiently wide field weakening (FW) region with IPM. The wide FW region requires complex machine design and the FW problem has not yet been fully resolved despite various efforts reported in the literature. In field weakening region, the field weakening current command may be required to adjust with the temperature to achieve optimal field weakening in PM machine. PM machine becomes less efficient in FW region. This is because, to weaken the flux, a negative direct axis stator current is required to apply and this additional direct axis current causes copper losses in the stator winding. PM machines also have more stringent position sensing requirement due to already established magnet flux and the controller must detect and track its position [22].
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2.4.2 Induction Machine
Induction machine is one of the most popular electric machine topologies due to its robustness, mature technology [16]. This mature manufacturing technology allows low manufacturing cost compared to the most of the other machine topologies. Induction machine produces a smooth torque and less noise and has relatively good efficiency. Induction machine has inherent protection against terminal short circuit fault in generation operation as short circuit removes the magnetising current component. In addition induction machine does not have over-voltage problem that PM machine may be subjected to in field weakening operation. The motoring and generator operation of induction machine is achieved by controlling the slip ( s ) of the machine via vector control or other means. The positive value of slip (i.e. s 0 ) corresponds to the motoring whereas the negative slip (i.e. s 0 ) corresponds to the generator operation.
Induction machine has lower torque/ volume and torque/weight ratio compared to the PM machine. Constant power operation of induction machine can be obtained usually within speed ratio of 1:3 - 5. In higher speeds, maximum power output is reduced due to leakage inductance of the induction machine. However, this may not have great impact on ISA application because ISA machine does not require to run at higher speed without compromising the output power [15].
2.4.3 Switched Reluctance Machine
Switched reluctance machine has a robust rotor structure and can achieve wide speed range [16]. The rotor has low moment of inertia and no rotor winding benefits high speed operation [15]. One of the attractive features of the switched reluctance drives for automotive application is its insensitivity to the high temperature [39]. Switched reluctance machine also has good efficiency characteristics. The motoring and generator operation of switched reluctance machine is achieved by applying current to the stator windings at correct region of the inductance vs angle curve (i.e. Lf () ). The positive gradient of the inductance vs angle curve (i.e. dL d 0 ) corresponds to
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Topologies motoring operation where as the negative gradient (i.e. dL d 0 ) corresponds to the generator operation.
The switched reluctance machine has disadvantages of torque ripple, vibration and acoustic noise [39]. The torque ripple and vibration are caused by the salient rotor. Radial magnetic field is the main cause for the acoustic noise. These effects can be somewhat compensated by proper selection of number of phases, poles and shaping the current pulse waveform. However, the shaping of current waveform may cause reduction in the torque capability of the machine.
2.5 Summary
This chapter reviewed main power generation and integrated starter topologies reported in the literature. Various improvements proposed for power electronics circuit of Lundell alternator were discussed. The main improvements reported in the literature focus on reducing the load dump transients, and increasing the output power and efficiency. Application of other electric machine and power electronic circuit topologies have also been considered for automotive power generation due to the inherent limitation of Lundell machine topology. Moreover, integrated starter alternators which combines the function of starter motor and alternator have been proposed in literature. Various electric machine topologies such as induction machine, permanent magnet machine and switched reluctance machine have been considered for the integrated starter alternator application. These machine topologies have various advantages and disadvantages. The selection of power electronics circuit configuration is tied up with the machine topology. The design of the electric machine and power electronics circuit depends on the application and required specification of the ISA. There exists a trade- off between electric machine and power electronics circuit design. A collective approach for design of ISA system should be more appropriate instead of individual design approaches for electric machine and the power electronics circuit.
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CHAPTER 3
INTEGRATED STARTER ALTERNATOR SYSTEM MODELLING
3.1 Overview
This chapter discusses the modelling of subsystems of the ISA system proposed in this thesis. The modelling of the subsystem of ISA is essential for the systematic control design for desired dynamic operation; and also for computer simulations which provide deep insight into the behaviour of the system. The main components of the proposed ISA in this thesis are: induction machine, three-phase inverter, battery and internal combustion engine (ICE). The models discussed in this chapter are used throughout this thesis for computer simulation studies and designing of various controllers. The induction machine of the ISA is modelled for both dynamic and steady-state conditions using two axes models (i.e. dq models) in stationary and synchronous reference frames. The dynamics and losses of the three-phase inverter are modelled separately. The dynamic model represents the dynamic relationship between AC and DC sides of the inverter in dq quantities. The losses of the inverter (conduction and switching losses) are also represented in terms of synchronous reference frame steady state dq quantities. The battery of the ISA system is modelled using resistive and capacitive (i.e. RC) components that represent main storage, surface charges and internal resistances. This battery model is represented in state-space and it provides adequate accuracy for the ISA system modelling. The internal combustion engine is modelled to represent cranking operation. The fuel supply into the cylinders of the engine during cranking operation is ignored for the simplicity. During running operation (i.e. generation operation of ISA), the engine speed is assumed to be stiff, for simplicity. Detailed modelling of each sub-system will be discussed in the rest of the chapter.
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3.2 Induction Machine Modelling
This section presents detailed modelling of the induction machine. For the purpose of analysis and control design of integrated starter generator, which demands challenging performance requirements, it is important to use a model that incorporates all the significant dynamic effects occurring during transient operation as well as in the steady state operation. A model valid for instantaneous variation of voltage and current; and adequately describes the behaviour and performance of the induction machine under both transient and steady state operation can be obtained by the utilization of space- phasor theory [40]. The derivation of induction machine model that utilizes space- phasor theory is described in Appendix A. The induction machine model assumes smooth air gap and symmetrical machine with sinusoidal distributed windings and is represented in arbitrary reference frame (i.e. reference frame rotating at arbitrary angular velocity). In this thesis, two special cases of arbitrary reference frame model is utilised; namely, stationary reference frame and synchronous reference frame [40-42]. In stationary reference model, the reference frame is fixed to the stator. In synchronous reference frame model, the reference frame rotates at synchronous speed. In addition to theses dynamic models, the steady state equivalent circuit model is also derived from the arbitrary reference frame complex space phasor model.
The next section discusses the dynamic model of induction machine ISA in stationary reference frame.
3.2.1 Dynamic Model in Stationary Reference Frame
This section discusses the representation of dynamic model of induction machine in stationary reference frame. The induction machine model in stationary reference frame can be obtained by substituting g 0 to the arbitrary reference frame model derived in Appendix A.
By substituting g 0 and subscript gs into (A.37), (A.38), (A.39) and (A.40) in Appendix A, and separating real and imaginary components of the complex space- phasors, stationary reference frame qd model can be obtained as follows:
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d vRis ss (3.1) qs s qsdt qs
d vRis ss (3.2) ds s dsdt ds
d 0 Ri' s ss (3.3) rqrdt qr r dr
d 0 Ri' s ss (3.4) rdrdt dr r qr
s ss qsL siLi qs m qr (3.5)
s ss dsLi s ds L m i dr (3.6)
s ss qrL riLi qr m qs (3.7)
s ss drLi r dr L m i ds (3.8) where, s s vvqs, ds are stator voltages of q and d axis in stationary reference frame.
s s iiqs, ds are stator currents of q and d axis in stationary reference frame.
s s iiqr, dr are rotor currents of q and d axis in stationary reference frame.
s s qs, ds are stator flux linkage of q and d axis in stationary reference frame.
s s qr, dr are rotor flux linkages of q and d axis in stationary reference frame.
' Rs , Rr are stator and rotor resistances
Lm is magnetising inductance LLLs sl m ; Lsl is stator leakage inductance ' ' LLLrrlm; Lrl is rotor leakage inductance r is electrical angular velocity of the rotor.
The induction machine used in the integrated starter alternator discussed in this thesis is a squirrel cage induction motor. Therefore, the rotor voltage in (3.3) and (3.4) is taken to be zero.
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The electromagnetic torque given in (A.44) of Appendix A can be written in stationary reference frame as follows:
L 3 P m s sss Tiiem() qs dr ds qr (3.9) 22Lr
The relationship between instantaneous three-phase stator voltages and qd axis stator voltages in stationary reference frame given in above equations can be written as follows:
FV 11FV s GW1 vas FVv GW qs 2 GW22 GWs GWvbs (3.10) v 3 GW HXds 33GW GW0 HXvcs HX22
Similarly, for stator currents,
FV 11FV s GW1 ias FVi GW qs 2 GW22 GWs GWibs (3.11) i 3 GW HXds 33GW GW0 HXics HX22
Figure 3.1 shows qd axis equivalent circuit representation for dynamic model of induction machine in stationary reference frame.
The stationary reference frame state-space qd model for induction machine based on stator current and rotor flux state variables can be obtained by substituting g 0 and subscript gs into (A.56) in Appendix A, and separating real and imaginary components of the complex space-phasors as follows:
FVRLRL'' smrm GW''0 2'r GWLLssLLrsLrF 1 V G 0 W FViissGWRLLR''FV L' GWqs GW0 smmr GWqs G s W ss''r '2 G W FVs d GWiiGWLLLLLssrsrGW 1 v ds ds GW0 GWqs (3.12) GWssGW' ' GW ' s dt RL R GWLs HXv GWqr GWrm 0 r GWqr ds HXGWssGWLLr HXGWGW00 dr GWrr dr GW GWRL' R' H 00X 0 rm r GWr HXLLrr
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where,
CSL2 ' ' m Ls is stator transient inductance and given by LLssDT1 and EULLrs
2 CSL '' m RRRssrDT. EULr
' s ' L rdr Rs sl Lrl Rr s s iqs iqr
s s v Lm s qs qs qr
s L ' rqr ' Rs sl Lrl Rr s s ids idr
s v s Lm s ds ds dr
Figure 3.1 qd axis equivalent circuit representation of dynamic model for induction machine in stationary reference frame
3.2.2 Dynamic Model in Synchronous Reference Frame
This section presents the representation of dynamic model of induction machine in synchronous reference frame. This is one of the most important reference frames used for vector control of AC machines. Different vector control techniques uses different synchronous reference frames such as rotor flux oriented reference frame, stator flux oriented reference frame and magnetising flux reference frame. The induction machine
44 Chapter3: Integrated Starter Alternator System Modelling
model in synchronous reference frame can be obtained simply by substituting g e to arbitrary reference frame model derived in Appendix A.
By substituting g e and subscript ge into (A.37), (A.38), (A.39) and (A.40) in Appendix A, and separating real and imaginary components of the complex space- phasors, synchronous reference frame qd model can be obtained as follows: d vRiee e e (3.13) qs s qsdt qs e ds d vRiee e e (3.14) ds s dsdt ds e qs d 0()Ri' ee e (3.15) rqrdt qr e r dr d 0()Ri' ee e (3.16) rdrdt dr e r qr ee e qsLi s qs L m i qr (3.17)
ee e dsLi s ds L m i dr (3.18) ee e qrLi r qr L m i qs (3.19)
ee e drLi r dr L m i ds (3.20) where,
ee vvqs, ds are stator voltages of q and d axis in synchronous reference frame.
ee iiqs, ds are stator currents of q and d axis in synchronous reference frame.
ee iiqr, dr are rotor currents of q and d axis in synchronous reference frame.
ee qs, ds are stator flux linkage of q and d axis in synchronous reference frame.
ee qr, dr are rotor flux linkages of q and d axis in synchronous reference frame.
e is the speed of rotation of synchronous reference frame.
The electromagnetic torque given in (A.44) of Appendix A, can be written in synchronous reference frame as follows:
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L 3 P m ee ee Tiiem() qs dr ds qr (3.21) 22Lr
The relationship between instantaneous three-phase stator voltages and qd axis stator voltages in synchronous reference frame given in above equations can be written as follows:
FV24FV cos cos( ) cos( ) vas FVve 2 GWGW qs GW33 GWe GWvbs (3.22) HXv 3 GW24 ds sin sin( ) sin( ) HXGWv HXGW33cs
Similarly, for stator currents,
FV24FV cos cos( ) cos( ) ias FVie 2 GWGW qs GW33 GWe GWibs (3.23) HXi 3 GW24 ds sin sin( ) sin( ) HXGWi HXGW33cs
The derivation for (3.22) and (3.23) are given in Appendix A.6.
Figure 3.2 shows qd axis equivalent circuit representation of dynamic model for induction machine in synchronous reference frame.
The synchronous reference frame qd axis state-space model of induction machine based on stator current and rotor flux sate variables can be obtained by substituting g e and subscript ge into (A.56) in Appendix A, and separating real and imaginary components of the complex space-phasors as follows:
FVRLRL'' smrm GW''er2' GWLLLLLssrsrFV1 GW0 FViieeGWRLLR''FV L' GWqs GW smmrGWqs GWs eeer'' '2 GWFVe d GWiiGWLLLLLssrsrGW 1 v ds ds GW0 GWqs (3.24) GWeeGW' ' GW ' e dt RL R GWLs HXv GWqr GWrm 0 r GWqr ds GWHXeeGWLLerHXGWGW00 dr GWrr dr GW GWRL' R' HX00 0 rm r GWer HXLLrr
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e ' e ' eds L ()erdr Rs sl Lrl Rr e e iqs iqr
e e v Lm e qs qs qr
e e ' ' eqs L ()erqr Rs sl Lrl Rr e e ids idr
e v e Lm e ds ds dr
Figure 3.2 qd axis equivalent circuit representation of dynamic model for induction machine in synchronous reference frame
3.2.3 Steady-state Equivalent Circuit
Steady state equivalent circuit of induction machine can be obtained from the complex variable dynamic model given by (A.45) and (A.46) in Appendix A.
By substituting stationary reference frame conditions (i.e. g 0 and subscript gs) g and vr 0 (for squirrel cage induction machine) to (A.45) and (A.46),
dd vRiLs ss iL () ii ss (3.25) s sslmssdt dt sr
dd 0(Ri''s L issss L i i) j (3.26) rrlmrrdt dt srr r
s ss By substituting rrsLrmiLi to (3.26) and rearranging,
dd jLiLiRiLiL ()s ss'' s () ii ss (3.27) rrrs m r r rldt r m dt sr
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Lets define f s is a complex dynamic variable of the machine and F is the steady state s s variable that corresponds to f s . The relationships between them can be written as s follows:
f s F (3.28) s s
d f s jF (3.29) dt s es
By applying relationships given in (3.28) and (3.29) to voltages and current, and substituting them into (3.25) and (3.27),
VRIjLIjLIIs ss e sls e m() s r (3.30)
'' j rrrmsrrerlremrs()LI L I RI j L I j L() I I (3.31) where,
Vs is the steady state peak phase voltage
Is is the steady state peak stator current
Ir is the steady state peak rotor current
e is the stator frequency
The above equations (3.30) and (3.31) are also valid for RMS values for Vs , Is and Ir .
By simplifying (3.31),
'' 0()()RIrr jLI rlr e r jL m I r I s() e r (3.32)
The slip of the induction machine ( s ) is given by:
s er (3.33) e
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By substituting (3.33) into (3.32),
R' 0(r I jLIjLI' I) (3.34) s rerlremrs
Figure 3.3 shows steady-state per-phase equivalent circuit of an induction machine. The core losses of the induction machine is usually represented by a resistor parallel to the magnetising inductance. Incorporation of core losses to the induction machine model will be discussed in Chapter 8 of this thesis.
L ' I Rs sl Lrl s Ir
ImsrII E ' V m Rr s L m s
Figure 3.3 Steady state per phase equivalent of induction machine.
3.3 Three-Phase Inverter Modelling
This section discusses the modelling of the three-phase inverter used in the ISA system. Inverter modelling, in the context of ISA application, can be divided into two parts, namely, dynamic modelling and loss modelling. Theses two models are treated separately in this thesis. The dynamic model is useful for systematic design of the ISA controller that is required to obtain desired dynamic performance. The inverter losses are assumed to be negligible in the dynamic model. On the other hand, the inverter loss model assumes the steady-state operation of the inverter (i.e. neglects the dynamics). Given the high working current in low voltage ISA application, loss modelling is important for investigating variation of inverter losses at various operating conditions of ISA.
Section 3.3.1 and 3.3.2 discuss the dynamic model and loss model respectively, of three phase inverter of the ISA.
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3.3.1 Dynamic model for three-phase inverter
Figure 3.4 shows a schematic diagram of the three-phase inverter. The three-phase inverter of ISA consists of six switches with anti-parallel diodes and a DC bus capacitor
(Cdc ) to filter out the switching ripple. The AC-side voltages and currents of the inverter are equal to the stator voltages and currents of the induction machine. Inverter voltages of phase a , b and c are vas , vbs and vcs , respectively. The inverter currents are ias , ibs and ics . The DC-side voltage and current are vDC and iI respectively. Rdc represents the leakage resistances of the DC bus capacitor Cdc . The currents through the capacitor and leakage resistance are iCAP and iR respectively. The current between battery and DC bus capacitor is iDC . The load current drawn from the DC bus is iL and the battery current is iBAT . All of these components are indicated in Figure 3.4. The directions of currents marked in the schematic diagram correspond to the motoring operation of ISA. Nevertheless, the model derived from this schematic diagram can be used for both motoring and generation modes of operation.
A i DC iI
iL i iCAP BAT iR ias vas R C a v dc dc vbs DC b i v c bs cs ics
' A Figure 3.4 Schematic diagram of three phase inverter
The dynamic inverter model is based on power balance of AC- and DC-sides. By neglecting losses of the inverter, power balance can be written as follows[43]: viDC I vias as vi bs bs vi cs cs (3.35)
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Equation (3.35) can be transformed into qd -stationary reference frame and written as follows:
3 v i() vis sss vi (3.36) DC I 2 qs qs ds ds
Similarly, (3.35) can be written in qd -synchronously rotating reference frame as below.
3 v i() viee vi ee (3.37) DC I 2 qs qs ds ds
A relationship between current and voltage of the DC-side of three-phase inverter can be obtained by applying KVL and KCL to the DC-side of the inverter.
By applying KCL to DC-side of the inverter, iiI DC i CAP i R (3.38)
The currents through DC capacitor and leakage resistance in (3.38) can be replaced by voltage expression as follows:
dv v DC DC iiIDC Cdc (3.39) dt Rdc
By combining (3.37) and (3.39), and rearranging,
2 dv v P 3 DC DC DC ee ee vvDC ()qsi qsv dsi ds (3.40) dt Rdc C dc C dc2 C dc
' where PDC is the power flow into the inverter across AA shown in Figure 3.4.
Equation (3.40) describes the dynamic relationship (i.e. dynamic model) between stator variables of induction machine and DC side variables of the inverter. Combination of induction machine and inverter model will be discussed in the Chapter 6 and this model will be used for designing proposed new DC voltage controller
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3.3.2 Loss Model for Three-phase Inverter
Main losses in the inverter are twofold; switching and conduction losses. Switching losses are caused by turning ON/OFF characteristics of switching devices and diodes. The conduction losses are caused by the on-state voltage drops and on-state resistance of the switching devices and diodes [44, 45]. Section 3.3.2.1 discusses inverter switching losses in detail and represents switching losses in terms of steady state AC side current, DC bus voltage and switching frequency. Section 3.3.2.2 discusses the inverter conduction losses in detail and represents in terms of steady state AC side current, voltage and power factor. Finally, in Section 3.3.2.3, the total converter losses are represented in terms of steady state dq axes that will be used in Chapter 8 for loss minimisation control of ISA.
3.3.2.1 Switching Losses of the Inverter
The switching losses are caused by the switching characteristics of the devices. Energy losses occur during switching ON process as well as switching OFF process as shown in
Figure 3.5. vx and ix are voltage across the switch when fully OFF and current through the switch when fully ON, respectively. i i i x vi x vi i v vx v vx 0 0
Figure 3.5 Voltage and current during (a) switching ON (b) switching OFF
The switching energy losses per switching cycle can be obtained as follows:
eeevidtvidton/ off on off OO (3.41)
tton off where,
eon is the energy loss during switching-on,
eoff is the energy loss during switching-off and
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eon/ off is the total energy loss per switching cycle.
Switching energy loss in an active switch or diode is increased with the voltage and current involved in the switching action. An approximate linear relationship can be assumed for total switching energy losses per cycle against currents and voltages as shown in Figure 3.6.
eeeon/ off on off eeeon/ off on off
EDATA EDATA
I DATA i x VDATA vx
Figure 3.6 Total switching energy loss per one cycle Vs (a) current at constant voltage (b) voltage at constant current.
The relationship between total switching energy loss per cycle with voltage and current can be written as follows:
eivAvion/ off(, x x ) x x (3.42)
E where A is assumed to be a constant and it can be calculated as A M using VIMM switching loss of a switch ( EM ) measured under voltage,VM and current IM .
Switching power loss ( pswi_ ) can be calculated by averaging the ON+OFF energy losses throughout the switching cycle as follows:
eiv(, ) on/ off x x pswi_ Avxxsw i f (3.43) Tsw
where, Tsw and fsw are switching period and switching frequency respectively.
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ice
vce
ia 0 VDC ia
v ak T i ak
Figure 3.7 A leg of a three-phase PWM inverter with sinusoidal output current
Figure 3.7 shows one leg of three-phase PWM inverter. As indicated in this figure, only switch T1 or diode D2 conduct during positive half cycle of current sinusoid where as switch T2 or diode D1 conduct during negative half cycle of current sinusoid. Each switching cycle occurring in positive half cycle of output current consists of following actions.
Action-1: Simultaneous switching-on of T1 and switching-off of D2
Action-2: Simultaneous switching-off of T1 and switching-on of D2
The currents through T1 and D2 for a positive output current are shown in Figure 3.8. As can be seen, two currents possess inverted nature and they are added together to form the output current. The sketch of switching waveforms and corresponding energy losses of switching action-1 and action-2 are illustrated in Figure 3.9 [46].
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ia
T sw Figure 3.8 Conduction of active switch T1 and diode D2 in positive half cycle of current
ia ia vice ce ice ice vice ce vce
VDC VDC vce 0 0
ia ia iak i viak ak ak viak ak
0 0 vak V V v DC DC ak Figure 3.9 Sketch of typical switching waveforms and associated energy losses in positive half cycle of current
As may be seen in Figure 3.9, the switching ON loss of D2 is small and hence neglected. Total switching energy losses for one leg per switching period in current positive half cycle can be obtained as follows:
eetot onT111 e offT e offD AviAvi T DC a D DC a (3.44)
where, AT and AD are constants and can be estimated as described in (3.42).
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By assuming the DC bus voltage is constant, the average switching power losses per leg during positive half cycle of current at steady state can be obtained as follows:
TTT 222222e tot psw OOOpdtsw_ i dt AviAvifdtT DC a D DC a sw TTTT000sw T 2 2 O AvT DC Av D DC f sw I peak sin tdt (3.45) T 0 2 f sw Av Av I T DC D DC peak
During negative output current half cycle, active switch T2 and diode D1 conduct and dissipate same amount of average switching power loss per leg in steady state as given in (3.45). The three-phase inverter consists of three legs. By assuming the steady-state operation of inverter, the total switching power loss can be written as follows:
6 f p sw Av Av I (3.46) tot_ sw T DC D DC peak
3.3.2.2 Conduction Losses of the Inverter
Conduction losses are caused by the forward voltage drops of active switches and diodes in the inverter [47, 48]. Forward conduction voltage drops of active switches and diodes are functions of current and can be represented approximately as follows:
vVce ceo ir ce ce (3.47)
vVirf fo ak f (3.48) where
vce and v f are the forward conduction drops of active switch and diode respectively,
Vceo and Vfo are the saturation voltage drops of active switch and diode respectively and
rce and rf are the dynamic forward resistance of active switch and diode respectively.
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ia vapwm_
va
DTsw
Tsw
T Figure 3.10 Conduction pattern of active switch T1 and diode D2 during positive current half cycle
Figure 3.10 shows conduction pattern of active switch T1 and diode during positive current half cycle. As can be seen in this figure, active switch T1 and diode D2 conduct for DTsw and (1DT ) sw time during each switching period.
Let output current of the leg, ia be,
iIapeaksin t (3.49)
The relationship between duty cycle ( D ) of the active switch T1 and modulation function ( m ) can be written as follows:
1 Dm(1 ) (3.50) 2
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The steady-state modulation function for SPWM (Sinusoidal Pulse Width Modulation) can be written as follows:
mMsin( t ) (3.51)
The average conduction power losses of active switch T1 during positive half cycle of current ( PTav1_ ) can be obtained as follows:
TT 22 Vi ir2 (1 m ) 22 ceo a a ce PTav1_ OO v cea i D dt dt TT002 T 2 1 22 O VIceo peak sin trIce peak sin t (1 M sin( t )) dt (3.52) T 0 VI CSCS rI2 2 ceo peak DTDT1cosMM ce peak cos 24 EUEU 4 24 4 3
The average conduction power losses of diode D2 during positive half cycle of current
( PD2_av ) can be obtained as follows:
TT 22 Vi ir2 (1 m ) 22fo a a f PDav2_ OO v fa i(1 D ) dt dt TT002 T 2 1 22 O VIfo peak sin t rIf peak sin t (1 M sin( t )) dt (3.53) T 0 VI CSCS rI2 2 fo peak DTDT1cosMM f peak cos 24 EUEU 4 24 4 3
The total conduction losses ( Ptot_ con ) of three-phase inverter is given by
PPPtot_1 con6( T_2 av D_ av ) (3.54)
2V The peak modulation index (M) for SPWM can be written as M peak . By vdc substituting (3.52) and (3.53) into (3.54), and simplifying,
22 PAIAIBIVBIVtot_1 con peak 2 peak 1 peak peak cos 2peak peak cos (3.55) where,
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3(VV ) 3(rr ) 3(VV ) 4(rr ) A ceo fo , A ce f , B ceo fo and B ce f 1 2 1 2 4 vdc
3.3.2.3 Representation of Total Inverter Losses In Steady-state Synchronous dq Reference Frame
This section discusses the representation of inverter losses in terms of steady-state qd components of output currents and voltages. This representation of loss model will be used for the loss minimisation algorithm discussed in Chapter 8.
Total inverter loss ( Pinvtot ) can be obtained by summing (3.46) and (3.55) as follows:
22 PAIAIBIVinvtot121 peak peak peak peak cos BIV2peak peak cos 6 f (3.56) sw Av Av I T DC D DC peak
ee ee ee By substituting I peakii ds qs and I peakVvivi peak cos (qs qs ds ds ) into (3.56), following relationship for total inverter losses in terms of dq axes voltages and currents can be obtained.
6 ' 'ee 22 ee 22 eeee PAEEfiiAiiBviviinvtot ((12T D )) sw qs ds (qs ds )(1qs qs ds ds ) (3.57) ee ee e22 e Bvi2 ()qs qs vi ds ds i qs i ds where,
' ' ETTDCAv and ED AvDDC can be considered as constant since the steady-state DC bus voltage is constant in the ISA.
3.4 Battery Modelling
Battery is an electrochemical device that stores electrical energy in form of chemical energy when charging. The stored chemical energy is converted to electrical energy when discharging through a load. For example, in lead-acid battery following reversible chemical reactions take place at positive and negative electrodes when charging and discharging the battery [49].
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discharging PbO (s) + 3H+-- + HSO +2e 2PbSO (s) + 2H O 24charging 42
discharging Pb(s) + HSO-+ PbSO (s) + H +2e- 44charging
The modelling of the battery can be very complicated depending on the approach. A sophisticated chemical model that takes into account the electric potential, electrolyte concentration, diffusion, active surface area, exchange current density and etc may predict accurate characteristics of the battery [50]. However, this will be a cumbersome approach for a model used in an application such as ISA. Given the complexity of the physical characteristics of battery, it is difficult to have a single model that describes battery accurately for different applications. For the ISA application, battery model only need to capture the dynamic behaviour and thus, performance based equivalent circuit model would serve sufficiently. Simplest battery model is a fixed voltage source in series with a resistance which represent the internal resistance of the battery. This model is not adequate for ISA application and suitable only for cases where dynamic behaviour is not required.
In this thesis, a RC model that capture dynamic behaviour relatively more accurate is used. Figure 3.11 shows the equivalent circuit of the RC battery model [51] [52].
Re Rt Is
Rc
Vo Cb VCb Cc VCc
Figure 3.11 RC Battery Model
The capacitor Cb is very large and represents ample capacity of the battery to store charge chemically. The capacitor Cc is small and mostly represents the surface effects
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of a spiral-wound cell, for example, the limiting behaviour of a battery to deliver current based on time constants associated with the diffusion of materials and chemical reactions. The resistances Rt , Rc and Re represents terminal resistance, surface resistance and end resistance respectively. The state space equations representing the model can be written as follows:
FVFV11 FVR dVCb c GWGWFVGW dt CRbe()() R c CR be R c VCb CRbe () R c GWGWGWGWI (3.58) dV GW11V GWR GWCc HXCc e HXGWGWGW dt HXCRce()() R c CR ce R c HXCRRbe() c
FVFVRRFVV RR VRGWGWceGWCb ceI (3.59) ot HXHXRRRRececHXVCc RR ec
The capacitance value of Cb can be determined from the open circuit voltage at 0% state of charge (SOC) , 100% SOC and ampere-hour of the battery. The energy stored in the capacitor, ECb is given by
11 EC Ampsec V C V222 C( V V ) (3.60) bS100%OC22bb 100%SOC 0%SOC
By rearranging (3.60),
Ampsec V C 100%SOC (3.61) b 1 ()VV22 2 100%SOC 0% SOC
Battery resistances (i.e. Rt , Rc , Re ) and the surface capacitance Cc are obtained by voltage profile of the battery during high current pulse discharge.
3.5 Internal Combustion Engine Modelling
This section briefly discusses the modelling of Internal Combustion Engine (ICE). The modelling of an internal combustion engine is complicated as it involves various physical processes and parameters and therefore, relies heavily on experimental data and empirical correlations. The ICE model discussed in this thesis is based on various assumptions and only limited to the cranking (i.e. starting) operation. Running engine is assumed to be infinite torque source (i.e. stiff speed source) during generation operation of the ISA, for simplicity. The engine cranking model is useful in determining the
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starting torque requirement and acceleration behaviour during cranking of the engine. Since cranking is a non-fired state of the engine, no combustion or heat transfer models are needed [53]. An overview of a single cylinder of four-stroke ICE is shown in Figure 3.12.
Figure 3.12 An overview of a cylinder of a four-stroke engine
In a four-stroke engine, there is a sequence of process under running condition namely, induction stroke, compression stroke, power stroke and exhaust stroke. Usual operation of each stroke is briefly described below [50].
Induction stroke: the inlet valve is opened and exhaust valve is closed during this stroke. The piston moves downward from top dead centre (TDC) to bottom dead centre (BDC). The air and fuel mix is allowed into the cylinder.
Compression stroke: the inlet and exhaust valves are closed during this stroke. The piston moves upward from BDC to TDC and compresses the air fuel mixture trapped in the cylinder.
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Power stroke: the inlet and exhaust valves are closed during this stroke. The air/fuel mixture is ignited when piston starts moving downwards from TDC. The high pressure creates by the combustion act on the piston and forced it down towards BDC which results the work.
Exhaust stroke: the inlet valve is closed and exhaust valve is opened during this stroke. The piston moves from BDC to TDC. The gases due to burnt fuel are forced out from the cylinder.
An IC engine needs to be cranked initially by an external means in order to start. Once the engine speed reaches to certain level, the fuel and ignition system activates and commences the combustion process. Even though the induction stroke, compression stoke and exhaust stroke during starting of the engine behave similar to the running condition, the power stoke does not create any additional work during starting as the ignition system is not activated. Therefore, the power stroke during starting can be called as expansion stroke. However, the trapped air/fuel mixer which is compressed during compression stroke, acts on the piston and releases the work during the expansion stroke. The torque component required to overcome the compression of the cylinders during starting is called cylinder pressure torque (or compression torque) and can be obtained considering variation in volume of air/fuel mixture trapped in the cylinder during compression and expansion strokes. The other torque component required to overcome during starting is the engine friction. Engine friction is caused by various components such as piston assembly, valve assembly, crankshaft bearing and oil pump. Modelling of these components is based on the empirical equations for simplicity.
Section 3.5.1 and Section 3.5.2 briefly discusses the modelling of cylinder pressure torque and frictional torque that need to be overcome by the ISA during the starting.
3.5.1 Cylinder Pressure Torque Modelling
From the geometry of crankshaft and connecting rod mechanism, an expression for the volume of the air/fuel mixture in the cylinder can be derived as a function of the crank angle as given below [54].
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CS2 2 LLCS VRA DT1cos DT sin2 (3.62) iiiDT EUCR 1 RREU where, i is the cranking angle in degrees, V ()i is the volume of i th cylinder at crank angle i , A is the piston area, R is the radius of crankshaft, L is the length of connecting rod and CR is the compression ratio. The minimum cylinder volume occurs when the piston is at TDC and can be obtained by substituting 0 to (3.62) as follows:
CS2 VRA 0 DT (3.63) EUCR 1
The maximum cylinder volume occurs when the piston is at BDC and can be obtained by substituting to (3.62) as follows:
CS2 VRA DT2 (3.64) EUCR 1
Assuming the compression and expansion process during starting is adiabatic (i.e. PV k ), the relationship between the cylinder pressure and crank angle can be obtained as follows:
CSV PP DT (3.65) EUV where, is the polytrophic index of the gas mixture
P is the cylinder pressure immediately before the beginning of the
compression stroke (i.e. end of the suction stroke). This pressure is equal to manifold pressure since the intake valve is opened in the suction stroke. In this
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thesis, this pressure is assumed to be equal to atmospheric pressure (i.e. 1atm) for simplicity.
The cylinder pressure and torque can be calculated by plugging (3.62) into (3.65). As may be seen, the cylinder pressure and torque depend on the geometrical properties of the engine and physical properties of the air/fuel gas mixture.
5 x 10 18
16
14
12
10
8 Pressure (Pa) Pressure 6
4
2
0 0 5 10 15 20 25 30 Angle (rad)
Figure 3.13 Pressure Vs cranking angle for a single cylinder
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150
100
50
0 Torque (Nm) Torque -50
-100
-150 0 5 10 15 20 25 30 Angle (rad)
Figure 3.14 Cylinder pressure torque vs. cranking angle for a single cylinder without ignition
Figure 3.13 and Figure 3.14 show simulated cylinder pressure and cylinder pressure torque per cylinder of a 1753cm3 four cylinder IC engine. As may be seen in Figure 3.14, the positive and negative portions of the cylinder pressure torque is symmetrical as there is no net torque production without ignition of the engine during cranking. However, there is another important torque component that needs to be overcome during the cranking. That is the frictional torque component. It should be noted that Figure 3.14 shows only the cylinder pressure component. The engine friction torque modelling will be discussed in Section 3.5.2
The torque contribution of each cylinder can be obtained by using following cranking angle relationship:
720 (1)i (3.66) i ncyl where, i 1,2,3 … corresponds to each cylinder ncyl is the number of cylinders.
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For four cylinders ICE, the pistons are connected to the crankshaft with 180º phase shift to the adjacent piston as given by (3.66) in order to distribute peaks of the torque.
3.5.2 Engine Friction Torque Modelling
Various moving components in the engine contribute to the friction. The main contributors are the piston assembly, valve assembly, crankshaft bearing and oil pump. Among them, largest portion of engine friction is caused by piston ring assembly and valve assembly [55]. Engine friction caused by all these components can be modelled to result in either mean or instantaneous values. The mean value approach averages the friction over each cycle where as instantaneous approach calculates the friction as function of angle. The mean value approach provide relatively less complicated model [56] and therefore, the mean frictional torque is used for the friction model describe in this thesis. Empirical formulas for each component of the friction are obtained in terms of frictional mean effective pressure (fmep).The friction model utilised in this thesis is based on empirical formulas of a study presented in [57]. These frictional components are function of engine speed, lubricant viscosity and geometry. The total frictional mean effective pressure is calculated by summing the components. The engine friction depends on the ambient temperature since the oil viscosity is a function of temperature. At low temperature, the viscosity is high and thus the friction is higher. The variation of viscosity due to the temperature can be described by the Vogel’s equation [58, 59] and this equation is used in this thesis to model friction of engine under cold cranking condition.
A cranking model for a 4-cylinder 1753cm3 including above discussed cylinder pressure torque, frictional torque and moment of inertia is developed using MATLAB/Simulink as a part of this thesis. This model is used for cranking simulation for the ISA presented in Chapter 5. Detailed description of modelling equations will not be discussed in this section as it is out of main focus of this thesis. However, a printed copy of Simulink files and M-files of the model that contains modelling equations is attached in Appendix B
3.6 Summary
This chapter discussed the modelling of subsystems of the ISA system. This includes modelling of induction machine, inverter, battery and internal combustion engine.
67 Chapter3: Integrated Starter Alternator System Modelling
Induction machine is modelled for steady-state and dynamic conditions. The dynamic model is dealt with two axis synchronous and stator reference frames. The inverter is modelled for dynamics relationship between AC- and DC-side. The losses of the inverter are treated in a separate model that represents inverter losses in terms of steady- state qd axis variables. This model accounts for both switching and conduction losses of active switches and diodes of the inverter. The battery is modelled with resistive and capacitive (RC) components. This model includes RC components that represent main storage, surface charges, terminal resistance, end resistance surface resistance. The engine is modelled for cranking operation which account for both cylinder pressure torque and frictional torque. The cylinder pressure component is assumed to be adiabatic during expansion. The frictional torque is based on empirical formals which represent friction of piston assembly, valve assembly, crankshaft bearing and oil pump.
The modelling discussed in this chapter is used for designing the controllers, minimising the losses to obtain optimum operating points and for various computer simulations presented throughout this thesis.
68 Chapter4: Parameter Variations of Induction Machine ISA
CHAPTER 4
PARAMETER VARIATIONS OF INDUCTION MACHINE ISA
4.1 Overview
This chapter presents a detailed investigation on parameter variations of induction machine ISA and determination of these parameter variations. As discussed in the previous chapter, the induction machine can be described using a model with several fixed parameters, namely, stator resistance stator leakage inductance, magnetising inductance, rotor leakage inductance and rotor resistance. In addition, the core loss is conventionally represented as a resistance across the magnetising branch of the equivalent circuit model. These parameters are usually determined using conventional no-load and locked-rotor tests. The conventional no-load test is typically carried out at rated voltage and frequency. The locked-rotor (stand-still) test is usually carried out at rated current and rated frequency. The parameters obtained from these tests are constants for a given motor. However, induction machine parameters show significant variation against changing operating conditions such as variations in stator frequency, slip frequency, magnetisation and temperature [60-62]. These parameter variations should be carefully taken into consideration in modelling ISA system, because the wide speed range causes the induction machine to be operated over wide stator frequency range and under variable magnetisation in field weakening region. In addition, the ISA is subjected to highly variable loading conditions in generation mode due to switching- on and -off of various on-board loads. This causes highly variable rotor slip frequency. All major parameter variation determined in this chapter is incorporated in the computer model used for simulation studies so that the computer model closely represents the actual ISA induction machine. With this model, the behaviour of ISA can be predicted more accurately compared to conventional constant parameter model. The major parameter variations and the causes for those variations are described below.
69 Chapter4: Parameter Variations of Induction Machine ISA
Magnetising inductance: magnetising inductance varies with magnetising current and the frequency due to the nonlinear relationship between flux and magnetising current (i.e. magnetic saturation). Induction machine ISA operates in wide speed range which results in varying magnetising current and stator frequency. This causes varying magnetising inductance.
Rotor leakage inductance and rotor resistance: rotor parameters of the induction machine vary due to rotor current displacement or skin effect [63]. The frequency of the rotor current increases with the slip and becomes equal to stator frequency at unity slip. At lower frequencies, the current density in the cross section of a rotor bar is some what uniform whereas at higher frequencies the current in a rotor bar concentrates more close to the airgap end. The higher current density close to the airgap increases the possibility of the flux produced by rotor current crossing the airgap and linking the stator core to become part of the mutual flux and hence reduces the rotor leakage flux. Consequently, rotor leakage inductance decrease with the increase of frequency of the rotor current (i.e. with the increase of slip). The higher concentration of current in a small area of the rotor bar results in increased rotor resistance at higher slip frequency.
Core losses: core loss of the induction machine is the sum of eddy current loss and hysteresis losses in the stator and rotor. The eddy current loss in the stator is 2 proportional to the air-gap flux ( a ) and square of the stator frequency ( fe ). The n hysteresis loss in the stator is proportional to fe and a . n is an empirical coefficient typically varying between 1.5 and 2.5 depending on the lamination properties of the motor. Rotor core loss also consists of eddy current and hysteresis components. Rotor 22 eddy current loss is proportional to sfe and a . Rotor hysteresis loss is proportional to n sfe and a [64]. s is the slip of the induction machine. The slip of the induction depends on the load and typically varies between 0 and 0.15 at the steady-state operation. Consequently, the rotor core losses are significantly small compared to the stator core losses and hence, can be neglected. The overall core loss of the motor varies with the stator frequency and air-gap flux of the induction machine ISA.
Conventionally, the magnetising inductance and core losses are calculated from the no- load test at rated voltage and frequency. Thus, the parameters calculated from this test are true only when machine is applied with rated voltage and rated frequency. Constant
70 Chapter4: Parameter Variations of Induction Machine ISA
values for rotor resistance and rotor leakage inductance are conventionally calculated from the locked-rotor test at rated frequency and rated current. The frequency induced in the rotor during this test is equal to the rated stator frequency (unity slip, s 1) and thus the calculated parameter values are true only at rated stator frequency of the machine. However, as discussed earlier in this chapter, the induction machine in ISA application operates in a wide frequency range, and under variable air-gap flux in field wakening region. In addition, the induction machine ISA is subjected to variable rotor frequency due to variable loading level in generation mode.
This chapter investigates the parameter variations of the induction machine based ISA for changing stator frequency, slip frequency and magnetisation. The rest of this chapter is organised as follows: Section 4.2 discusses representation of variable parameters equivalent circuit for the induction machine ISA. The variable parameters determination method and test result are presented in detail in Section 4.3. Section 4.4 discusses the incorporation of parameter variation into a computer model that is to be used for various simulation studies presented in this thesis.
4.2 Variable Parameter Equivalent Circuit
The conventional steady-state equivalent circuit of an induction machine discussed in Chapter 3 consists of constant parameters for stator resistance, stator leakage inductance, magnetising inductance, rotor leakage inductance and rotor resistance. The core loss of the induction machine is usually represented as a constant resistance across the magnetising inductance. As discussed in the previous section, the major parameter variations are represented as follows:
' ' The rotor resistance ( Rr ) and rotor leakage inductance ( Lrl ) are represented as functions of frequency of rotor current (i.e. slip frequency, f2 ),
'' RrrRf()2 (4.1)
'' LLfrl rl ()2 (4.2)
The magnetising inductance ( Lm ) can be given as a function of the magnetising current
( Im ) and stator frequency ( fe ),
71 Chapter4: Parameter Variations of Induction Machine ISA
LLIfmmme(,) (4.3)
The core losses ( Pfe ) is represented as a function of magnetising current and stator frequency,
PPIffefeme(,) (4.4)
The steady-state induction machine equivalent circuit model that takes into account these major parameter variations is shown in Figure 4.1.
' R Lsl Lrl ()f2 ' Is s Ir
Em Im
V Em ' s R Rr ()f2 m PI(,) f L (,)If feme mme s
Figure 4.1 The steady-state variable parameter equivalent circuit model for induction machine
The current through the core-loss resistance ( Rm ) in the equivalent circuit is significantly smaller compared to the magnetising current ( Im ). The equivalent impedance Zeq for the steady-state equivalent circuit can be written as follows:
R' jLjL()r ' V em erl ZRjLs s (4.5) eq I s e sl R' s r j ()LL' s erlm
The power into the stator the induction machine ( Ps ) can be expressed as follows:
R' PIRI332'2r P (4.6) s srs fe
Next section discusses the determination of variable parameters of the induction machine ISA
72 Chapter4: Parameter Variations of Induction Machine ISA
4.3 Determination of Parameter Variations
This section presents determination of parameter variations of induction machine ISA. In order to obtain the variation of the parameters caused by the change in stator frequency, magnetisation and slip frequency, the induction machine is subjected to stand-still and no-load tests under variable frequency and variable voltage. In addition to the above tests, impressed stator current test [65] is carried out for the induction machine ISA in order to verify rotor resistance value obtained from the variable- frequency stand-still test at near zero frequency (i.e. approximate DC value).
Sections 4.3.1 and 4.3.2 describe the variable-frequency stand-still test and variable- frequency variable-voltage no-load test, respectively. The constant parameters of the induction machine ISA obtained form conventional 50Hz no-load and stand-still tests are listed in Table 4.1.
Table 4.1 Constant parameters obtained from conventional stand-still and no-load tests.
4.3.1 Variable-frequency Stand-still Test
In view of investigating the variations of rotor resistance and rotor leakage inductance with slip frequency, variable-frequency stand-still test was conducted. An experimental set-up of variable-frequency stand-still test is shown in Figure 4.2. As can be seen in this figure, variable-voltages and variable-frequencies are generated using a three-phase PWM inverter by adjusting the magnitude and frequency of sinusoidal reference of the pulse width modulator. The stand-still test was carried out at rated current for frequency
73 Chapter4: Parameter Variations of Induction Machine ISA
range form very low to rated stator frequency. Line-voltage, line-current and three- phase power of the induction motor were measured using a Power Analyser.
ISA Rectifier Inverter Power Analyzer Induction Machine
3-Phase Supply PZ 4000
Locked rotor
PWM
Vref f ref
Figure 4.2 An experimental set-up of variable-frequency stand-still test.
The slip frequency of the induction machine is equal to the stator frequency (i.e. 2 e ) at stand-still. It can be assumed that magnetising reactance is very large
R' compared to rotor impedance at stand-still (i.e. r jLjL' ). Thus, equation s erl em (4.5) can be simplified to following at stand-still.
V LR '''' Zeq ()()()()RRsr j erlslsr L L RR j2 L rlsl L (4.7) 3ILR
where VLR and I LR are respectively line voltage and line current during stand-still test.
The core losses during the stand-still test can be neglected since significantly lower voltage is applied to the machine compared to the rated voltage. The total active power is assumed to be dissipated as copper loss and hence the power at stand-still test ( PLR ) can be expressed as below.
2' PIRRLR3( LR s r ) (4.8)
74 Chapter4: Parameter Variations of Induction Machine ISA
The rotor resistance and rotor leakage inductance at each slip frequency can be obtained by solving (4.7) and (4.8) using measured voltage, current and power, provided that the stator resistance and stator leakage inductance are known. The stator resistance ( Rs ) is determined using DC resistance measurement. The stator leakage inductance is calculated by assuming that the stator and rotor leakage inductance are equal to at rated frequency lock-rotor conditions. The stator leakage inductance is then assumed to be constant for all frequencies.
The variable-frequency stand-still test was conducted over frequency range of 4 - 50Hz. The rotor resistance and rotor leakage inductance variations with slip frequency were obtained from the results of this test. Figure 4.3 shows the variation of rotor resistance and referred value of rotor resistance against slip frequency (referred rotor resistance,
2 CSL ref m ' RrrDTR ). Figure 4.4 shows the variation of rotor leakage inductance with slip EULr frequency. As can be seen from these figures, the rotor parameters significantly vary with slip frequency, and the conventional stand-still test at rated frequency does not provide accurate parameters for operational slip frequencies. The lowest rotor resistance occurs when the slip frequency is close to zero and increases with the slip frequency. The rotor resistance at zero slip is 2.32 m$ which is 75% of the value at 50Hz. The rotor resistance value at 50Hz obtained by variable-frequency stand-still test is about 3.16 m$ as shown in Figure 4.3.
The rotor leakage inductance, as shown in Figure 4.4, decreases as slip frequency increases and has significantly higher value at low slip frequencies compared to the 50Hz value. As discussed earlier, the current in a rotor bar concentrates more close to the airgap end at higher frequencies. The higher current density close to the airgap increases the possibility of the flux produced by the rotor current crossing the airgap and linking the stator core to become part of the mutual flux and hence reduces the rotor leakage flux. As a result, rotor leakage inductance decreases with the increase of frequency of the rotor current (i.e. with the increase of slip frequency).
75 Chapter4: Parameter Variations of Induction Machine ISA
Figure 4.3 Variation of rotor resistance and referred value of rotor resistance with slip frequency.
Figure 4.4 Rotor leakage inductance variation with slip frequency.
76 Chapter4: Parameter Variations of Induction Machine ISA
4.3.2 Variable-frequency Variable-voltage No-load Test
In order to investigate the variations of the magnetising inductance with stator frequency and magnetising current, a no-load test was carried out under variable- frequency and variable-voltage conditions. A block diagram of the variable-frequency variable-voltage no-load test is illustrated in Figure 4.5.
2 syn p
V f ref ref
Figure 4.5 An block diagram of variable-frequency variable voltage no-load test.
As can be seen in Figure 4.5, the variable-voltages and variable-frequencies for induction machine are generated using a PWM inverter by adjusting the magnitude and frequency of the sinusoidal reference to the pulse width modulator. The induction machine ISA is driven by a speed controlled motor drive during this test. The speed command for the controlled motor drive is derived from the applied stator frequency as indicated in Figure 4.5 to ensure that the induction machine rotates at synchronous speed corresponding to each test frequency. The line-current, line voltage and three- phase power of the induction machine were measured at various frequencies and various stator voltages using a Power Analyser.
77 Chapter4: Parameter Variations of Induction Machine ISA
The slip of the induction motor is zero under no-load condition (i.e. s 0 ), since the induction machine rotates at synchronous speed. Equation (4.5) can be simplified for no-load condition as follows:
V NL ZeqRjLL s e() sl m (4.9) 3I NL where
VNL is the stator line voltage during no-load test and
I NL is the stator line current during no-load test.
The core losses, as described in Section 4.1 and 4.2, are also function of magnetising current and stator frequency, and can be obtained by applying no-load condition (i.e. s 0 ) to (4.6),
2 Pcl PIRNL3 NL s (4.10) where
PNL is the three-phase power into the stator during no-load test.
The variations of magnetising inductance and core losses of the induction machine are obtained from (4.9) and (4.10) for various magnetising currents and stator frequencies using the variable-frequency variable-voltage no-load test.
The variable-frequency variable-voltage no-load test is conducted on the prototype ISA over range of 5 - 270Hz with varying magnetising current. Figure 4.6 shows the variation of the magnetising inductance with the magnetising current at various stator frequencies. Magnetising inductance varies significantly with magnetising current, for example at 60Hz, the difference between maximum and rated magnetising inductance is about 17%.
Figure 4.7 shows the variation of core-loss with magnetising current for various values of stator frequency. As can be seen, the core-loss increases with the magnetising current as well as stator frequency.
78 Chapter4: Parameter Variations of Induction Machine ISA
Figure 4.6 Variation of magnetising inductance with magnetising current at various frequencies.
Figure 4.7 Stator core loss variation with magnetising current at various stator frequencies.
79 Chapter4: Parameter Variations of Induction Machine ISA
Next section discussed the impressed stator current test that was conducted in order to verify the zero frequency rotor resistance obtained by the stand-still test.
4.3.3 Impressed Stator Current Test
ref The stator resistance and the referred value of rotor resistance ( Rr ) can be determined by applying different level of DC stator currents in stepped manner to two phases of the induction machine and then analysing the resultant voltage transients [65]. A block diagram of the setup of this test is illustrated in Figure 4.8. A step stator current sequence suitable for this test and the resultant stator line voltage waveform are shown in Figure 4.9.
a
b
ia
1
i* a
Figure 4.8 A block diagram of impressed stator current test set-up.
80 Chapter4: Parameter Variations of Induction Machine ISA
s s iiqs s i qs tt3
is qs tt2 (a)
t1 t2 t3 t4 vs s 1 s qs i tt3 vvqs2 sL qs tt3 vs qs tt2 (b)
t t t 1 t2 3 4
vs qs tt3
Figure 4.9 Stator current (a) and stator voltage (b) during impressed stator current test.
The space-phasor definition for voltage and current can be written as follows:
2 vjvvtavtavtss(() () 2 ()) (4.11) qs ds3 as bs cs
2 ijiitaitaitss(() () 2 ()) (4.12) qs ds3 as bs cs where,
2 j ae 3 ,
vtas (), vtbs ( ) and vtcs ( ) are instantaneous phase voltages of phase a , b and c respectively and
itas (), itbs ( ) and itcs ( ) are instantaneous currents of phase a , b and c respectively.
The instantaneous voltages of phase a , b and c during this test can be written as follows:
81 Chapter4: Parameter Variations of Induction Machine ISA
1 1 vt() v, vt() v and vt ( ) 0 (4.13) sas2 Lsbs2 Lsc where,
vsL is the instantaneous line voltage between phase a and b .
The induction machine voltages can be represented as qd axis components in stationary reference frame by substituting the phase voltages given in (4.13) into (4.11) as follows:
1 vvs (4.14) qs2 sL
The instantaneous current during the impressed current test can be written as follows:
itsas() i, itsbs( ) i and itsc ( ) 0 (4.15)
By substituting phase currents given in (4.15) into (4.12),
s iiqs s (4.16)
By combining (3.1), (3.5) and (3.7) in stationary reference frame induction machine model given in Chapter 3, following relationship for stationary reference frame q axis current can be obtained.
L s ss' dd m s vRiLiqs s qs s qs qr (4.17) dt Lr dt
The induction machine remains stand-still during the test (i.e. r 0 ). By applying this condition to (3.3) and combining with (3.7) given in Chapter 3,
d TLs ssi (4.18) rqrqrmqsdt
Combining (4.17) and (4.18),
L s ref s ' d s m s vRRiLiqs() s r qs s qs qr (4.19) dt Lrr T where,
82 Chapter4: Parameter Variations of Induction Machine ISA
2 CSL ref ref m ' Rr is the referred rotor resistance given by RrrDTR , EULr
L ()LL ' r mrl Tr is rotor time constant given by Tr '' and RRrr
L2 ' ' m Ls is the stator transient inductance given by LLss(1 ) . LLs r
The stator current is controlled to be DC during periods (tt12), (tt23) and (tt34) as d indicated in Figure 4.9. Therefore, the rate of change of is is zero (i.e. is 0 ) qs dt qs d during these periods. By plugging is 0 and combining (4.18) and (4.19), the dt qs s q axis stator voltage ( vqs ) can be represented as follows:
d TvvRis ss (4.20) rqsqssqsdt
s As described by (4.20) and illustrated in Figure 4.9(b), vqs decays exponentially during time periods (tt12), (tt23) and (tt34). The decaying time constant is equal to rotor s time constant. Also, in the end of periods (tt12) and (tt23), the stator voltage (vqs ) reaches steady-state as shown in Figure 4.9(b). The stead-state relationship version of (4.20) can be written as follows:
s s vRiqs s qs (4.21)
Stator resistance is given by,
ss ()vv () qs t23 t qs t t Rs ss (4.22) ()ii () qs t23 t qs t t where,
s s s s ()v , ()v , ()i and ()i are as indicated in Figure 4.9. qs t2 t qs t3 t qs t2 t qs t3 t
83 Chapter4: Parameter Variations of Induction Machine ISA
s The value of voltage vqs in the beginning of period (tt34) (i.e. tt3 t ) can be obtained form (4.19) as follows:
srefss Lm vRRiqs ()s r qs qr (4.23) tt tt tt 333LTrr
The rotor flux at the end of period (tt23) (i.e. tt3 t) can be obtained from (4.18) as follows:
ss Li (4.24) qr m qs tt33 tt
s However, the rotor flux can not be changed suddenly. Therefore, the rotor flux qr immediately before and after tt3 (i.e. tt3 t and tt3 t ) is equal. This can be written as follows:
ss (4.25) qr qr tt33 tt
ref By combining (4.23), (4.24) and (4.25), the referred value of rotor resistance ( Rr ) can be obtained as below.
vRiss qs s qs ref tt33 tt Rr (4.26) iiss qs qs tt33 tt where,
vs , is and is are as indicated in Figure 4.9. qs qs qs tt3 tt3 tt3
Figure 4.10 shows the current applied to stator and the resultant stator voltage during impressed stator current test. The rotor time constant is obtained by exponential curve fitting for stator voltage transient during period T2. The referred value of rotor resistance obtained using the starting value of the stator voltage in decaying period T3 as discussed above.
84 Chapter4: Parameter Variations of Induction Machine ISA
Figure 4.10 Stator current and voltage during impressed stator current test.
The parameters obtained for the induction machine ISA are shown in Table 4.2. The referred rotor resistance value obtained from this test is 1.84 m$ . This value very closely agrees with the extrapolated 0Hz value (i.e. DC value) of the referred rotor resistance obtained from the variable-frequency stand-still test shown in Figure 4.3.
Table 4.2 Impressed stator current test results.
4.4 Variable Parameter MATLAB/Smiulink Model for IM ISA
The computer modelling and simulation studies presented in this thesis were conducted using Matlab/Simulink. This simulation software includes Power System Toolbox which provides dynamic model for induction machine simulations. However, this model is a constant parameter model and the parameter variations are not taken into account. As discussed earlier in this chapter, constant parameter models do not accurately describe the behaviour of induction machine ISA. A more accurate Matlab/simulink
85 Chapter4: Parameter Variations of Induction Machine ISA
model for induction machine ISA was obtained by incorporating parameter variations obtained in Section 4.3. The parameter variations are included into the stationary reference frame dynamic model as look-up tables as indicated in Figure 4.11. 2
2
' ' Lrl ()2 R () s s r 2 Rs Lls is ir
s s vs j r r
Pcoreloss(,) eI m LImm(, e )
e Im I e m Figure 4.11 Dynamic model including parameter variation obtained in section 4.3
86 Chapter4: Parameter Variations of Induction Machine ISA
4.5 Summary
This chapter discussed the parameter variations of induction machine ISA. The variations of rotor resistance, rotor leakage inductance, magnetising inductance and core losses are considered in this chapter. These parameter variations are obtained by conducting stand-still and no-load test at variable-frequency and variable-voltage conditions.
The rotor resistance of the induction machine increases with the slip frequency. Its value at very low slip frequency (i.e. 0Hz) is 75% of the value at 50Hz. The rotor resistance value at 50Hz obtained from the variable-frequency stand-still test agrees closely with the value obtained by the conventional stand-still test. In addition the referred rotor resistance value at very low slip frequency which was calculated from the variable- frequency tests is compared to the value obtained from the impressed stator current test. These values agree very closely with a difference of only about 2%. The rotor leakage inductance also shows significant variation with respect to the slip frequency. It decreases with the increase of slip frequency. The leakage inductance values at lower slip frequencies are greater than double the value at 50Hz.
The magnetising inductance shows significant variation with respect to the magnetising current and stator frequency. It has higher values at magnetising currents less than the rated value. For example, the magnetising inductance at 60Hz is maximised at about 37A rms magnetising current and this maximum magnetising inductance value is 17% higher than the value at rated magnetising current. The influence of the stator frequency on magnetizing inductance is significant only when magnetizing current is low. The induction machine ISA shows higher magnetising inductance for low stator frequencies at low magnetising currents. The core-losses of the induction machine ISA increases with increase of stator frequency as well as magnetisation.
Finally, the obtained parameter variations are incorporated into a Matlab/Simulink model that was used in the simulation studies presented throughout this thesis.
87 Chapter 5: Control of Proposed Integrated Starter Alternator
CHAPTER 5
CONTROL OF PROPOSED INTEGRATED STARTER ALTERNATOR
5.1 Overview
This chapter develops the overall control design of the proposed integrated starter alternator. A schematic block diagram of the proposed ISA is shown in Figure 5.1. As can be seen in this figure, the ISA system consists of induction machine, three-phase PWM inverter, DC bus capacitor and 36V battery. The onboard electric loads are powered by the DC bus. The controller produces the PWM signals for the inverter. Rotor flux oriented control (RFOC) is utilised for the control of the induction machine ISA. The rotor flux required for RFOC is estimated using current model in the starting mode (i.e. low speed) and voltage model at high speeds. Synchronous reference frame controllers are used for current control of the ISA. Space Vector Modulation (SVM) is applied for PWM signal generation. The nonlinearities of switching devices are compensated in order to reduce the distortions in current waveforms. The stator voltage feedback signals required for control algorithm are computed using reference signals to SV modulator and DC bus voltage.
In rotor flux oriented reference frame, q and d axis components of stator current correspond to torque and flux respectively of the induction machine. The Starting Torque Profile Block (STB), shown in Figure 5.1, is a lookup function that provides motoring torque reference for required cranking acceleration characteristics. The Torque Estimation Block (TEB) calculates q axis current reference corresponds to the torque command for power assist or regenerative braking which is provided by the vehicle controller. DC Voltage Control Block (VCB) consists of DC bus voltage control algorithm which regulates DC bus voltage to 42V. The proposed DC voltage controller that provides tight voltage regulation will be discussed in detail in Chapter 6. The choice between these three algorithms for q axis current reference generation is made by the vehicle controller as indicated in the figure. The detailed operation of the vehicle
88 Chapter 5: Control of Proposed Integrated Starter Alternator
controller is beyond the scope of this thesis and therefore will not be discussed in this thesis.
Induction Inverter/ Rectifier Machine 42V DC Bus
LOAD ICE & IM BATTERY
v r ias ibs DC
Space Vector vDC Modulator (SVM) + vas Stator Device Nonlinearity Voltage Compensation vbs Estimation
r i e as Rotor Flux e Estimation dr Synchronous ibs e + iqs Reference Frame vas Transformation ie Current Controllers vbs abs dqe ds
e e e* e* vqs vds iqs ids
e* Changeover iqs Command From Vehicle Controller I max max (Limit) qs Iqs Flux Reference max Iqs Block (FRB) Fig. 8.4 e1* iqs
e e Generation Motoring e v e* dr vqs ds e iqs r
ie1* Motoring / qs Generation
* DC Voltage Torque Starting Estimation vDC Control Torque (42V) Block (VCB) Block Profile Fig. 6.9 (TEB) Block (STB)
e e* e vDC i * r e dr qs T dr
Power Assist / Regen Braking Torque Reference From Vehicle Controller
Figure 5.1 Schematic block diagram of proposed ISA.
89 Chapter 5: Control of Proposed Integrated Starter Alternator
Flux Reference Block (FRB) shown in the overview block diagram represents generation of flux reference. This includes flux reference generation for cranking operation, field weakening at high speeds and loss minimised control in generation mode. The proposed field weakening design will be discussed in detail in Chapter 7. The proposed loss minimised control will be discussed in detail in Chapter 8. Rest of this chapter discusses details of flux estimation, stator voltage computation, nonlinearity compensation of inverter switching devices, and current control design. In addition, results of cranking simulation study will also be presented in a later section of this chapter.
5.2 Rotor Flux Oriented Control (RFOC)
The rotor flux oriented control expresses the induction machine stator currents in a way that they can be manipulated to produce conditions similar to a DC machine, where the air gap flux is decoupled from the torque producing armature current (i.e. orthogonal). This allows induction machine to offer DC machine-like performance instead of sluggish response provided by scalar control. Rotor flux oriented control (RFOC) technique is utilised for the ISA controller to achieve rapid starting and tight DC voltage regulation.
The rotor flux oriented control requires aligning d axis of synchronous reference frame to the rotor flux. This reference frame is called Rotor Flux Oriented Synchronous Reference Frame (RFOSRF) in this thesis. Therefore, the conditions for rotor flux orientation can be written as follows:
e qr 0 (5.1)
By substituting (5.1) to (3.21), the electromagnetic torque can be obtained as follows:
L 3 P m ee ee TiKiem qs dr qs dr (5.2) 22Lr
By substituting condition in (5.1) to 4th row of matrix equation (3.24) and rearranging,
d TLeei e (5.3) rdrdrmdsdt
90 Chapter 5: Control of Proposed Integrated Starter Alternator
As may be seen in (5.2), the torque expression of induction machine in rotor flux e orientation is similar to that of the separately excited DC machine. The rotor flux ( dr ) e of the induction machine can be controlled via d axis current (ids ) as given in (5.3), e independent to the torque producing q axis current (iqs ).
5.2.1 Rotor Flux Estimation during Starting
This section discusses the estimation of rotor flux in starting mode (i.e. low speed) of ISA. The flux is estimated using stator current and speed feedback and is so called current model flux estimation which was originally formulated by Blaschke [41].
Rotor flux angle of the induction machine is obtained by substituting condition (5.1) to 3rd rows of matrix equation (3.24).
L m ie (5.4) er e qs Trdr
The rotor flux angle can be obtained by integrating (5.4). The magnitude of rotor flux is obtained from (5.3).
Rotor flux model described by theses equations can be graphically represented as in Figure 5.2. r p 2 e iqs L 1 m ias T s e abcs r e ibs i qde cs e e ids Lm dr sTr 1 e
Figure 5.2 Schematic diagram for flux estimation at starting (i.e. low speeds)
91 Chapter 5: Control of Proposed Integrated Starter Alternator
5.2.2 Rotor Flux Estimation at Higher Speeds
This section discusses the rotor flux estimation of the induction machine ISA in higher speeds. The current model flux estimation discussed in the previous section does not perform well in high speed field weakening region due to magnetising inductance variation which results in errors in flux estimation. In addition, the current model flux estimation depends on the rotor resistance which highly varies with the temperature and the skin effect of the rotor bars. The ISA utilises voltage model flux estimation in high speed operation. Voltage model flux estimation uses stator voltage and current to estimate the rotor flux of the induction machine [41]. This estimation is based on the stationary reference frame induction machine model discussed in Chapter 3. The stator flux in stationary reference frame can be obtained by integrating (3.1) and (3.2) as follows:
sss qsO ()vRidt qs s qs (5.5)
sss dsO ()vRidt ds s ds (5.6)
By manipulating (3.5), (3.6), (3.7) and (3.8), the rotor flux in stationary reference frame can be obtained as follows:
L s r ss' qr ()qsLi s qs (5.7) Lm
s Lr ss' dr ()dsLi s ds (5.8) Lm
By substituting (5.5) and (5.6) to (5.7) and (5.8) respectively, the rotor flux can be obtained as follows:
L LL' s r ssrs s qr O ()vRidtiqs s qs qs (5.9) LLmm
92 Chapter 5: Control of Proposed Integrated Starter Alternator
L LL' s r ssrs s dr O ()vRidtids s ds ds (5.10) LLmm
The ideal integration in (5.9) and (5.10) becomes difficult, since even slightest amount of DC offset of the voltage signals tends to build up at the output of the integrator. This problem can be solved by using a low pass filter (LPF) to replace the integrators present in (5.9) and (5.10) as follows [66, 67].
LT LL' s rL()vRissrs i s (5.11) qr qs s qs qs LTsmL(1) Lm
LT LL' s rL()vRissrs i s (5.12) dr ds s ds ds LTsmL(1) Lm
where,TL is the time constant of the low pass filter.
The rotor flux angle and magnitude can be calculated as follows:
CS s tan1 DTqr (5.13) e DT s EUdr
ess22 dr qr dr (5.14)
The rotor flux model described by these equations can be graphically represented as shown in Figure 5.3
s s vas v T L abcs qs L r qr v Ts1 L bs L m s s tan 1 qds T L e v vds L r dr cs TsL 1 Lm R R ' 1 ' s s s Ls Ls ias s iqs abcs 22 e ab dr ibs qds s ids ics
Figure 5.3 Schematic diagram for flux estimation in generation mode
93 Chapter 5: Control of Proposed Integrated Starter Alternator
Next section discusses the estimation of stator voltages which is required for voltage model rotor flux estimation as well as field weakening operation discuss in this thesis.
5.3 Computation of Stator Voltages
As may be seen in the Figure 5.3, stator voltage feedbacks are required for rotor flux estimation at higher speed operation of ISA. Also, stator voltages are required for the field weakening algorithm that will be discussed in Chapter 7. The required stator voltage signals are constructed (i.e. computed) using DC bus voltage and the reference signal to the Space Vector (SV) Modulator. There is an advantage in computation of voltages compared to measurement of voltages. The measured voltage signals need filtering to remove the switching frequency pulses. The delays caused by the filters may affect the accuracy of rotor flux estimation. In addition the extra hardware required for voltage measurement can be eliminated by computing the stator voltages.
V (110) V3(010) 2 e
a v * left V
V (011) V (100) 4 v 1 a n right vDC b c c b
g V (101) V5(001) 6 V (000) 0 V7(111)
Figure 5.4 (a) Three-phase inverter, (b) Space vectors of three-phase inverter.
Figure 5.4 show a three-phase inverter and corresponding space vectors. As can be seen in Figure 5.4 (b), a given voltage vector can be uniquely defined by tleft , tright and sector . where,
v v left right tTleft S and tTright S V1 V2
94 Chapter 5: Control of Proposed Integrated Starter Alternator
These three variables (i.e.tleft , tright and sector ) are generated from the reference signals to the SV Modulator. Switching pulse pattern for each sector is shown in Figure 5.5. SV modulation generates required voltage vector by turning-on the corresponding switches of end vectors of a sector for certain time durations. For example a voltage vector in sector-1 is generated using vectors V1 , V2 , V0 and V7 for time durations tright , tleft , t0 and t0 respectively in a symmetrical manner. Voltage vectors in other sectors are generated similarly utilizing left and right end vectors of the sector for given time durations.
V V V V V V V 0 1 2 7 2 1 0 T S t t t t t right left t t left right t 0 0 0 0 2 2 2 2 2 2 2 2
vag vag
v v bg bg
vcg vcg
Sector-1 Sector-2
vag vag
v v bg bg
vcg vcg
Sector-3 Sector-4
vag vag
v v bg bg
vcg vcg
Sector-5 Sector-6
Figure 5.5 Phase voltage pulse pattern (or switching pulse patterns) for six sectors
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The time period t0 given in Figure 5.5 is defined as follows:
1 tTtt() (5.15) 0 2 S left right
The instantaneous voltages of middle points of the legs of each phase relative to negative rail for a vector in sector 1 can be written as follows:
vvttag DC() left right t0 (5.16)
vvttbg DC() left 0 (5.17)
vvtcg DC 0 (5.18) where,
vag , vbg and vcg are mid point voltages of phase a , b and c relative to negative rail of the inverter (i.e. g ).
The instantaneous phase voltages with respect to star point of the induction motor
(i.e. n ) , van , vbn and vcn can be obtain by substituting (5.16), (5.17) and (5.18) to following equations.
1 vvvv(2 ) (5.19) an 3 ag bg cg
1 vvvv(2 ) (5.20) bn 3 bg ag cg
1 vvvv(2 ) (5.21) cn 3 cg ag bg
Phase voltages of induction machine corresponding to the other sectors can be obtained similarly using patterns given in Figure 5.5. These estimated phase voltages are used for
96 Chapter 5: Control of Proposed Integrated Starter Alternator
rotor flux estimation in higher speed operation; and field weakening operation to be discussed in Chapter 7.
The above computation of voltages using the reference signals to SV modulator assumes ideal operation of inverter. However, practical inverter output voltages can be significantly different from the required voltage by the reference signals applied to SV modulator due to nonlinearity of the inverter. The causes for the inverter nonlinearity and its compensation will be discussed in next Section.
5.4 Inverter Nonlinearity Compensation
This section discusses the inverter nonlinearity and its compensation. The nonlinearity of the inverter is mainly caused by switching dead-time, and non-idealities of active switches and anti-parallel diodes. The practical semiconductor switches, unlike ideal switches, have finite switching-on and switching-off times. The switching-on time is usually shorter compared to the switching-off time [68]. Therefore, in practical inverters, switching of turning-on device is delayed by few microseconds from the turning-off device of the same leg to prevent simultaneous conduction that causes shoot through fault (i.e. shorting of positive and negative of the inverter through upper and lower switches in a leg) . This time delay is called as dead time. Although the dead time is small (i.e. few micro seconds), it affects linearity of the inverter. In addition turn-on and turn-off time delays also contribute to the inverter nonlinearity [69].
Another factor that affects the linearity of the inverter is the conduction voltage drops of switches and anti-parallel diodes. The conduction drop of a switching device increases with the current passing through the device. The voltage drops in the switches cause output pulse voltage to be different from the voltage required by the reference. The conduction voltage drop of an IGBT can be described by a fixed component and a component that increases with the current passing through the device. In this thesis, the conduction drops of the switching devices and diodes are approximated to two piece characteristics as shown in Figure 5.6.
97 Chapter 5: Control of Proposed Integrated Starter Alternator
i ic f
rce rf
v v vceo vce fo f
Figure 5.6 Approximated forward characteristics of (a) IGBT (b) diode
The conduction voltage drop can be expressed as follows:
vvce ceo ri ce c (5.22)
vvrif fo f f (5.23) where, vce and v f are total voltage drop of IGBT and diode respectively vceo and v fo are saturation voltages of IGBT and diode respectively rce and rf are dynamic resistance of IGBT and diode respectively
ic and i f are magnitude of the current through IGBT and diode respectively
The effect of non-ideal nature of switching devices on output voltage of inverter depends on direction of the current. Figure 5.7 shows one leg of a three phase inverter conducting positive (i.e. current is out from the middle point) and negative current (i.e. current is into the middle point).
98 Chapter 5: Control of Proposed Integrated Starter Alternator
a ia 0 a ia 0 vDC vDC
g g
Figure 5.7 One leg of a three phase inverter with (a) positive current (b) negative current
The effect of current direction on the inverter output voltage pulse can be described as follows:
When ia 0 and T1 is ON (T2 is OFF): T1 is conducting and pulse voltage is dropped by conduction voltage drop of the switch (vce ).
When ia 0 and T2 is ON (T1 is OFF): D2 is conducting and pulse voltage is dropped by conduction voltage drop of the diode (v f ).
When ia 0 and T1 is ON (T2 is OFF): D1 is conducting and pulse voltage is increased by conduction voltage drop of the diode (v f ).
When ia 0 and T2 is ON (T1 is OFF): T1 is conducting and pulse voltage is increased by conduction voltage drop of the switch (vce ).
Figure 5.8 illustrates the effects of the time delays and voltage drops on output voltage vector in sector-1. The time delays indicated in the figure are; td : dead time; toff : switch-off time; ton : switch-on time. The voltage vectors in other sectors can be represented in a similar manner.
99 Chapter 5: Control of Proposed Integrated Starter Alternator
T S t t t t t t t t 0 right left 0 0 left right 0 2 2 2 2 2 2 2 2
(0)ia v v v v f ce (0)ia f ce t off t off vag tton d tton (0)ib d
v (0)ib bg
(0)ic
vcg (0)ic
Figure 5.8 Voltage pulses correspond to voltage vector in sector-1 for a practical inverter compared with ideal inverter
As can be seen in Figure 5.8, the deviation between practical and ideal voltage pulses are changed with the current direction. The voltage difference (or error) for phase a in sector-1 can be written as follows:
If ia 0
1 vvtttvttttvttttttag_00 error& DC off d on f d on off ce right left d on off ' (5.24) TS
100 Chapter 5: Control of Proposed Integrated Starter Alternator
If ia 0
1 vvttttvtttvttttttag_0 error & ce off d on DC d on off f right left 0 off d on ' (5.25) TS where,
vag_ error is the voltage error due to non-linearity of the inverter with respect to negative rail
The voltage errors in phases b (vbg_ error ) and c (vcg_ error ) with respect to negative rail can be written similarly.
The voltage errors with respect star point for phases a (van_ error ), phases b (vbn_ error ), phases c (vcn_ error ) can be obtained as follows:
1 vvvv(2 ) (5.26) an____ error 3 ag error bg error cg error
1 vvvv(2 ) (5.27) bn____ error 3 bg error ag error cg error
1 vvvv(2 ) (5.28) cn____ error 3 cg error ag error bg error
The corresponding modulation error functions of phase a , b and c can be obtained as follows:
v v v an_ error bn_ error cn_ error ma , mb and mc (5.29) vDC vDC vDC 3 3 3
The non-linearity of the inverter can be compensated by subtracting the modulation error functions of phase a , b and c from the modulation signal as illustrated in Figure 5.9.
101 Chapter 5: Control of Proposed Integrated Starter Alternator
macomp_ Tleft ma abc v * left V mbcomp_ Tright m v b right
Tright m sector ccomp_ Tleft mc sector
ma mb mc
ia ib
ic v DC
Figure 5.9 Schematic block diagram for inverter nonlinearity compensation
5.5 Designing of Current Controllers
This section discusses designing of current controllers for the induction machine ISA. Synchronous reference frame current controllers are utilised to control the stator current in order to achieve desired dynamic performance. The control design is based on the dynamic equations of induction machine in rotor flux oriented reference frame (RFORF).
5.5.1 Current, Flux and Torque Dynamics in RFORF
The q axis current dynamics in rotor flux oriented references frame can be obtained by substituting (5.1) into 1st row of matrix equation (3.24) as follows:
L '''d ee e m ee Lis qs RiLi s qs e s ds r dr v qs (5.30) dt Lr
2 CSL ' m By substituting RRRssrDT to (5.30), EULr
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2 CSLL ''d ee mme e e Lis qs() RR s r DT iqs e Li s ds r dr v qs (5.31) dtEU Lrr L
By substituting (5.4) to (5.31) and simplifying,
L ''d eee em e LiRivs qs sqs qs e Li sds e dr (5.32) dt Lr
The d axis current dynamics in rotor flux oriented references frame can be obtained by substituting (5.1) into 2nd row of matrix equation (3.24) as follows:
LR' '''d eeeeemr Lis ds e LiRi s qs s ds 2 dr v ds (5.33) dt Lr
2 CSL ' m By substituting RRRssrDT to (5.33) and simplifying, EULr
d LiRiv''eee Li e (5.34) s dt ds s ds ds e s qs
Equations (5.32) and (5.34) describe the dynamics of q and d axes currents respectively. These current dynamics can be represented as transfer functions shown in Figure 5.10. In this figure, all the differential equations are represented in Laplace e e domain (i.e. functions of “s”). mqs and mds are q and d axes modulation functions,
e e e e and the relationship between mqs , mds and vqs , vds for SVM modulation can be written as follows:
ve ve e qs e ds mqs and mds (5.35) vDC vDC 3 3
e e The numerical range of modulation functions mqs , mds are between -1 to +1.
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e L m dr e Lr
e ve e mqs qs 1 iqs ' sLs Rs vDC L' 3 es ' esL e ve e mds ds 1 ids ' sLs Rs vDC
3
Figure 5.10 q and d axis current dynamics in RFORF
As can be seen in Figure 5.10, q axis current dynamic is coupled with d axis current and rotor flux. d axis current dynamic is coupled with the q axis current. Significance of the effect of cross-coupling and its cancellation of will be discussed in Section 5.5.2.
In addition, the rotor flux and torque dynamics can be added to the current dynamics illustrated in Figure 5.10. The rotor flux and torque dynamics are governed by equations (5.3) and (5.2) respectively. The current, rotor flux and torque dynamics in rotor flux oriented reference frame are illustrated in Figure 5.11.
104 Chapter 5: Control of Proposed Integrated Starter Alternator
3 Lp m Te 22Lr e L m dr e Lr
e e e e v iqs mqs qs 1 iqs ' sLRs s vDC L' 3 es ' esL e ve ie e mds ds 1 ds Lm dr ' sLRs s sTr 1 vDC
3
Figure 5.11 Current, rotor flux and torque dynamics in RFORF
5.5.2 Current Controllers
This section discusses designing of current controllers of the ISA system. Proper designing of current regulation is very important aspect of the ISA design as poor designing of current controllers could deteriorate the overall performance in both motoring and generation modes. The bandwidth of the current regulators should be high enough to ensure the required dynamic performance of ISA, especially; to develop maximum available torque quickly during starting operation. Synchronous reference frame PI current regulators are utilized in order to obtain required high dynamic performance. However, the transient response of synchronous PI controllers deteriorate as the synchronous frequency increases due to excitation frequency cross-coupling adding to machine model by the synchronous reference frame transformation [70]. This ' e was shown in Figure 5.10. As illustrated Figure 5.10, the cross-coupling terms esdsLi ' e and esqsLi interact with q and d axis current dynamics, respectively. The increasing significance of cross coupling effect with increasing speed in generation
105 Chapter 5: Control of Proposed Integrated Starter Alternator
mode of ISA is depicted in Figure 5.12. This figure also indicates the back EMF variation with the speed of the induction machine.
Figure 5.12 Cross-coupling and back EMF voltages variation with the speed in generation mode.
The cross-coupling causes deterioration of current control performance in higher speeds. This problem could be overcome by cancelling the cross-coupling terms as described below.
By rearranging q and d axis current dynamics described in (5.32) and (5.34),
d LiRiu' eee (5.36) s dt qs s qs qs
d LiRiu' eee (5.37) s dt ds s ds ds
L s ee' m e where, uvqs qs e Li s ds e dr and Lr
eee ' uLivds e s qs ds
106 Chapter 5: Control of Proposed Integrated Starter Alternator
Equations (5.36) and (5.37) describe the transfer function (TF) of q and d axes
e e currents and resultant voltages, uqs and uds respectively. q and d axis current dynamics and PI-regulators in closed loop are shown in Figure 5.13. The cancellation of coupling terms (i.e. decoupling) are indicated in this figure. Decoupling of back EMF component is also shown in this figure, even though it is not very significant since the current dynamics are much faster than the dynamics of the back EMF.
L L m e m e edr edrL e* Lr e e r e e e* e iqs K mqs vqs 1 iqs iqs iqs K 1 iqs K Ic K Ic Pc ' Pc ' s sLRs s s sLRs s vdc vdc ' e 3 3 ' e esdsLi esdsLi
e* e e e e e* e ids K mds vds 1 i ids ids K 1 i K Ic ds K Ic ds Pc ' Pc ' s sLRs s s sLRs s vdc vdc ' e 3 3 ' e esqsLi esqsLi
Figure 5.13 Closed qd axes current loops with decoupling.
The closed-loop decoupled transfer function for q and d axis current can be written as follows:
ie isKKe qs ds Pc Ic (5.38) ee**2' iisLsRKKqs ds s() s Pc Ic
The closed loop current response holds second order characteristics as given in (5.38). The gains of controller is evaluated by comparing (5.38) with the optimum coefficients of ITAE (“Integral of Time multiplied by Absolute magnitude of the Error”) criterion for a ramp input for second order system [71]. PI gains derived according to ITAE criterion are given below.
' KLRPcn3.2 css (5.39)
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2' KLIc nc s (5.40)
The dynamic response of the closed q and d axes current loops depend on the natural frequency ( nc ) of the loops, and hence the value of nc is chosen for the desired dynamic response.
e e As mentioned earlier the numerical range of modulation functions mqs , mds are between -1 to +1. The modulation function outputs of the PI current controllers are limited so
ee22 that mmqs ds is always less than unity. Figure 5.14 shows implementation of this limit using polar coordinates.
e e mqs r mqs x, y r,
me me ds r, x, y ds
Figure 5.14 Implementation of limits to the modulation function
5.6 Modelling and Experimental Results for Current Control Design
The simulation tests were conducted to investigate the performance of the current controllers. Figure 5.15 shows the simulation results for the currents of the induction machine ISA during 120A load application at 2000rev/min in generation mode. As can be seen both q and d axis currents are closely following the current reference. Figure 5.16 shows the transients in current when load of 120A is dumped at 2000rev/min. Figure 5.17 demonstrates the coupling effect of the current control discussed in Section 5.5.2. This figure corresponds to the load dump of 100A at 3500rev/min in generation mode of ISA. Improvement to the current control results in by decoupling can be clearly seen in Figure 5.17.
108 Chapter 5: Control of Proposed Integrated Starter Alternator
500 (A) c
, I 0 b , I a I -500
200 Feedback 0 Reference (A) e qs I -200
-400 200
150 (A) e ds
I 100
50 0 0.02 0.04 0.06 0.08 0.1 Time (sec)
Figure 5.15 Phase currents, q and d axis current references and feedbacks in generation mode 120A DC load application at 2000rev/min
500 (A) c
, I 0 b , I a I -500
200 Feedback 0 Reference (A) e qs I -200
-400
100
50 (A) e ds
I 0
-50 0 0.02 0.04 0.06 0.08 0.1 Time (sec)
Figure 5.16 Phase currents, q and d axis current references and feedbacks in generation mode 120A DC load dump at 2000rev/min
109 Chapter 5: Control of Proposed Integrated Starter Alternator
Without Decoupling 60
40
20 d-current (A) 0 Feedback Reference With Decoupling 60
40
20 d-current (A) 0 0 0.05 0.1 0.15 0.2 Time (sec)
Figure 5.17 d axes current references and feedbacks in generation mode for 100A load dump at 3500rev/min (i) without decoupling (ii) with decoupling
500 (A) c
, I 0 b , I a I -500
200 Feedback 0 Reference (A) e qs I -200
-400 200
150 (A) e ds
I 100
50 0 0.02 0.04 0.06 0.08 0.1 Time (sec)
Figure 5.18 Experimental results for phase currents, q and d current ref and feedbacks in generation mode 120A DC load application at 2000rev/min
110 Chapter 5: Control of Proposed Integrated Starter Alternator
500 (A) c
, I 0 b , I a I -500
200 Reference 0 Feedback (A) e qs I -200
-400 100
50 (A) e ds
I 0
-50 0 0.02 0.04 0.06 0.08 0.1 Time (sec)
Figure 5.19 Experimental results for phase currents, q and d axis current references and feedbacks in generation mode 120A DC load dump at 2000rev/min
Figure 5.18 and Figure 5.19 shows experimental results for the prototype ISA for 120A load application and dump at 2000re/min. These experimental results show good agreement with the modelling results shown in Figure 5.15 and Figure 5.16.
5.7 Cranking Simulation The ISA system including the engine is modelled and simulated in order to investigate cranking performance of the ISA. The engine modelling discussed in Chapter 3 was used for these simulations. Figure 5.20 shows the cold cranking (-20°C) torque requirement of the engine and ISA starting torque profile up to 700 rev/min. The ISA produces significantly higher starting torque to ensure cranking of the engine under cold conditions. Figure 5.21 shows the variation of torque command, battery current and q axis current during cranking. The starting torque command is reduced with increasing speed in order to reduce the discharging current of the battery during starting (i.e. cranking current).
111 Chapter 5: Control of Proposed Integrated Starter Alternator
160
140 ISA torque 120
100
80
60 Torque (Nm)
40
20 Cold cranking(-20oC) Specification 0 100 200 300 400 500 600 700 Speed (rev/min)
Figure 5.20 Cold cranking (-20°C) specification and ISA starting torque.
150 100
T (Nm) 50 0
300
200 (A)
dc 100 I
0 600
400 (A) q I 200
0 100 200 300 400 500 600 700 Speed(rev/min)
Figure 5.21 Torque, battery current and q axis current during cranking.
112 Chapter 5: Control of Proposed Integrated Starter Alternator
Figure 5.22 shows the starting characteristics of the ISA under cold cranking condition (-20°C). As can be seen, the starting torque command is issued at about t = 200ms and ISA cranks the engine. The engine reaches 400rev/min in about 300ms under cold conditions, which is sufficient for the starting of the engine.
Figure 5.23 shows the speed and battery discharging current during cranking for three different operating conditions; (i) 20°C at 36V (ii) -20°C at 36V and (ii) -20°C at 18V. The speed of the engine reaches 400rev/min in about 350ms when cranking at room temperature (20°C) with 36V battery voltage. In the case of cold cranking (-20°C), it takes about 500ms to reach 400rev/min. Cranking under cold condition with low battery voltage (18V), which is the most onerous among them, takes about 550ms to reach 400rev/min. It can be noted in the case of cranking under low battery voltage (18V), the speed of engine can not exceed 500rev/min due to the limited DC bus voltage. Also, it requires high battery discharging current compared to other cases since higher current is needed to deliver the same power at lower voltage.
113 Chapter 5: Control of Proposed Integrated Starter Alternator
1000
500 (rpm) r N 0 200
100 T(Nm) 0 300 200 (A) dc
I 100 0 400 (A) s
I 200
0 20 (V)
LL 10 V 0
500
(A) 250 q I 0 0 0.2 0.4 0.6 0.8 1 1.2 Time (sec)
Figure 5.22 Cold cranking performance (i) speed, (ii) torque, (iii)battery current, (iv) stator current, (v) stator voltage and (vi) q axis stator current
114 Chapter 5: Control of Proposed Integrated Starter Alternator
800 -20oC at 36V 600 20oC at 36V 400 -20oC at 18V 200
Speed (rev/min) 0 0 0.2 0.4 0.6 0.8 1
500 -20oC at 18V 20oC at 36V 400
(A) 300 -20oC at 36V dc I 200 100 0 0 0.2 0.4 0.6 0.8 1 Time (sec)
Figure 5.23 Speed and battery discharging current during cranking (i) 20°C at 36V (ii) -20°C at 36V and (ii) -20°C at 18V.
5.8 Summary
This chapter presented the overall control of the proposed integrated starter alternator. The rotor flux oriented control is applied for the control of the induction machine of ISA. The flux estimation at low speed is based on the current model whereas voltage model is utilised in high speeds. The voltage integrators of the voltage model flux estimation were replaced with low pass filters to prevent integration of DC offsets. The voltage required for flux estimation was constructed from the measured DC bus voltage and reference signals to SV modulator. The compensation of nonlinearity of the inverter caused by dead time switching on/off time and conduction drops of the switches was implemented.
Current control design was discussed in detail. Effect of coupling between q and d axis current loops and its increased significance in high speed region was highlighted. Decoupling control was utilised for achieving high dynamic performance in current
115 Chapter 5: Control of Proposed Integrated Starter Alternator
control. Simulation and experimental results for current control performance were presented in this chapter.
This chapter also presented cranking simulation results of the proposed ISA. The cranking simulation conducted at different ambient temperatures and state of charge of the battery.
116 Chapter 6: DC bus voltage controller
CHAPTER 6
DC BUS VOLTAGE CONTROLLER
6.1 Overview
As discussed in Chapter 1, tight DC voltage control is essential requirement in a future automotive power system in order to minimise the cost associated with power electronic devices to be used in various loads. Load disturbances and engine speed variations are the main causes for DC bus voltage transients that vary significantly away from its set point. The worst case scenario for DC voltage transient is sudden disconnection of full load that was being supplied by the ISA. The DC voltage regulator of the ISA should maintain its performance throughout a wide speed range. For example, the DC voltage controller gain values that give optimum performance at certain speed may not necessarily provide similar performance at a different speed. In addition, DC bus voltage tends to overshoot when operation mode of ISA transfer from the motoring to generation operation. A fast DC voltage regulator can help reducing the size of the DC bus capacitor. The main purpose of DC bus capacitor is to filter out the ripple in the DC current of the inverter. Nevertheless, it provides a valuable support for attenuating over voltage transients. Proper DC voltage controller design which ensures fast q axis current command minimises the additional support required by the DC capacitors to limit the over voltages and hence, the capacitor size can be minimised.
This chapter presents the proposed new sophisticated DC bus voltage control design to achieve superior voltage regulation that provides solutions to above mentioned problems. The proposed DC voltage controller is based on a linearised mathematical model of the ISA, and includes decoupling for speed and flux. In addition an anti- windup technique is also employed to prevent voltage overshoot during transition from motoring mode to generation mode.
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Chapter 6: DC bus voltage controller
Next section discusses the integration of inverter and induction machine model in rotor flux oriented reference frame to obtain the model required for proposed DC bus voltage regulator design. In Section 6.3, control designing of DC voltage controller with decoupling for rotor flux and stator frequency will be discussed with simulation results. This section also discusses voltage overshoot caused by the winding-up phenomenon and its mitigation. Section 6.4 discusses the sizing of the DC bus capacitor and its support on DC over voltage reduction. Section 6.5 presents experimental results for the prototype ISA with proposed DC voltage control.
6.2 DC Voltage Dynamics of ISA
This section discusses the DC voltage dynamics of ISA which describes the relationship between induction machine variables and DC bus voltage. The synchronous reference frame inverter model discussed in Chapter 3 is combined with the induction machine model in rotor flux oriented reference frame discussed in Chapter 5 in order to obtain the dynamic relationship between DC voltage and induction machine variables. It can be assumed that DC voltage dynamics are much slower compared to the current dynamics in the induction motor. This assumption allows simplifying induction machine model by ignoring current transients (i.e. by considering steady-state qd currents). Equations (5.32) and (5.34) can be simplified by ignoring current transients as follows:
L ee' e m e vRiLiqs s qs e s ds e dr (6.1) Lr
ee ' e vRiLids sds e sqs (6.2)
The dynamic relationship between AC- and DC-side of the inverter is described by (3.40) given in Chapter 3:
vdv 2 P 3 DC DC DC ee ee vvDC ()qsi qsv dsi ds (6.3) dt Rdc C dc C dc2 C dc
By substituting (6.1) and (6.2) into(6.3),
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Chapter 6: DC bus voltage controller
dvvL2 P 3 DC DC DC ee22 m ee vRDC ((s iqsi ds ) e dri qs ) (6.4) dt Rdc C dc C dc2 C dc Lr
At higher speeds, the voltage drop across the stator resistance is small compared to the back EMF voltage component. The generation mode of ISA usually occurs at high speeds. (i.e. 900rev/min – 6000rev/min). Thus, following can be written for generation mode.
L ee22 m ee Rs ()iiqs ds e dri qs (6.5) Lr
ee22 By ignoring Rs ()iiqs ds component of (6.4),
dv v2 P3 L DC DCDC e m ee viDC dr qs (6.6) dt Rdc C dc C dc2 C dc L r
Equation (6.6) hold a non-linear relationship between DC bus voltage ( vDC ) and e q axis current (iqs ). This equation required to be linearised in order to apply linear control theory. A linear equation can be obtained by multiplying (6.6) by 2 and rearranging as follows:
dv2 L DC 2 3m ee RCdc dc 22( vDC R dc PDC e dri qs ) (6.7) dt 2 Lr
Equation (6.7) describes the dynamics of DC voltage as a linear differential equation.
2 The variable vDC instead of vDC is taken as the output variable in the linearised dynamic equation. The inverter dynamics given in (6.7) can be combined with the induction machine dynamic model in rotor flux orientated reference frame given by (5.32), (5.34), (5.35), (5.3) and (5.2). Figure 6.1 shows graphical interpretation of combined dynamics of induction machine and inverter of ISA application.
119
Chapter 6: DC bus voltage controller
3 p Lm Te 22Lr e e L 3 L m dr dr m e e Lr 2 Lr
e ve e e mqs qs 1 iqs iqs ' sLs Rs v DC L' 3 es
' esL e ve e ie e mds ds 1 ids ds Lm dr ' sLs Rs sTr 1 vDC 3 2 vDC v 1 DC 2Rdc 3 sRdc C dc 2
PDC v DC
Figure 6.1 Combined Dynamic Model for Induction machine and Inverter
Next section discusses the designing of the voltage controller based on the combined dynamic model of induction machine and inverter in rotor flux oriented reference frame.
6.3 Proposed DC Voltage Controller
6.3.1 Controller Design
This section discusses the design of the DC voltage controller based on the linearised transfer function discussed in the previous section. The transfer function given in (6.7)
2 e shows that vDC instead of vDC , has a liner relationship with iqs . Therefore, closed-loop
2 control of vDC is more appropriate control variable for DC bus voltage regulator design. Figure 6.2 shows a block diagram of proposed DC bus voltage controller. As can be 2 ee seen the control variable is vDC . The decoupling term 32emdrqsLi L r is included to the output of the PI controller in order to remove the coupling of stator frequency and rotor flux in the DC voltage dynamics (i.e. decoupling). 120
Chapter 6: DC bus voltage controller
DC Voltage Control Block (VCB)
e*1 2* * iqs v vDC KIVd DC 2 + - U - KPVd (42V) s 2 vDC 3Lm 2 U 2Lr
e v e dr DC
2 Figure 6.2 vDC controller with stator frequency and rotor flux decoupled
Decoupling Closed pDC current loop e* e v2 K iqs iqs 2R DC 2* + K IVd - -+ dc vDC - PVd 1 s sRdc C dc 2
3 L 2 m e 3 Lm e edr vDC edr 2 Lr 2 Lr
pDC
2* K 2Rdc 2 + K IVd - v - PVd + vDC DC s sRdc C dc 2 2 vDC
Figure 6.3 Closed DC bus voltage control loop.
121
Chapter 6: DC bus voltage controller
Voltage Loop Bode Diagram Current Loop 10
0
-10
-20
-30 0
-45 Phase (deg) Magnitude (dB)
-90 0 1 2 3 4 10 10 10 10 10 Frequency (rad/sec)
Figure 6.4 Bode plot for closed current and DC voltage loops (current loop natural frequency ( nc ) 100Hz, closed voltage loop natural frequency ( nvd ) 15Hz)
2 Figure 6.3 depicts vDC controller in closed-loop. The inner q axis current loop is 2 much faster than outer vDC loop and therefore, the closed loop transfer function of 2 q axis current can be assumed unity gain when designing outer vDC control loop. ee The decoupling term 32emdrqsLi L r shown in Figure 6.3 removes the coupling of rotor flux and stator frequency on the DC bus voltage control loop. PDC is the load on the DC bus and it acts as disturbance to the control loop (i.e. load disturbance). The 2 transfer function for outer vDC closed loop can be written as follows:
vKR2 22sR K DC PVd dc dc IVd (6.8) 2* 2 vRCsRKsRKDCdcdcdcPVd(2 2) 2 dcIVd
The gains of PI controller can be obtained by applying ITAE criterion. The Proportional gain, KPVd and Integral gain, KIVd obtained by the ITAE criterion are as follows: 122
Chapter 6: DC bus voltage controller
(3.2 RC 2) 2 C nvd dc dc nvd dc KPVd and KIVd (6.9) 2Rdc 2
The interaction between inner and outer loop is avoided by choosing natural frequency of outer loop ( nvd ) is much less than natural frequency of the inner loop [71]. Figure 6.4 shows bode plot for inner closed current loop and outer voltage loop.
6.3.2 Simulation Results
Simulation tests were conducted in order to investigate the performance of the proposed DC voltage controller.
200 100 0 Speed (krev/min) 50
(V) 40 DC V 30 50 0
T(Nm) -50 0 -200 (A) e qs
I -400 20
(V) 15 m
V 10 0.8 0.6 (pu) r F 0.4
200
(A) 100 e ds I 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 2 Time(sec)
Figure 6.5 Load current, DC bus voltage, torque, q axis stator current, peak phase voltage, rotor flux and d axis stator current during application and dump of 133A load 123
Chapter 6: DC bus voltage controller
Figure 6.5 shows transients of the ISA during sudden load application and dumps. The results shown in this figure are for a load of 133A at 2000rev/min which corresponds to the maximum power output operating point of the ISA discussed in this thesis. As may be seen, the DC voltage quickly returns to 42V after short transient periods for both load application and dump. It can be noticed that the DC bus voltage transient stays below 50V during the load dump. As can be seen, q axis current and torque adjust quickly to regulate the DC bus voltage to the set value. This figure also indicates the transients in stator voltage, flux and d axis current. The above test was conducted without battery connected.
5
Speed 0 (krev/min) 43 42 (V)
DC 41 V 40 0 -20
T(Nm) -40 0
(A) -100 e qs I -200 20
(V) 15 m
V 10
1
(pu) 0.5 r F 0 200
(A) 100 e ds I 0 0 1 2 3 4 5 6 7 Time(sec)
Figure 6.6 Speed, DC bus voltage, torque, q axis stator current, peak phase voltage, rotor flux and d axis stator current during engine acceleration and deceleration.
124
Chapter 6: DC bus voltage controller
Figure 6.6 shows the DC voltage regulation and other transients of ISA during fast acceleration and deceleration of the engine. As may be seen, the speed of the engine accelerates from 800 rev/min to 5000rev/min within one second and decelerates from 5000rev/min to 800rev/min within one second during this simulation test. This test was also conducted without battery connected to the DC bus.
Figure 6.7 compares the DC bus voltage regulation during acceleration of engine speed from 800rev/min to 5000rev/min with and without decoupling the flux in DC voltage control loop. As can be seen, the flux decoupling provides superior performance in DC bus voltage regulation. A zoomed in view of Figure 6.7 is shown in Figure 6.8. As may be seen, q axis stator current reference can not change fast enough in the case without flux decoupling compared to the case with flux decoupled control.
5
3 Speed 1
(1000rev/min) 0
43 With Decoupling 42 (V)
DC 41 V Without Decoupling 40
-100
(A) -150 e qs I
-200 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time (sec)
Figure 6.7 Speed, DC bus voltage and q axis stator current during engine acceleration with and without rotor flux decoupling.
125
Chapter 6: DC bus voltage controller
42.2 With Decoupling 42
(V) 41.8 DC
V 41.6 Without Decoupling 41.4
-109 Without Decoupling -110 -111 -112 (A) e qs
I -113 -114 With Decoupling -115 0.6 0.7 0.8 0.9 1 1.1 1.2 Time (sec)
Figure 6.8 DC bus voltage and q axis stator current during engine acceleration with and without rotor flux decoupling, zoomed in view of Figure 6.7
6.3.3 Anti-windup Control Loop
Significant DC voltage overshoots were observed when ISA changeover from motoring to generation mode; and when regaining control of DC bus voltage after sort-duration high demand DC voltage unregulated operation. Following describe these two events.
(1) Changeover from motoring to generation: the changeover from motoring generation mode occurs when the engine speed is reached to a certain value that is suitable for starting the engine (this speed is 600rev/min for the prototype ISA discussed in this thesis). In addition, the changeover between motoring and generation modes may be required at higher speeds for power assist feature depending on the ISA configuration.
(2) Short-duration high demand DC voltage unregulated operation: this occurs when onboard power consumption exceeds the power capability of the ISA in generation mode for short duration which usually occurs at high speed operation. During this short period, the ISA produces its maximum power capability and the remaining power requirement is supplied by the battery. As a result of power shortage from the ISA, the
126
Chapter 6: DC bus voltage controller
DC bus voltage can not be regulated to set value (42V for the prototype ISA) and stays below the set value. When the temporary requirement for high power demand is over, the DC voltage control needs to be regained and charging of the battery is initiated.
DC bus over voltage transients were observed during the above described events due to the winding-up phenomenon of the integrator in DC voltage controller. As a result of integrator windup, the q axis current reference does not respond to the input error
2* 2 change (i.e. difference between vDC and vDC ) fast enough and causes overshoot in the 2 DC bus voltage. An anti-windup loop is included in the proposed vDC controller in 2 order to solve this problem. vDC controller shown in Figure 6.2 with anti-windup loop is shown in Figure 6.9.
2 Figure 6.9 vDC controller including anti-windup control loop (The dotted line box is the “DC Voltage Control Block (VCB)” of Figure 5.1)
127
Chapter 6: DC bus voltage controller
DC Voltage (V) 55 Without antiwindup 50
45 40 With antiwindup 35
q-Current Reference (A) 200
0
-200 Without antiwindup
-400 With antiwindup
-600 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 Time (sec)
Figure 6.10 Changeover to generation mode at 600 rev/min (i) DC voltage (ii) q current
A simulation test is carried out to evaluate the effectiveness of the proposed DC voltage controller with anti-windup loop when changeover from motoring to generation mode. Figure 6.10 shows the comparison of the simulation results with and without anti- windup loop. As can be seen in this figure, the q current reference responds quickly in the case with anti-windup loop which results in lower overshoot in DC bus voltage.
Figure 6.11, Figure 6.12 and Figure 6.13 show simulation results for regaining DC voltage regulation after short-duration high power demand. The simulation is conducted at 5000rev/min with 100A initial load which exceeds the power capability of the ISA. At time = 0.2, the load is change to 40A which is within the power capability of the ISA. As may be seen in Figure 6.11, the DC voltage is approximately 35V when the load is 100A and then regulates to 42V when load is dropped to 40A. A significant DC voltage overshoot can be observed for the case without anti-windup loop. The controller with anti-windup loop is able to respond quickly and change the q reference so that the voltage regulation is regained without large overshoot.
128
Chapter 6: DC bus voltage controller
120 100 80 (A) L
I 60 40 20
60 Without AWL 50 (V)
DC 40 V With AWL 30
0 With AWL
-100 Without AWL
reference(A) -200 e* qs
I 0 0.2 0.4 0.6 0.8 1 Time (sec)
Figure 6.11 DC load current, DC voltage and q axis current reference (i) with anti-windup loop (AWL) and (ii) without anti-windup loop.
120 100 80 (A) L
I 60 40 20
-20 With AWL -40
(V) -60
DC Without AWL I -80 -100
20 With AWL 0
(A) -20 BAT I -40 Without AWL -60 0 0.2 0.4 0.6 0.8 1 Time(sec)
Figure 6.12 DC load current, DC bus current and battery current (i) with anti- windup loop (AWL) and (ii) without anti-windup loop. 129
Chapter 6: DC bus voltage controller
0
-2000 With AWL -4000 (A) e*1 qs I -6000 Without AWL -8000
-50
-100
(A) With AWL e* qs I -150 Without AWL
-200 0 0.2 0.4 0.6 0.8 1 Time (sec)
e*1 e* Figure 6.13 iqs and iqs current (i) with anti-windup loop (AWL) and (ii) without anti-windup loop.
Figure 6.12 shows the load current, DC bus current and battery current during the above test. As may be seen, the ISA supply 93A and battery supply 7A when load is 100A. When the load current is dropped to 40A, the ISA supplies 46A for both load and charging current for the battery while regulating DC voltage to the set value. Currents for both cases (i.e. with and without anti-windup loop) are indicated in this figure.
e*1 e* Figure 6.13 shows q axis current during this test. iqs and iqs are the q axis current before and after the saturation block, respectively of the block diagram shown in Figure 6.9. As may be seen, the integrator of controller without anti-windup loop continues to integrate during the period where the DC voltage regulation is not possible due to physical limitation of the system. The controller with anti-windup loop, the integrator stops at an output value (magnitude) slightly higher than the saturation limit. This allows quick response when load is reduced to a value within its power capability.
Next section discusses the sizing of the DC bus capacitor bank of the ISA inverter.
130
Chapter 6: DC bus voltage controller
6.4 Sizing DC bus Capacitor for ISA inverter
This section discusses sizing of DC bus capacitors in the ISA inverter. The DC bus capacitors are used to filter out the ripples caused by current pulses in DC side of the inverter. The proper sizing of capacitor bank is important as it helps protecting battery from ripples as well as for the operation of inverter when battery is not in the circuit. The lager the DC bus capacitor, the smoother is the DC voltage. However, capacitors are usually bulky and expensive. As a result, there exists a trade-off between the quality of DC bus voltage, the compactness and the cost. Thus, the capacitors should be chosen based on permitted magnitude of ripple in the DC bus voltage. In addition to reducing ripples, DC bus capacitor provides valuable support for reducing overshoots in the DC bus voltage.
iDC iI
iCAP ias vas Cdc a v ibs vbs DC b i v c cs cs
Figure 6.14 A schematic diagram of three phase inverter
Figure 6.14 shows three phase inverter with DC bus capacitor. By applying KCL to the DC-side of the inverter,
dv CiiiDC (6.10) dc dt CAP DC I
The DC voltage ( vDC ) can be written in steady-state as the summation of constant DC component v and ripple component ( v ). DC con ripple
vv v (6.11) DC DCcon ripple 131
Chapter 6: DC bus voltage controller
By substituting (6.11) to (6.10),
dv() v dv CCiDCcon ripple ripple i (6.12) dc dt dc dt DC I
The frequency of the voltage ripple is equal to the switching frequency as the current pulses present in the DC side are due to switching of the devices. As a result of high switching frequency applied to inverter, the magnitude of voltage ripple is significantly small compared to the average DC bus voltage. Also, the periodic time of the ripple is small. By assuming small magnitude and periodic time of the ripple, equation (6.12) can be rewritten as follows[72]:
v CiiDC (6.13) dc t DC I
By rearranging (6.13), change in DC bus voltage can be written as follows:
iit DC I vDC (6.14) Cdc
As can be seen in (6.14), the maximum change in voltage occurs when iDC is maximum and iI is minimum. The maximum iDC corresponds to maximum load. The minimum iI value for any switching combination of inverter is I m , where I m is the peak stator current of the motor. The time period for this voltage change is half a switching cycle 1 (i.e. t ). The minimum capacitor value required to maintain DC bus voltage 2 fsw ripples less than given peak-peak ripple voltage v can be obtained as follows: DC A
iIt iI C DC m DC m (6.15) dc 22 vf vf DCswDCswAA
The DC bus capacitor value is calculated for the highest loading condition of generation mode of ISA as follows: iA133 , I 220 2A , vV0.5 (i.e. approx 1.2%) and f 10kHz DC m DC A sw
132
Chapter 6: DC bus voltage controller
By substituting above values into (6.15), the minimum DC bus capacitor value is calculated to be ,CmFdc 44 . A capacitor bank of 50mF is selected for the prototype ISA system discussed in this thesis.
A simulation test was conducted with varying sizes of capacitors to investigate the effect of the capacitor value on DC bus voltage regulation and the quality of the DC bus voltage (i.e. ripple content). The test was conducted without battery connected to the DC bus.
60
58
56 20mF 54
52
(V) 50 DC V 48 50mF 46 80mF 44
42
40 0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 Time (sec)
Figure 6.15 DC bus voltage transients for 133A load dump at 2000rev/min for DC bus capacitance of (i) 20mF (ii) 50mF (iii) 80mF.
Figure 6.15 shows simulation results for DC bus voltage for 133A load dump at 2000rev/min. The load 133A at 2000rev/min corresponds to the maximum power point of the ISA in generation mode and therefore this is the maximum load dump that can be possible in ISA operation. The simulation results for 20mF, 50mF and 80mF are shown in this figure. With 20mF, the load dump transients almost exceeds the specified range by the proposed 42V standard, whereas with 80mF and 50mF and the transient over voltages stay below 50V. Figure 6.16 shows the transients when 133A load at 2000
133
Chapter 6: DC bus voltage controller
rev/min is applied. As may be seen, the case with 20mF shows oscillations during transient before regaining the DC voltage regulation.
45
40 80mF 50mF 35 (V) DC V 20mF 30
25
20 0 0.1 0.2 0.3 0.4 0.5 Time (sec)
Figure 6.16 DC bus voltage transients for 133A load application at 2000rev/min for DC bus capacitance of (i) 20mF (ii) 50mF (iii) 80mF.
50mF
42.2
(V) 42 DC V 41.8
80mF
42.2
(V) 42 DC V 41.8
20mF
42.2
(V) 42 DC V 41.8 0.6 0.65 0.7 0.75 0.8 0.85 0.9 0.95 1 Time (sec)
Figure 6.17 DC bus voltage ripples for (i) 50mF (ii) 80mF (iii) 20mF. 134
Chapter 6: DC bus voltage controller
Figure 6.17 shows the ripple in the DC bus voltage for different values of capacitance. As may be seen, the DC bus with 20mF capacitor bank shows higher ripple content compared to the other two cases. The larger capacitor value (80mF) results in lower amount of ripple on the DC bus.
6.5 Experimental Results
The prototype ISA with the proposed control design was tested for voltage regulation during load and speed variations. Figure 6.18 shows transient of the prototype ISA during load application/dump test at 2000rev/min. In this test, a load of 120A is applied at t 1sec and dumped at t 6sec. DC voltage, d axis current and q axis current transients are shown in this figure.
150 100 DC Load Current(A) 50 0
50 DC Voltage (V) 45 40
35
100 d-Current (A)
50
0
0
-200 q-Current (A)
-400 1 2 3 4 5 6 7 8 9 10 Time (sec)
Figure 6.18 Experimental results for 120A load application / dump at 2000rev/min (i) DC load current (ii) DC voltage (iii) d current (iv) q current
135
Chapter 6: DC bus voltage controller
Figure 6.19 Experimental results: DC bus voltage transients for full load dumps at different speeds.
5000 Speed (rev/min) 3000
1000 0 50 DC Voltage (V) 45
40
100
50 d-Current (A) 0 100 0
-100 q-Current (A) -200 1 2 3 4 5 6 7 8 9 10 Time (sec)
Figure 6.20 Experimental results for transient during acceleration and deceleration of engine speed at no-load (i) speed (ii) DC voltage (iii) d current (iv) q current 136
Chapter 6: DC bus voltage controller
Figure 6.19 shows experimental results for DC bus voltage transients during full load dumps at speeds of 1000, 2000 and 4000 rev/min. As can be seen in this figure, the voltage overshoots stay below 50V and settles back to 42V within a short time. The voltage transients are well within the limits specified in proposed 42V ISO/WD 21848-2 standards.
Figure 6.20 shows experimental results for transients during acceleration and deceleration of the engine speed at no load. The DC voltage transients due to speed acceleration and deceleration ramps are almost unnoticeable due to good voltage regulation.
6.6 Summary
This chapter proposed a sophisticated DC voltage controller for the ISA that provides tight voltage regulation required by future automotive power system. The proposed regulator design is based on a linearised model for the combined inverter and induction
2 machine. The control variable is selected to be vDC instead of vDC as this allows application of the linearised transfer function for control design. The proposed DC controller is included with decoupling to remove the effect of speed and rotor flux on the DC voltage regulation. Moreover, an anti-windup loop is added to prevent voltage overshoot caused by the winding up of the integrator of the DC voltage regulator. An extensive simulation study was conducted to investigate the performance of the proposed DC voltage control. Simulation results demonstrate the performance of the DC voltage controller. Sizing of the DC bus capacitor of the ISA inverter was also discussed in this chapter. Finally, the experimental verification of the proposed controller was presented. Experimental results show excellent DC voltage regulation during load application, load dump and engine speed acceleration / deceleration. The voltage dump transient stay well within the specified voltage limits of proposed 42V standard under various operating conditions of ISA.
137
Chapter 7: Field weakening operation of integrated starter alternator
CHAPTER 7
FIELD WEAKENING OPERATION OF INTEGRATED STARTER ALTERNATOR
7.1 Overview
This chapter presents proposed designing of field weakening (FW) control of generation mode of ISA. Optimal field weakening is important in ISA application in order to exploit maximum capabilities of the ISA hardware throughout the operational speed range of the generation mode. The optimal field weakening in generation mode of ISA can be seen as the choice of proper rotor flux and q axis current reference that results in maximum DC power output within maximum voltage and current limits of the induction machine and inverter. In classical field weakening method, commonly known as 1/ r method, the rotor flux reference is reduced as inversely proportional to the rotor speed at speeds above the base speed. However, the flux reference yielded by 1/ r method is too high and it reduces available voltage margin that is essential for maintaining the torque producing current; hence, the maximum torque can not be achieved [73, 74]. Several techniques have been proposed to solve the field weakening problem [73-77]. These methods can be divided into two broad categories; (i) calculation of rotor flux reference and limit of q axis current using induction machine parameters [74, 75], (ii) obtaining rotor flux reference and q axis current limit by controlling stator voltage of the induction machine [76, 77]. The later method is less sensitive to the parameter variation, particularly, variation of magnetising and leakage inductances. In the proposed ISA, a stator voltage controlled field weakening method is applied for generating mode (i.e. induction generation) of ISA. The method applied for the proposed ISA is a modified implementation based on the method reported in [76] for induction motor drive. There is a noteworthy distinction between purpose of FW in generation mode of ISA and conventional motor drive. In a motor drive, optimal FW weakening is a dynamic event that produces maximum torque to obtain maximum acceleration. In generation mode of ISA, optimum field weakening is a steady-state
138 Chapter 7: Field weakening operation of integrated starter alternator
event that produces the maximum power of the ISA system for a given short time period. During this period DC bus voltage regulation may not be functioning as the ISA power capability is not sufficient to satisfy DC power demand. The battery’s contribution is required to fulfil the DC load demand during this short time period. The allowed duration for this operation depends on the amount of remaining charge in the battery. After this allowed period, the load required to be reduced to a level so that ISA power generation is sufficient for satisfying the load as well as charging the battery. This operation usually occurs in higher speeds where power capability of ISA is weakened.
The modifications to the stator voltage control field weakening proposed in this thesis are as follows:
(1) Designing of controllers with decoupling terms to remove the coupling of stator frequency so that the controllers maintain performances throughout the wide speed range.
(2) Adding an anti-windup loop to prevent integrator winding up of q axis voltage regulator when ISA operates at speeds where power is limited by the current limit (i.e. constant torque region)
(3) Adding an anti-windup loop to prevent integrator winding up of d axis voltage regulator when induction machine does not operate with maximum power in high speed region (i.e. FW-2 region).
(4) Applying nonlinear dynamic compensator (NDC) to control d axis voltage in order to reduce oscillations caused by large load application at high speeds region (i.e. FW-2 region).
Next section of this chapter discusses the principle of the stator voltage control based field weakening method in generation mode of induction machine ISA.
7.2 Stator Voltage Control Field Weakening in Generation Mode
7.2.1 Principle of Operation
The optimum field weakening in generation mode of ISA can be stated mathematically as maximisation of PDC (i.e. DC output power) under following constraints.
139 Chapter 7: Field weakening operation of integrated starter alternator
ee222 ()vvVqs () ds max (7.16)
ee222 ()iiIqs () ds max (7.17) where
Vmax and Imax are the maximum peak phase voltage and peak phase current capability of the inverter or induction motor.
Steady state relationship between stator current and voltage of induction machine in d rotor flux oriented reference frame can be obtained by substituting Li' e 0 and sqsdt d Li' e 0 to (5.30) and (5.34) respectively as follows: sdsdt
L ee' e m e vRiLiqs s qs e s ds e dr (7.18) Lr
ee ' e vRiLids sds e sqs (7.19)
The voltage drop due to stator resistance is significantly small compared to the back EMF and cross coupling terms at higher speeds at which field weakening is of interest. By neglecting stator resistance drops, equations (7.18) and (7.19) can be simplified as follows:
L ee' m e vLiqs e s ds e dr (7.20) Lr
ee ' vLids e s qs (7.21)
ee By substituting steady state version of (5.3) (i.e. drLi m ds ) into (7.20) and simplifying
L ee' m e vLiLiqs e s ds e m ds Lr LL22 e mm edsiL() s (7.22) LLrr e esdsLi
140 Chapter 7: Field weakening operation of integrated starter alternator
By substituting (7.21) and (7.22) into (7.17),
2 2 CSve CSve DTds DTqs 1 (7.23) ' DT EUesLImaxEUes LI max
The output power in generation mode of ISA is given by,
PviDCDCDC (7.24)
The negative sign in (7.24) corresponds to output power with respect to the direction of DC bus current marked in Figure 3.4 of Chapter 3.
By substituting (3.38) to (7.24),
PviiiDC DC ()I CAP R (7.25)
The average DC current through the capacitor is zero (i.e. iCAP 0 ). The current through the capacitor leakage resistance is small and therefore can be neglected (i.e. iR 0 ). By applying these conditions to (7.25),
PviDCDCI (7.26)
By substituting (3.37) to (7.26),
3 Pvivi()ee ee (7.27) DC 2 qsqs dsds
By substituting (7.21) and (7.22) into(7.27),
3 vvee vv ee P () qs ds ds qs DC ' 2 esLL es 31CS 1 DTvvee (7.28) ' ds qs 2 EUesLL es 3 L2 m vvee ' ds qs 2 LLLrsse
The magnitude of output power can be represented by taking absolute value of (7.28) as follows:
141 Chapter 7: Field weakening operation of integrated starter alternator
3 L2 K Pv m eevv ev e (7.29) DCq' sdsqsds 2 LLLrsse e
3L2 m where , K ' is a constant for given induction machine. 2LLLrss
The optimum field weakening optimisation can be described as maximization of (7.29) under constraints (7.16) and (7.23). Equation (7.16) characterises the voltage limit circle ee in vvqs ds plane. The maximum peak phase voltage, Vmax is usually limited by the inverter maximum voltage capability rather than the motor maximum voltage capability. ee Equation (7.23) represents current limit ellipse (CLE) in vvqs ds plane. The current limit ellipse (CLE) expands with increasing stator frequency (i.e. increasing speed). ee The constant DC output power contours have hyperbolic shape in vvqs ds plane and are characterised by (7.29). These power hyperbolas (PH) move away from the origin when DC output power increases at a given speed. The voltage limit circle, current limit ellipse and power hyperbolas are indicated in Figure 7.1.
Figure 7.1 Voltage limit circle, current limit ellipses and DC power ee hyperbolas in vvqs ds plane
142 Chapter 7: Field weakening operation of integrated starter alternator
The maximisation of DC power output at a given speed correspond to the power ee hyperbola that stays furthest away from the origin of vvqs ds plane but yet has at least a point inside or boarder of the common area to both voltage limit circle and current limit ellipse. This single common point is the maximum power operating point for given frequency. At low frequencies, the maximum power operating point is the tangent point between power hyperbola and the current limit ellipse. Above base frequency, the maximum power hyperbola passes through intersection point between the current limit ellipse and voltage limit circle. At high speeds, the maximum power point is the tangent point between power hyperbola and voltage limit circle. Figure 7.2 shows ee e e generation quadrant of vvqs ds plane (i.e. both vqs and vds axes are positive) with three sample maximum power operating points (MPOP) at different stator frequencies.
Figure 7.2 Three sample MPOPs at three different stator frequencies in generating operation (i.e. at three different frequencies)
The MPOP at 40Hz belongs to constant torque operation region of the induction machine ISA and this operation is constrained only by the current limit ellipse. The MPOP at 100Hz operation is constrained by both voltage and current limits (i.e. both voltage limit circle and CLE) and this operating point belongs to felid weakening
143 Chapter 7: Field weakening operation of integrated starter alternator
region-1 (FW-1). The MPOP at 250Hz operation is constrained only by voltage limit and this operating point belongs to field weakening region-2 (FW-2). All these are illustrated in Figure 7.2.
ee Figure 7.3 Operation trajectory over complete speed range in vvqs ds plane.
Figure 7.3 shows the maximum power operation trajectory of induction machine ISA ee over complete speed range in vvqs ds plane. The dashed-line and solid-line with arrows show starting and generating operation of ISA respectively. Straight-line section corresponds to constant torque operation. The arc on the voltage limit circle corresponds to FW-1 region. The single point in the end of the arc corresponds to FW-2 region. The direction of arrows indicates increase in stator frequency (i.e. speed). The point corresponding to maximum power operation in FW-2 is the point that tangent of power hyperbola touches the voltage limit circle. The power hyperbola is symmetrical around ee 45° axis of the vvqs ds plane and therefore, the maximum power in FW-2 occurs when following condition is satisfied.
V vveemax (7.30) qs ds 2
144 Chapter 7: Field weakening operation of integrated starter alternator
The conditions for maximum power generation of ISA can be summarized as follows:
The rated flux is applied to induction machine, if stator voltage is below Vmax . This operation is constant torque operation.
Increase the speed while rated flux is applied causes increase in stator voltage. At certain speed, the stator voltage tends to increase beyond the maximum
voltage Vmax . The magnitude of stator voltage is then regulated to Vmax . This operating region is FW-1.
e The magnitude of d axis stator voltage, vds increases with the increase of
speed for given q axis current. Above certain speed, d axis stator voltage V attempts to exceed max . Then, the magnitude of d axis stator voltage, ve 2 ds V needs to be regulated to max . In addition, the stator peak voltage (i.e. 2
ee vvqs ds ) needs to be controlled to Vmax . When these two conditions are satisfied, the ISA generates maximum power in FW-2 region.
As may be noticed in this section, the inverter losses and stator resistance of the induction machine was neglected when deriving condition for field weakening operation. In case of generation mode, unlike in the motoring operation, theses assumptions are used to simplify the field weakening optimisation. Next section investigates significance of the error caused by the assumptions of neglecting of inverter losses and stator resistance of the induction machine on field weakening.
7.2.2 Effect of Stator Resistance and Inverter Losses on Field Weakening Optimization
This section discusses and analyses the effects of assumptions made in the FW optimisation presented in the previous section. The main assumptions are: (1) neglecting the stator resistance of the induction motor (2) neglecting the losses in the converter. The field weakening optimisation can be rewritten without these assumptions for generation mode of ISA as follows:
Maximisation of PDC under following constraints
145 Chapter 7: Field weakening operation of integrated starter alternator
ee222 ()vvVqs () ds max (7.31)
ee222 ()iiIqs () ds max (7.32) where, 3 PviviP()ee ee (7.33) DC2 qsqs dsds conlosses_ ee e vRiLiqs sqs e sds (7.34)
ee ' e vRiLids sds e sqs (7.35)
By substituting converter losses given in (3.57) to (7.33)
3 PviviAiiAiiBvivi()ee ee e22 e ()() e 22 e ee ee DC2 qsqs dsds121 qs ds qs ds qsqs dsds (7.36) 6 Bvi()eeeeee vi i22 i () E ' E ' f iee 22 i 2 qs qs ds ds qs ds T D sw qs ds
30
20 PH
Voltage 10 Limit
e qs 0 V
-10 Current Limits
-20
-30 -30 -20 -10 0 10 20 30 Ve ds
ee Figure 7.4 DC power contours on vvqs ds plane with current ellipses and voltage limit circles (without assumption:-doted line, with assumption:-solid line)
146 Chapter 7: Field weakening operation of integrated starter alternator
DC power ( PDC ) contours with voltage and current constraints for the cases with and without assumptions are shown in Figure 7.4. The contours given in this figure were calculated for induction machine ISA and IGBT inverter using MATLAB tools. Figure 7.5 shows the calculated maximum DC power output Vs speed with and without assumptions.
5500
5000
4500
4000
3500 Power (watts)
3000
2500 Without Assumptions With Assumptions
2000 0 1000 2000 3000 4000 5000 Speed (RPM)
Figure 7.5 Calculated maximum DC power output with and without assumptions
As may be seen in Figure 7.4 and Figure 7.5, the assumption made for simplifying optimisation does not have large impact on the power capability of the ISA. Therefore, in this thesis, the simplified optimisation discussed in Section 7.2.1 is used for FW control.
Next section discusses the implementation of field weakening control in generation mode of ISA.
147 Chapter 7: Field weakening operation of integrated starter alternator
7.3 Implementation of Field Weakening Optimization
7.3.1 Overview of the Proposed Implementation
The conditions for field weakening operation for generation mode discussed in Section 7.2.1 are implemented as shown in Figure 7.6. The proposed implementation consists of e d axis stator voltage controller ( vds -NDC), flux controller and q axis stator voltage controller. The stator voltage of the induction machine is regulated to Vmax via q axis stator voltage controller in FW-1 and FW-2 regions. The q axis stator voltage is selected as control variable instead of peak stator voltage in order to obtain approximate linear relationship for controller design. Details of this will be discussed in later part of this section. It was observed that large DC bus load disturbances cause oscillations in the FW-2 region due to the nonlinear nature of the system seen by the d axis stator voltage controller when proportional integral (PI) controller is used. A Nonlinear Dynamic Compensator (NDC) is applied in order reduce the oscillations in FW-2 region as shown in Figure 7.6. Nonlinear nature of the system and design of NDC will be discussed in detail in the later section of this chapter. The dependency on the stator frequency, e is removed (i.e. decoupled) from both d and q axis stator voltage control loops so that controllers can maintain desired dynamics throughout wide speed range without interaction with other controllers. A flux controller is included in the proposed implementation to ensure fast flux rotor flux build-up. As indicated in the Figure 7.6, anti-windup loops are included for both d and q axis stator voltage control loops. The purpose and design of anti-windup loops will be discussed in the later part of this chapter.
Next sections discuss the proposed field weakening implementation in detail which e e includes systematic designing of flux controller, vqs controller and vds controller given in Figure 7.6.
148 Chapter 7: Field weakening operation of integrated starter alternator
e dr
K Aqs e vqs e* K ids e 2 e Ilvd rated vds Vvmax ds s e1* e* dr dr
e vqs ie* ds e 2* e Iimax ds e vds
i I max KIlvq p qs Vmax 2 s
U e vds K Ads
Figure 7.6 Block diagram of proposed implementation of stator voltage controlled field weakening method
7.3.2 Field Weakening Controller Designing
7.3.2.1 Flux Controller
This section discusses the designing of the flux controller which ensure fast build up of the rotor flux. The flux controller enables induction machine to be operated on comparatively high d axis current momentarily, which results faster flux build-up compared to applying required direct d axis reference.
Closed current loop e* e* e e K i i L dr dr + If ds 1 ds m - KPf s sTr 1 e dr
Figure 7.7 Rotor flux transfer function and controller in a closed loop
149 Chapter 7: Field weakening operation of integrated starter alternator
Figure 7.7 shows rotor flux transfer function derived in Chapter 5 in a closed loop with flux controller. As can be seen in this figure, the inner d axis current dynamic is assumed much faster compared to the flux dynamics and therefore the transfer function of d axis current is assumed to be unity for simplifying flux controller design. The closed-loop transfer function of rotor flux can be written as follows:
e sK L K L dr Pf m If m (7.37) e*2 drsT r s(1 KPf L m ) L m K If
By applying ITAE criterion to (7.37), the gains of rotor flux PI controller can be obtained as follows:
(3.2 T 1) nf r KPf (7.38) Lm
2 T nf r KIf (7.39) Lm where,
KPf and KIf are proportional and integral gain of the rotor flux controller respectively.
nf is the natural frequency of the closed rotor flux control loop.
The natural frequency of outer flux loop is chosen to be much lesser than the natural frequency of inner current loop (i.e. nc nf ) to avoid interaction between inner and outer control loops.
e 7.3.2.2 q axis voltage controller ( vqs Controller)
As discussed earlier in Section 7.2, the stator voltage of the induction machine (i.e.
ee vvqs ds ) needs to be controlled to Vmax in order to satisfy the condition for maximum power operation in both FW-1 and FW-2 regions. In the proposed field weakening
ee implementation, the regulation of stator voltage (i.e. vvqs ds ) is achieved via
e vqs controller, since this allows approximate linearization of the transfer function for
150 Chapter 7: Field weakening operation of integrated starter alternator
e e vqs controller design. Relationship between rotor flux and vqs is given in (7.20) and can be re-written as follows:
L ee' m e vLiqs e s ds e dr (7.40) Lr
' e As illustrated in Figure 5.12 in Chapter 5, the cross coupling term, esdsLi is e significantly small compared to the back EMF component, emrdrLL . By
' e neglecting cross coupling term, esdsLi , an approximated relationship between q axis e e voltage ( vqs ) and rotor flux ( dr ) can be written as follows:
L ee m vqs e dr (7.41) Lr
e The closed-loop control block diagram for vqs controller design is shown in Figure 7.8.
Closed flux loop e* e* e e v vqs qs KIlvq dr dr Lm +- 1 e s Lr vqs e e
e Figure 7.8 vqs controller and approximated transfer function in a closed loop.
e As can be seen in Figure 7.8, the output of vqs controller is divided by stator frequency, e in order to decouple it from the control loop. The closed-loop transfer function can be written as follows:
vKLe qs Ilvq m (7.42) e* vsLKLqs r Ilvq m
151 Chapter 7: Field weakening operation of integrated starter alternator
The closed loop transfer function given in (7.42) has first order characteristics and therefore, only integral gain is used. The integral gain of controller can be obtained as follows:
L nlvd r KIlvq (7.43) Lm where, e KIlvq is the integral gain of vqs controller.
e nlvd is natural frequency of the vqs closed loop
The natural frequency of this closed-loop is chosen to be much lesser than that of the inner flux loop (i.e. nf nlvd ) to avoid interaction between inner and outer loops.
e 7.3.2.3 d axis voltage controller ( vds Controller)
As discussed earlier in Section 7.2, for maximum power operation in FW-2 region, the V condition of ve max needs to be satisfied. The q axis current reference needs to ds 2 be limited to a value which corresponds to the condition for maximum power in FW-2
e region. This current limit is obtained via controlling vds as illustrated in Figure 7.6.
e The closed loop design of vds controller based on approximate linear relationship is described below.
e A relationship between absolute value of d axis voltage, vds and absolute value of
e q axis stator current, iqs can be written from (7.21) as follows:
ee ' vLids e s qs (7.44)
e Figure 7.23 shows vds controller and approximate transfer function in a closed control loop.
152 Chapter 7: Field weakening operation of integrated starter alternator
q-current loop e e* e* e v vds ds iqs iqs K Ifvd ' +- 1 L s s ve ds e e
e Figure 7.9 vds closed control loop
As can be seen in Figure 7.9, the output of integral controller is divided by stator frequency, e in order to decouple the stator frequency from control loop. The closed
e loop transfer function for vds controller can be written as follows:
ve KL' ds Ifvd s (7.45) e* ' vds sKIfvd L s
This is a first order system and the integral gain can be estimated as follows:
nfvd KIfvd ' (7.46) Ls where,
e KIfvd is the integral gain of the vds controller
nfvd is the natural frequency of the closed control loop.
Next section discusses the simulation results for the modelled system with above discussed control design.
7.3.3 Simulation Results
Simulation tests were conducted to evaluate the performance of the proposed field weakening method. Figure 7.10, Figure 7.11, Figure 7.12 and Figure 7.13 show results for engine acceleration test where the engine is accelerated from 1000rev/min to 5000rev/min in 1second. The DC bus of the ISA is loaded to 100A during this test. As
153 Chapter 7: Field weakening operation of integrated starter alternator
may be seen in Figure 7.10, the stator voltage of the machine is regulated to peak stator e e voltage, VVm 18 , and vqs and vds voltages are regulated to 12.72V (i.e.
ee vvqs ds 18 2 ) at higher speeds, which satisfy the requirement for maximum power. The selected load of 100A exceeds the power capability of the ISA system at higher speed. This can be noted in Figure 7.11 where DC voltage regulation is lost at higher speeds and the battery starts supplying the additional power required to satisfy 100A load. This operation is allowed for short periods under normal ISA operation as discussed in Chapter 6. Figure 7.12 shows torque, q axis stator current, flux and d axis stator current during the engine speed acceleration. The tracking of conditions for maximum power can be seen in Figure 7.13. Initially, the speed is 1000rev/min and the induction machine operates with rated flux at point-A. When speed is increased, the stator voltage is regulated to VVm 18 , which corresponds to the circle indicated. At e e higher speeds (i.e. FW-2), vqs and vds voltages are regulated to 12.72V (i.e.
ee vvqs ds 18 2 ), which corresponds to point-B. The arrows in the figure indicate the direction of movement of the operating point under speed acceleration.
6
4
Speed 2
(1000rev/min) 0
20
18 (V) m
V 16
14 20 Ve (V) 16 qs
e ds 12 8 & V 4 Ve e qs ds
V 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time (sec)
Figure 7.10 Speed, peak stator voltage and qd axis voltages during engine accelerate from 1000 to 5000rev/min with 100A load on DC bus
154 Chapter 7: Field weakening operation of integrated starter alternator
60
DC 50 (V) V 40
30
-60 -80
(A) -100 DC I -120 -140
10
(A) 0 BAT I -10 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time (sec)
Figure 7.11 DC voltage, DC bus current and battery current during engine accelerates from 1000 to 5000rev/min with 100A load on DC bus
20 0 -20 (Nm)
Torque -40 -60 -100
(A) -200 e qs I -300
1 0.8 0.6 (pu)
Flux 0.4 0.2
150
(A) 100 e ds I 50 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time(sec)
Figure 7.12 Torque, q axis current, rotor flux and d current during engine accelerates from 1000 to 5000rev/min with 100A load on DC bus
155 Chapter 7: Field weakening operation of integrated starter alternator
20
18
16 A 14
12 B (V) 10 e qs V 8
6
4
2
0 0 2 4 6 8 10 12 14 16 18 20 Ve (V) ds
Figure 7.13 Trajectory of q axis voltage and d axis voltage during engine accelerate from 1000 to 5000rev/min with 100A load on DC bus
6
4
Speed 2
(1000rev/min) 0
25
20 (V) m
V 15
10 Without stator freqency decoupling with stator frequency decoupling 30 e
(V) V 20 qs e ds
10 & V Ve e qs ds
V 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time (sec)
Figure 7.14 Speed, peak stator voltage and qd axis voltages during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus (i) with and (ii) without stator frequency decoupling
156 Chapter 7: Field weakening operation of integrated starter alternator
Figure 7.14 shows the effect of decoupling of stator frequency from d and q axis stator voltage control loops. As can be seen in this figure, the stator voltage shows a significant overshoot for the case without stator frequency decoupling. These overshoot may cause problems to the current control loop as the overshoots reduce the available voltage margin required for the high performance stable current control.
Figure 7.15, Figure 7.16 and Figure 7.17 show simulation test results for sudden reduction of load from 100A to 40A at 5000rev/min. As discussed earlier, 100A exceeds the power capability of the ISA at 5000rev/min and it requires support of the battery to satisfy the load requirement. This can be noted in the first graph of Figure 7.15. During this period, the ISA supplies the maximum power capability by regulating ee VVm 18 and vvqs ds 18 2 as can be seen in this figure. When load is reduced to
e e 40A, Vm regains the regulation to 18V after a short transient. vqs and vds voltages
ee are not required to regulate to vvqs ds 18 2 , since the operation at maximum power capability is not required to satisfy 40A load. When load is dropped to 40A the battery starts charging as the ISA can generates sufficient power to supply load and charging the battery. As may be seen in Figure 7.16, the DC bus voltage regulates to 42V when load is reduced to 40A. Torque, q axis stator current, rotor flux and d axis stator current are also shown in this figure. It may be noticed in this figure that the rotor flux of the machine is increased to slightly higher value when load is reduced. This is because the back EMF component of the motor needs to increase in order to compensate - R der to maintain the same stator terminal voltage. Figure 7.17 show the movement of ee operating point on vvqs ds plane during the load reduction. Initially the operating point is at A, which corresponds to maximum power. Then, it moves to point-B when load is reduced to 40A. Direction of movement of operating point is indicated by arrows.
157 Chapter 7: Field weakening operation of integrated starter alternator
120 I 80 L 40 0 -40 -80 I Current (A) I BAT -120 DC
20
18 (V) m
V 16
14
18 16 (V) 14 ds Vqs 12 V 10 ds & V 8 qs
V 6 0 0.2 0.4 0.6 0.8 1 Time (sec)
Figure 7.15 Load current, DC bus current, battery current voltage, peak stator voltage during sudden reduction of load from 100A to 40A at 5000rev/min.
50
(V) 40 DC V 30 0
-5 (Nm)
Torque -10 0
(A) -100 e qs I -200 0.25
0.2 Fr (pu) 0.15 100
50 (A) e ds I 0 0 0.2 0.4 0.6 0.8 1 Tim e (s ec)
Figure 7.16 DC voltage, torque, q stator current, rotor flux and d stator current during sudden reduction of load from 100A to 40A at 5000rev/min.
158 Chapter 7: Field weakening operation of integrated starter alternator
20
18 B
16
14 A 12
(V) 10 qs V 8
6
4
2
0 0 2 4 6 8 10 12 14 16 18 20 V (V) ds
Figure 7.17 Trajectory of q axis voltage and d axis voltage during sudden reduction of load from 100A to 40A at 5000rev/min.
7.3.4 Anti-windup Loops
The proposed implementation of stator voltage control field weakening consists of anti- e e windup technique for both vqs and vds controllers in order to stop winding-up of integrators. The purpose and operation of these anti windup loops are discussed below.
e 7.3.4.1 Winding up of vqs controller
e Winding up of integrator of vqs controller occurs when the speed is below FW-1 region. This is because the flux is saturated at rated flux by the saturation limit. The rated flux at speeds below FW-1 corresponds to stator voltages lower than Vmax . The integrator increases its output as it attempts to correct the difference. This causes the winding up of the integrator. In case the speed is increased to a speed within the FW-1 region, then the rotor flux needs to be reduced below the rated value to regulate the stator voltage. This may not happen quickly as the integrator output has already reached a high value due to winding up and needs longer time to reduce to the required value. The temporary higher voltage caused by the slower change flux reference temporarily
159 Chapter 7: Field weakening operation of integrated starter alternator
reduces the voltage margin available for current control and hence it may lose the control. To overcome this problem integrator needs to be stopped when the flux reference saturates at rated flux. In the proposed implementation, this is achieved by an anti-windup loop as indicated in Figure 7.6. Simulations for ISA with proposed implementation were conducted in order to investigate the effect of anti wind-up loop. Figure 7.18 and Figure 7.19 shows simulation results for a case where ISA operates at 800rev/min in generation mode for some time and then speed ramps up to 5000 rev/min in 1 second. Figure 7.18 shows the behaviour of input and output of the flux saturation e1* e* function (i.e. dr and dr ) indicated in Figure 7.6. As may be seen in Figure 7.18, the controller with anti-windup loop responds quickly and adjusts the output of flux saturation block to the required value when speed is ramped up.
5
3 Speed 1
(1000rev/min) 0
2 Without AWC (pu) 1 e1* dr
With AWC 0
2 Without AWC (pu) 1 e* dr
With AWC 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 Time (sec)
e Figure 7.18 With and without anti-windup loop for vqs control when speed ramp from 800 to 5000rev/min in 1 sec (i) speed (ii) input of flux e1* saturation function ( dr ) in Figure 7.6 (iii) output of flux e* saturation function ( dr ) in Figure 7.6
160 Chapter 7: Field weakening operation of integrated starter alternator
Figure 7.19 shows DC voltage, q axis stator current, d axis stator current and flux
e for vqs control with and without anti-windup loop. As can be seen, the control of the system is disturbed in the case without anti-windup loop due to delayed adjustment of flux reference.
50
(V) 40 DC V 30
100 0
(A) -100 e qs
I -200 -300 200
(A) 100 e ds I 0
1 0.5
Flux (pu) 0 0 0.5 1 1.5 0 0.5 1 1.5 (a) Time (sec) (b) Time (sec)
Figure 7.19 DC voltage, q axis stator current, d axis stator current and flux when speed is ramped up form 800rev/min to 5000rev/min in 1 sec e (a) with anti-windup for loop vqs control (b) without anti-windup
e loop for vqs control
e 7.3.4.2 Winding up of vds controller
e Winding up of vds controller occurs when the induction machine operates with
e e d axis stator voltage ( vds ) below the value of vds 18 2 . This occurs when induction machine operates under all speed and load conditions other than the maximum
161 Chapter 7: Field weakening operation of integrated starter alternator
power operating point in FW-2 region. The simulation results for effect of anti-windup e loop on vds control are shown in Figure 7.20 and Figure 7.21.
100
50
Load (A) 0 0 0.2 0.4 0.6 0.8 1 800 Without AWC 600 p I 400 With AWC 200 0 0 0.2 0.4 0.6 0.8 1 400 Without AWC 300 With AWC m ax qs I 200
100 0 0.2 0.4 0.6 0.8 1 Time (sec)
e Figure 7.20 With and without anti-windup loop for vds control when load of 100A is applied at 5000rev/min (i) load current (ii) input of the q axis current saturation function (ip ) indicated in Figure 7.6 (iii)
max output of the q axis current saturation function ( Iqs ) indicated in Figure 7.6
Figure 7.20 shows the input of q axis current saturation function (ip ) and the output of
max q axis current saturation function ( Iqs ) indicated in Figure 7.6, when a load of 100A is applied at 5000rev/min. As may be seen, when ISA operates with no load on the DC bus, the input of q axis current saturation function (ip ) increases to a value above 600 due to integrator windup. With anti-windup loop, the integrator is stopped and output stays at about 320. When the load is applied, the integral controller become active and e max regulate vds to 18 2 V, and derives the q current limit ( Iqs ). However, in the case
162 Chapter 7: Field weakening operation of integrated starter alternator
without anti-windup loop, the ISA loses its control due to the delayed change in q axis
max current limit ( Iqs ), as can be seen in Figure 7.21(b).
50
(V) 40 DC V 30 100 0 -100 (A)
qs -200 I -300 200 100 (A)
ds 0 I -100 0
(A) -100 DC I -200 0 1 2 0 1 2 (a) Time (sec) (b) Time (sec)
Figure 7.21 DC voltage, q axis stator current, d axis stator current and DC bus current when load of 100A is applied (a) with anti-windup e loop for vds control (b) without anti-windup loop for
e vds control
Oscillations in the q axis current can be observed when load is sudden load is applied in FW-2 region. These oscillations can be noticed in Figure 7.20 and Figure 7.21 (a). The reason for these oscillations and its mitigation will be discussed in the next section.
7.3.5 Oscillation Caused by Large Load Disturbances and its Mitigation
As mentioned in the previous section, oscillations in q axis current occur when a large load is applied in FW-2 region. The reason for this is the nonlinearity of the system that
163 Chapter 7: Field weakening operation of integrated starter alternator
e needs to be controlled by the vds controller. A control block diagram for the system
e that needs to be controlled by the vds controller (i.e. plant) is depicted in Figure 7.22. In this figure, the approximate relationship given in (7.21) is used to describe the
e e* relationship between vds and iqs of the induction machine. ip is the manoeuvrable
e variable (i.e. input to the plant) and vds is the control variable.
e* ids
2* e Imax ids e
max e i v p Iqs vmax ds 2 e e vds iqs
e vds e1* e* ve iqs iqs e ds ' vds U Ls e
e Figure 7.22 vds controller and the system that needs to be controlled by
e vds controller (i.e. plant) in closed loop
Figure 7.23 shows graphical representation of an example for approximated relationship
e between manoeuvrable variable, ip and control variable, vds of the plant (i.e. the
e system needs to be controlled by vds controller). The example given in this figure corresponds to stator frequency of e =1570 rad/s in FW-2 region under various level of loading.
164 Chapter 7: Field weakening operation of integrated starter alternator
20
18 vmax Linear 275A 16 2 A 250 A MPOP 14 225 A
12 209 A
| 175 A 10 e ds Saturation Region
|v 150 A 8 125 A 6 100 A 4 75 A 50 A 2 ie1*=25A qs 0 0 50 100 150 200 250 300 350 i p
Figure 7.23 An example approximated characteristics of the plant (system that e needs to be controlled by vds controller) (This example corresponds to stator frequency of e =1570 rad/s).
As can be seen in Figure 7.23, the system (i.e. the plant) shows combination of linear and saturation characteristics. The saturation voltage value for a given stator frequency depends on the level of loading (i.e. q axis current). The point - A corresponds to maximum power operating point (MPOP) in FW-2 region. Operation of the system close to point - A can be described as follows:
V V If ve is slightly larger than max (i.e. ve max ), the plant is operated on linear ds 2 ds 2 section.
V V If ve is slightly lower than max (i.e. ve max ), the plant is operated completely in ds 2 ds 2 saturation region.
165 Chapter 7: Field weakening operation of integrated starter alternator
Given the nonlinear nature of the plant around the maximum power operating point, a
e linear pure PI controller is not suitable for controlling vds . It causes oscillations in q axis current and DC bus voltage at FW-2 region when large load disturbance is
e applied. Figure 7.24 shows simulation results for field weakening with vds PI controller when large load disturbance is applied.
DC Voltage V (V) 44 dc
42
40
38 d-Stator Voltage Ve (V) 20 ds
15
10
5
0 q-Current, ie (A) qs 100 0
-100
-200
-300 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 Time (s)
Figure 7.24 Transients during full load disturbance at 5000rev/min with a PI e for vds -controller (i) DC bus voltage (ii) d axis stator voltage
(iii) q axis stator current.
In the proposed implementation, a non-linear dynamic compensator (NDC) instead of
e pure PI controller is used for vds -controller as shown in Figure 7.6, in order to reduce theses oscillations. As can be seen in Figure 7.6, the NDC used in this application consists of error limiting element and an integral controller, and NDC provides comparative phase advance for large amplitude error signals [78]. The integral component of the NDC compensator is designed as discussed in Section 7.3.2.3. Figure
166 Chapter 7: Field weakening operation of integrated starter alternator
7.25 shows simulation results for large load disturbance when the NDC is utilised for
e vds -controller.
DC Voltage V (V) 44 dc
42
40
38 d-Stator Voltage Ve (V) 20 ds
15 10
5
0 q-Current, ie (A) qs 100
0
-100 -200
-300 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 Time (s)
Figure 7.25 Transients during full load disturbance at 5000rev/min with NDC e for vds -controller (i) DC bus voltage (ii) d axis stator voltage
(iii) q axis stator current.
7.3.6 Experimental Results for Stator Voltage Regulation and Power Capability in Generation Mode
The prototype ISA was tested for verifying the proposed field weakening operation. Figure 7.26, Figure 7.27 and Figure 7.28 show the experimental results for acceleration of engine form 1000 rev/min to 5000 rev/min in approximately 2.2 second with 100A load on the DC bus. These results agree with the simulation results presented in Section 7.3.3. It may be noted that the speed acceleration time in experimental study is longer than that of the simulation study. This is due to the limitation of the drive system of the experimental setup.
167 Chapter 7: Field weakening operation of integrated starter alternator
6
4
Speed 2
(1000rev/min) 0
25
20 (V) m
V 15
10
20 Ve qs
(V) 16
e ds 12 8 e & V 4 V ds e qs
V 0 1 1.5 2 2.5 3 3.5 4 4.5 Time (sec)
Figure 7.26 Measured speed, peak stator voltage and qd voltages during engine accelerates from 1000 to 5000rev/min with 100A load
20
18
16
14 A 12
(V) 10 B e qs V 8
6
4
2
0 0 2 4 6 8 10 12 14 16 18 20 Ve (V) ds
Figure 7.27 Trajectory of measured q axis voltage and d axis voltage during engine speed accelerate from 1000rev/min to 5000rev/min with 100A load on DC bus 168 Chapter 7: Field weakening operation of integrated starter alternator
60
(V) 40
DC Voltage 20 -100
(A) -200 e qs I -300
1 0.8 0.6 (pu)
Flux 0.4 0.2
150
(A) 100 e ds I 50 1 1.5 2 2.5 3 3.5 4 4.5 Time(sec)
Figure 7.28 Measured DC bus voltage, q current, rotor flux (estimated) d current during engine accelerates from 1000 to 5000rev/min with 100A load on DC bus
6
Proposed Method 5
4 1/ r 3
2 Maximum Power Output (kW) 1
0 0 1000 2000 3000 4000 5000 Speed (rev/min)
Figure 7.29 Experimental results: maximum power output for proposed method and 1/ r method.
169 Chapter 7: Field weakening operation of integrated starter alternator
The prototype ISA was tested for the power generation capability with stator voltage control FW method and the conventional 1/ r method. The comparison is shown in Figure 7.29. The stator voltage control method provides significant increase in power generation capability, particularly in FW-2 region compared to the conventional method.
7.4 Summary
This chapter proposed an improved field weakening implementation for the integrated starter alternator in generation mode. The proposed field weakening implementation is based on stator voltage control of the induction machine. The proposed field weakening implementation included decoupling to remove the effect of stator frequency on the voltage regulation. Anti-windup loops were utilised to prevent winding up of the integrators of q and d axis voltage controllers when ISA operates in speeds where power is limited by the current limit in constant torque region; and in FW-2 region when operates below the maximum power point. Oscillations in q axis current and DC bus voltage were observed in FW-2 when large load is applied. The cause of the oscillations is identified as the nonlinearity of the plant seen by the d axis voltage controller. In the proposed implementation, these oscillations were mitigated by introducing Nonlinear Dynamic Compensator (NDC) to control the d axis voltage. The simulation and experimental results presented in this chapter demonstrates the effectiveness of the proposed improvements. The stator voltage control method of field weakening was compared with conventional (1/ r ) field weakening method for power capability in the generation mode of the ISA. The experimental results have clearly demonstrated that stator voltage control allows extracting significant amount of extra DC power from the ISA in generation mode.
170 Chapter8 Loss Minimised control of integrated starter alternator
CHAPTER 8
LOSS MINIMISED CONTROL OF INTEGRATED STARTER ALTERNATOR
8.1 Overview
Efficient operation of the power generator in automobile is one of the main performance requirements for future automotive onboard power system. There are two aspects for designing high efficient ISA system; efficient hardware (i.e. electric machine, inverter and current carrying conductors) and loss minimised or efficiency improved control. This chapter focuses on the later aspect which is loss minimised control of the ISA. The loss minimised control is usually possible at light load operation of the induction machine. Loss minimised light load operation results in significant energy saving in the long run as ISA operates below its maximum load capability for considerable portion of its life time.
Efficiency of the induction machine at light loads (i.e. loads less than rated loads) can be improved by properly selecting flux which results in better balance between copper and core losses of the machine. The steady-state rotor flux of field oriented control induction machine is proportional to the d axis stator current. The core losses of induction machine vary with the flux in the core. Higher flux in the core results in higher core losses. Thus, the core losses can be reduced by reducing the d axis current. However, when d axis current is reduced, q axis stator current has to be increased in order to produce the same electromagnetic torque (or power), since the electromagnetic torque is proportional to the product of d and q axes stator currents.
The increase in q axis stator currents causes increased stator current and hence increased copper loss in the machine. Therefore, there is a unique combination of d and q axes stator currents that results in optimum efficiency of the induction machine for a given electromagnetic torque. Loss minimised operation can be achieved by forcing induction machine to operate at this unique combination.
171 Chapter8 Loss Minimised control of integrated starter alternator
The techniques allowing efficiency improved operation of induction machine given in literature can be broadly divided into three categories. The first category is simple state controllers where a single variable such as power factor or slip of the induction machine is controlled to obtain better balance between copper and iron losses [79-81]. The second category is so called search controllers, in which flux or a corresponding variable is reduced until the measured input power is minimised [82-84]. Theoretically, search controller approach does not require knowledge of the parameters of the system as it is based on an iterative optimisation algorithm. However, practical search controllers have various drawbacks such as slow adaptation, continuous disturbance in the torque and need for precise information on load [85]. The third category is model based loss minimisation, in which the optimum flux reference is calculated using an induction machine loss model [86-91]. Obviously, this approach requires knowledge on parameters of the system in order to calculate the optimum flux reference. Most of the literature that discuss the loss model approach simply ignore the losses of the inverter by assuming the motor losses are comparatively high in case of medium-power and low-power drive. Some of the literature model the losses of the inverter as a linear resistance for simplicity [88].
This thesis utilises a model based loss minimisation technique taking into account the inverter losses. The model based approach suit well for the ISA application as the motor and inverter belong to a system, unlike general purpose variable speed drive application, the knowledge of the motor parameters is always available in such a case. Inverter losses are significant in 42V ISA application since the system operates with low voltage and high current. The proposed method utilises the inverter model that was discussed in Chapter 3 with some simplifications. Target speed range for efficiency improvement for the proposed ISA is 800-2000rev/min; this is because at higher speeds, the power output of the ISA is relatively low and therefore likelihood of light load operation is quite low compared to lower speed range (i.e. 800-2000rev/min). In addition this is the speed range that an engine of a mid size car usually operates in majority of the urban driving time.
Next section discusses the loss model of the system used for the loss minimised control proposed in this chapter. In Section 8.3, condition for loss minimisation and its
172 Chapter8 Loss Minimised control of integrated starter alternator
implementation will be discussed. Section 8.4 presents the experimental results for the loss minimisation method.
8.2 Loss Model of the System
This section discusses the system loss model consists of motor and three phase inverter. Figure 8.1 illustrates the power flow of the ISA system in generation mode. As illustrated in the figure, Pmech , PAC and PDC are mechanical power input, AC electric power generated by the induction motor and DC power output of the system, respectively.
PAC Pmech PDC P P mtot invtot
Figure 8.1 Power flow of the ISA system in generation mode
There are two loss component in the system; total induction machine losses ( Pmtot ) and total inverter losses ( Pinvtot ). The total loss of the system ( Psysloss ) is given by,
(( )*+, ( PPPsysloss mtot invtot
The motor losses are obtained using the model presented in [91]. The derivation of this model is given in Appendix-A. As may be noticed, this model neglects the stator and rotor leakage inductances of the induction motor for simplicity. This does not cause significant error in loss values in the speed range in which efficiency improvement is required by the ISA (i.e. 800-2000rev/min) as the voltage drops due to leakage reactance are not large at lower speeds. The core losses are included in the model as a
173 Chapter8 Loss Minimised control of integrated starter alternator
resistance parallel to the magnetising branch. The total induction machine losses given in (A.92) can be rewritten as follows:
ee22 PRiRtot Q qs D() r i ds (8.48)
CS CS 3 RR' 3 L22 where, RRDTcs r and RR() DTmr Qm DTs ' Dm rDT s ' 2 EURRrcs 2 EURRrcs
The loss model for three phase inverter discussed in Chapter 3 is used to obtain total loss modelling of the system. Equation (3.57) representing total inverter losses can be rewritten as follows:
6 ' 'ee 22 ee 22 eeee PAEEfiiAiiBviviinvtot ((12T D )) sw qs ds (qs ds )(1qs qs ds ds ) (8.49) ee ee e22 e Bvi2 ()qs qs vi ds ds i qs i ds
In (8.49), the loss terms associated with constants B1 and B2 (i.e. last two terms) are significantly small compared to the first two terms and thus can be neglected for simplicity. An example comparison of total estimated losses for the inverter used for the prototype, with and without theses terms (i.e. terms associated with B1 and B2 ) are shown in Figure 8.2.
With above assumption, the total inverter losses can be written as follows:
3 PRiiKii()()ee22 ee 22 (8.50) invtot2 I qs ds I qs ds
2 6 where, R A and KA() EEf'' I 3 2 IT1 Dsw
The total system losses (i.e. induction machine and inverter) can be written by substituting (8.48) and (8.50) into (8.47).
eeee222 2 PPPRiRiKiisysloss mtot invtot Q qs D() r ds I ( qs ds ) (8.51)
CS CS 3 RR' 3 L22 where, RRRDTcs r and RRR() DTmr QIsDT ' DrDT I s ' 2 EURRrcs 2 EURRrcs
174 Chapter8 Loss Minimised control of integrated starter alternator
1100
1000
With B and B terms 900 1 2
800
700
600 Total inverter loss (W)
500
400 Without B and B terms 1 2
300 -300 -200 -100 0 100 200 300 q-axis current (A)
Figure 8.2 Total estimated inverter losses with and without the e simplification of the model (at iAds 150 )
e ee222 Equation (8.51) can be represented in terms of iqs and im by substituting iiids m qs as follows:
e 22 PRRiRiKisysloss Q D() r qs D () r m I m (8.52)
Equation (8.52), represents the losses in the ISA system in terms of variables, q axis
e current (iqs ) and peak phase current (im ) of the motor.
Next section derives and discusses the condition for loss minimized control of the ISA system.
8.3 Loss Minimization
Loss minimisation problem can be stated as minimization of Psysloss for given mechanical power input ( Pmech ) at given speed. This can be represented mathematically as follows:
175 Chapter8 Loss Minimised control of integrated starter alternator
Max ( Psysloss ) under constraint PCmech 0 (8.53)
e By taking absolute value of q axis current (iqs ) of (8.52) and differentiating with respect to im ,
dP die sysloss e qs 2()2() RQDrqsRi RiKDrmI (8.54) dimm di
dP The minimum losses occur when sysloss 0 . The condition for minimum losses can be dim dP obtained by substituting sysloss 0 into (8.54) as follows: dim
e diqs 2()RiK Drm I (8.55) e dim 2() RD rQqsRi
If the friction and windage losses are neglected, the mechanical power input to the ISA in generation mode can be obtained from (5.2) of Chapter 5.
22 LLL2 333PPPmmmee ee e2 e Pimech qs dr r iiiiiqs ds r r qs m qs (8.56) 22LLLrrr22 22
By differentiating (8.56),
CSdieedi DT2 eeeqs 23qs 224iimqs iqsm i i qs 2 DT dP31P L EUdimmdi mech m (8.57) r 24 dimr22 L 2 2 ee iimqs i qs
dP For given constant mechanical power input at constant speed, mech 0 . Thus, dim equation (8.57) can be reduced to:
diee i i qs qs m (8.58) di e 2 2 m (2iiqs m )
176 Chapter8 Loss Minimised control of integrated starter alternator
By substituting (8.58) to (8.55), following relationship for magnitude of q axis current
e (i.e. iqs ) and im for minimised loss operation of the system can be obtained.
32ee22 2()RiKiiiRDrm ImmqsDr 2 () RKi Q 2 Iqs 0 (8.59)
Equation (8.59) is a cubic equation and solution of the equation provides im value for
e loss minimised operating for given iqs . The equation is implemented as a look-up table to avoid high computational requirement for solving the cubic equation online. The solution for (8.59) is graphically shown in Figure 8.3.
300
250
200 m i 150 1000 rev/min 2000 rev/min 3000 rev/min 100 4000 rev/min 5000 rev/min
50
0 0 50 100 150 200 250 300 |ie*| qs
e* Figure 8.3 Loss minimised trajectories of im and iqs for various speeds
A block diagram for implementation of loss minimisation technique together with field weakening method, which was discussed in the previous chapter, is shown in Figure 8.4. As may be seen, d axis current reference is adjusted based on speed and q axis current reference for minimising losses. The d axis current reference pass through a low pass filter to remove the fast changes caused be the dynamics in q axis current
177 Chapter8 Loss Minimised control of integrated starter alternator
reference. The maximum limit of d axis current reference is set by the field weakening algorithm. This implementation allows loss minimised operation at light load conditions and maximum power field weakening operation when high power output is required. The two dimensional lookup-table for the loss minimisation
e relationship (8.59) is indicated in this figure. The inputs of the lookup table are iqs and speed. The output of the lookup table is im .
e1* ie* iqs max qs Iqs max Iqs max Iqs
e dr e vqs e* e max ids vds Ids e
e* max Ids iqs e1* U im 2 ids ii2 e mqs r
Figure 8.4 Block diagram of flux reference generation of the ISA based on loss minimisation and field weakening (This is the “Flux Reference Block (FRB)” of Figure 5.1)
Experimental results of the prototype ISA with the loss minimisation method will be presented in the next section.
8.4 Experimental Results
The proposed loss minimisation technique is implemented within control system of prototype ISA. The prototype ISA is tested for efficiency improvement at various loading levels and operational speeds. Figure 8.5 shows induction machine and inverter
178 Chapter8 Loss Minimised control of integrated starter alternator
efficiency with output power when ISA speed is 2000 rev/min. As can be seen in Figure 8.5, the loss minimization control improves the efficiency for loading level up to 1kW.
Figure 8.5 Efficiency Vs DC power output at 2000 rev/min for conventional and loss minimized (LM) control (i) induction machine (ii) three phase inverter
Figure 8.6 Induction machine efficiency Vs speed for conventional and loss minimized (LM) control (i) for 250W DC load (ii) 500W DC load and (iii) 2000W DC load.
179 Chapter8 Loss Minimised control of integrated starter alternator
Figure 8.7 Rotor flux, stator current and torque variation with speed at 500W output power for conventional and loss minimized (LM) control
Figure 8.6 shows the variation of induction machine efficiency with operational speed of the ISA for output power levels of 250W, 500W and 2000W. The results indicate that significant efficiency improvement can be obtained for light load condition using the proposed loss minimized control. Figure 8.7 shows variation of rotor flux, stator current and torque vs. speed for 500W loading condition.
8.5 Summary
This chapter has proposed and tested loss minimised control of an integrated starter alternator. A model based loss minimised control method taking into account the induction machine and the inverter loss is applied to the ISA in order to improve the efficiency in generation mode. The loss model for the motor is based on simplified dq axes synchronous reference frame model that ignores the leakage inductances. The inverter model is based on a simplified version of the loss model derived in Chapter 3. The error caused by the simplification of inverter model was shown to be insignificant by means of an example simulation. The derivation of total system loss minimised conditions are also discussed. Moreover, this chapter discussed the implementation of the loss minimised conditions together with maximum power field wakening within a 180 Chapter8 Loss Minimised control of integrated starter alternator
same control system. Finally, experimental verification of the proposed loss minimised control is presented for varying load and operational speed. The experimental results demonstrates that proposed loss minimised control provides significant efficiency improvements under light load conditions of the ISA.
181 Chapter9: Conclusions
CHAPTER 9
CONCLUSIONS
9.1 Conclusion of this Thesis
This thesis started with a goal of addressing the challenging requirements of a future automotive power system using a squirrel cage induction machine as the low-cost generator. The essential requirements of an integrated starter alternator in generation mode are: tight DC voltage regulation, maximised power generation capability, increased efficiency and lowest possible cost. The induction machine was selected because of the many advantages of this machine topology such as lower cost; robustness against harsh operating conditions, inherent protection against short circuit fault, smooth torque, low vibrations and noise. Chapter 2 of this thesis has included a detailed comparison of different electric machines and review of power generation and integrated starter alternator topologies reported in the literature.
Detailed and up-to-date models of all the subsystems required for extensive computer simulations and systematic control design were developed in Chapter 3. The subsystems included induction machine, inverter, battery and the internal combustion engine. The dynamic models developed in this chapter provided the platform for understanding the problems and the basis for systematic control design which resulted in excellent performance in DC voltage regulation and field wakening operation achieved, which are reported in Chapter 6 and Chapter 7, respectively. The manipulation of inverter loss model in terms of two-axis quantities proposed in this chapter helped the straightforward integration of inverter and induction machine loss models. This formed the basis for efficiency improvement reported in Chapter 8.
Parameter variations of the induction machine were investigated in Chapter 4 in order to obtain a computer model for induction machine that closely resembles the actual machine. Variations of rotor resistance, rotor leakage inductance, magnetising inductance and core losses were determined by variable-frequency and variable-voltage
182 Chapter9: Conclusions
locked-rotor and no-load tests. The rotor resistance value at very low slip frequency (DC value) was found to be 75% of conventionally obtained value and increasing with the slip frequency. This rotor resistance variation very closely agrees with the results obtained from impressed stator current test and conventional 50Hz value for two extremes. The rotor leakage inductance was found to increase greatly with decreasing slip and it was greater than double the value of conventional rotor leakage inductance value. The magnetising inductance has significant variation with magnetizing current. At 60Hz, the highest magnetizing inductance was found to be 17% higher than the conventional value measured at the rated magnetizing current. The variation of the magnetizing inductance with stator frequency was also found in Section 4.3.2. of this chapter. Induction machine model incorporated with parameter variations found in this chapter was used in the computer simulation studies presented throughout the thesis in order to mimic the actual machine accurately.
Overview of ISA control design including current control of the proposed ISA was discussed in chapter 5. Systematic current control design including decoupling was found to be important in achieving good control in high speed region of ISA. Simulation and experimental results have demonstrated performance and improvements in current control of ISA. In addition, this chapter illustrated how the control methodologies proposed in Chapter 6, 7 and 8 were incorporated into the overall control of ISA.
Chapter 6 proposed and tested a new sophisticated DC voltage controller for the ISA which provides tight DC voltage regulation required by future automotive on board power system. This voltage controller is based on the linearized model of combined inverter and induction machine proposed in this chapter. The proposed DC voltage controller is also incorporated with decoupling to remove the effect of speed and rotor flux; and an anti-windup technique. The proposed DC voltage controller has provided solution to the load dump transient problem. It controls the full load dump DC over voltage transient amplitude to be less than 50V and settling time of 150ms, which is well within the maximum dynamic over voltage envelop of 58V for 400ms in the drafted 42V PowerNet standards. The proposed DC voltage controller has demonstrated similar excellent full load dump transient control over the operational speed range of ISA. The DC voltage overshoots caused by the changeover from motoring to generation mode of ISA are eliminated by this DC voltage controller proposed in this thesis. The
183 Chapter9: Conclusions
DC controller mitigates these overshoots to be below 45V, which is well below the 58V limit allowed in the drafted 42V PowerNet standard. This DC voltage controller also mitigates the overshoots which occur after short term unregulated operation of the ISA where both the battery and the ISA are supplying the on-board demand. The proposed DC controller has eliminated the voltage dip problem during fast acceleration of the engine. In addition it provides fast control for the DC voltage dips caused by large load application. Extensive simulation and experimental studies have clearly demonstrated the above mentioned performance of the proposed DC voltage controller.
Chapter 7 has proposed and tested an improved field weakening implementation which is based on stator voltage control method. The proposed technique included q and d axis voltage regulators with stator frequency decoupling, and an anti-windup technique. A Nonlinear Dynamic Compensator (NDC) was utilised for d axis voltage regulator instead of a linear PI regulator. This chapter has investigated the cause for the oscillation problem occurred with large load application in high speed field weakening- 2 region. The cause of the oscillations was identified as the nonlinearity of the plant seen by the d axis voltage controller. The proposed implementation has provided a solution for this oscillation problem as were clearly demonstrated in this chapter. The overshoots which occurred in the stator voltage during acceleration was eliminated by the stator frequency decoupling of the proposed field weakening control. This has been clearly demonstrated by the simulation and experimental tests. Elimination of these overshoots is important because it reduces the voltage margin available for the current control. The proposed field weakening implantation was employed with anti-windup technique to q axis voltage controller and d axis voltage NDC. The simulation results have clearly demonstrated the effect of anti windup loops in stabilizing field weakening operation in field weakening-1 and field weakening-2 regions. The experimental verifications were carried out for dynamic operation as well as steady- state operation of the ISA. The power generation capability of the ISA with proposed field weakening implementation was tested and compared with conventional 1/ r - field weakening. The test results have demonstrated that proposed implementation allows extracting significant amount of additional DC power in generation mode over wide speed range as illustrated in Section 7.3.6 of Chapter 7.
184 Chapter9: Conclusions
Loss minimised control of the integrated starter alternator was proposed and tested in Chapter 8. The proposed method is based on a loss model which takes into account the inverter losses. A system loss model which combines the loss models of the induction machine and the inverter was developed in terms of dq axes currents in order to facilitate easy incorporation to overall ISA control system. The optimisation of the loss model with simplifications was presented. The proposed simplification for the inverter loss model was justified using computational comparison. The experimental verification of the proposed loss minimised control was carried out for varying load and operational speed conditions. The test results have demonstrated that the proposed loss minimised control allows significant efficiency improvements under light load conditions as shown in Section 8.4 of Chapter 8. This chapter has addressed significant efficiency improvement by means of the loss-minimised control.
9.2 Suggestions for Future Work
In addition to the control design discussed in this thesis, improved hardware design (i.e. electric machine and converter) is important for better performance of the system. Typical induction machine power capability reduces significantly at high speed region. This is due to the large voltage drop across the leakage inductance of the induction machine at higher frequency. An induction machine designed with low leakage inductance may be able to provide better power capability in high speed region. The starting torque requirement of ISA should also be considered in the design of induction machine for ISA application. Power converter design with low loss switches will increase the efficiency of the inverter and thus, the system efficiency. Further work is necessary for improving the efficiency of the electric machine and the inverter to achieve the level of efficiencies expected in future automotive power generation.
Application of direct-flux-vector control for integrated starter alternator has been reported [92]. Field oriented control requires a voltage margin between inverter voltage and the machine back emf in order to control the current at high speed operation. Direct- flux-vector control may not require this voltage margin as the current control is not applicable. This may lead to higher voltage utilization and result in better power capability of the machine in high speeds. Detailed comparison investigation into optimum power generation capability with direct-flux-vector control would be valuable.
185 Chapter9: Conclusions
Low DC bus voltage of 42V poses constraints on the peak stator voltage rating of the electric machine. The maximum possible RMS line-line AC voltage is 29.6V with two level boost converter. This voltage corresponds to the maximum that can be achieved either by space vector modulation or sinusoidal PWM with third harmonic injection. The electric machine peak stator voltage rating needs to be lower than this voltage to provide sufficient margin for current control. For example, the prototype ISA used in this thesis has base stator voltage of 22V. This voltage is unusually low for high power electric machine (i.e. 6kW) and may pose restriction for enhancing performance such as efficiency. It is worth investigating how this low peak voltage level affects the design of the machine. In addition, the low voltage high current power electronics inverter design may require special consideration on selecting the devices in order to reduce the losses of the inverter. If higher DC bus voltage is selected, a DC/DC converter may be used to obtain required low voltage for on-board loads. Alternatively, a totally different power electronics topology providing the higher AC voltage and lower DC voltage (i.e. Buck AC to DC, Boost DC to AC) may be worth investigating.
The converter cost depends on the voltage rating of the power electronics devices for a given kVA rating. The cost of the electric machine may also depend on the level of voltage as a result of machine design constraints. Proper analysis of cost of the system with different voltage levels may also be useful in selecting the desired voltage.
Electrolytic capacitors are well known for the reliability and temperature related problems. Further research into minimizing the size of the DC bus capacitor would be useful as if the required DC bus capacitor size is small, ceramic or polypropylene capacitors may be used instead of the electrolytic capacitors.
Packaging of the system is also of interest as compact design will lead to space saving in the engine bay. High temperature semiconductor switches such as silicon carbide devices may help compact packaging due to reduced cooling requirement in the future. Electric machines with stator mounted power semiconductors, instead of separate power electronics in a separate enclosure would be very suitable for ISA application. New power electronic converter topologies with less passive components would also allow compact packaging of the system.
Most of the present commercially available hybrid electric vehicles use interior permanent magnet (PM) machines due to their attractive features which were discussed
186 Chapter9: Conclusions
in Chapter 2. However, there are many issues, including high system cost, remain unsolved. Research into system cost reduction of interior permanent magnet electric drive would help reducing the price gap between the hybrid electric vehicles and conventional vehicles.
187 References
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194 Appendix A: Induction Machine Modelling
APPENDIX A
DERIVATION OF INDUCTION MACHINE MODEL
A.1 Space-Phasor Representation of Three-phase Quantities
In this space-phaser representation, it is assumed that induction machine has a smooth air gap and is a symmetrical machine with sinusoidal distributed windings. Figure A.1 shows stator of an elementary symmetrical three phase induction machine. The three phase axes and two phase quadrature axes are indicated in this figure.
b
a
c' b'
120 a 120 q
b c
a'
c
d
Figure A.1 Stator of an elementary three phase symmetrical induction machine
Physical quantities that fluctuate with time such as currents, voltages and flux linkage can be represented as complex space-phasors as follows:
2 iitaitait(() ()2 ()) (A.1) s 3 as bs cs where,
195
Appendix A: Induction Machine Modelling
iis qs ji ds , is the space-phasor of stator current where iqs and ids are two-axes components,
itas (), itbs ( ) and itcs ( ) are the instantaneous currents of phase - a , -b and - c respectively,
2 j a is the spatial operator which is given by ae 3 .
By substituting two axes components into (A.1), the matrix version of space-phasor definition can be obtained as follows:
FV 11FV GW1 ias FVi GW qs 2 GW22 ( (((( -*. ( GW GWibs HXi 3 GW ds 33GW GW0 HXics HX22 Equation (A.2) assumes balanced operation of induction machine and hence zero sequence component is not taken into account.
Similarly, space-phasors for other physical quantities of the induction machine can be defined.
A.2 Stator and Rotor Flux Linkage of an Induction Machine
This section discusses the relationship between flux linkage and currents of an induction machine.
Figure A.2 indicates the magnetic axes of a three phase induction machine.
196
Appendix A: Induction Machine Modelling
bs axis
br axis
ibs
ar axis
iar i br as axis
ias
icr
ics
cs axis axis cr
Figure A.2 Magnetic axes of a three phase induction machine
The flux linkage of the stator phase a winding that caused by stator current can be written as follows:
assLi as as L abs i bs L acs i cs (A.3)
Similarly,
bssLi bas as Li bs bs Li bcs cs (A.4)
cssLi cas as Li cbs bs Li cs cs (A.5)
The above equation can be represented in a matrix as follows:
FVF VFV assLL as abs L cas i as GWG WGW GWGbssLLLi bas bs bcs WGW bs (A.6) GWG WGW HXHcssLL cas cbs Li cs XHX cs
197
Appendix A: Induction Machine Modelling
where,
ass is the flux linkage of stator phase a caused by stator currents. Flux linkage of phase b and c are represented in a similar manner.
Las is the self inductance of the stator phase a . Self inductances of phase b and c are represented in similar manner.
Labs is the mutual inductance between stator phase a and phase b . The mutual inductances between other phases can be represented similarly.
The flux linkage of stator winding caused by rotor currents can be written as:
FV FVLLLiFV asrGW as,,, ar as br as cr ar GW GW GWbsrbsarbsbrbscrbrGWLLLi,,,GW (A.7) GW GWGW HXcsrHXLLLi cs,,, ar cs br cs crHX cr where,
asr is the flux linkage of stator phase a caused by rotor currents. Flux linkage of phase b and c are represented in a similar manner.
Las, ar is the mutual inductance between stator phase a and rotor phase a windings. The mutual inductances between other phase windings can be represented similarly.
The flux linkage of rotor winding caused by stator currents can be written as:
FV FVLLLiFV arsarasarbsarcsasGW,,, GW GW GWbrsGWLLLi br,,, as br bs br csGW bs (A.8) GW GWGW HXcrsHXLLLi cr,,, as cr bs cr csHX cs
ars is the flux linkage of rotor phase a caused by stator currents. Flux linkage of rotor phase b and c are represented in a similar manner.
Lar, as is the mutual inductance between rotor phase a and stator phase a windings. The mutual inductances between other phase windings can be represented similarly. 198
Appendix A: Induction Machine Modelling
The flux linkage of rotor winding caused by rotor currents can be written as:
FVF VFV arrLL ar abr L car i ar GWG WGW GWGbrrLLLi bar br bcr WGW br (A.9) GWG WGW HXHcrrLLLi car cbr cr XHX cr where,
arr is the flux linkage of rotor phase a caused by rotor currents. Flux linkage of phase b and c are represented in a similar manner.
Lar is the self inductance of rotor phase a . Self inductances of rotor phase b and c are represented in similar manner.
Labr is the mutual inductance between rotor phase a and phase b . The mutual inductances between other phases can be represented similarly.
As can be seen in (A.6), (A.7), (A.8) and (A.9), number of inductance parameters require for describe the relationship between flux linkages and currents. However, number parameters required can be reduced greatly by considering symmetrical nature of three-phase induction machine.
Self inductance consists of two components, namely, leakage inductance and magnetising inductance as follows:
LLLas asl asm (A.10) where,
Lasl is the leakage component of self inductance.
Lasm is the magnetising component of self inductance
By considering symmetry of the three phase winding,
LLLLLas bs cs sl sm (A.11)
199
Appendix A: Induction Machine Modelling
Also, it can be proved following relationship for mutual inductance between stator phases by assuming sinusoidal mmf distribution.
L LLLsm (A.12) abs bcs cas 2
By substituting (A.11) and (A.12) to matrix given in (A.6),
FVLL sm sm GWLLsl sm FV 22FV ass GWias GWGWLLGW sm LL sm i (A.13) GWbss GW22sl sm GWbs HXGW GWHXGWi css LL cs GWsm sm LL HXGW22sl sm
Using space-phasor definition
2 s ()aa2 ss() 3 ass bss css 2 LL L L [()Li L ism i sm i a sm i Li L i sm i 32222sl as sm as bs cs as sl bs sm bs cs LL aiiLiLi2 ()sm sm ] 22as bs sl cs sm cs
By simplifying above equation
CS3 s DTLLis (A.14) s()ssEUsl2 sm
Following relationships can be obtained for mutual inductances between stator and rotor windings by considering symmetry, winding ratio and angle between rotor and stator windings.
Nr LLLas,,, ar bs br cs cr Lsmcos r (A.15) Ns
200
Appendix A: Induction Machine Modelling
Nr 2 LLLas,,, br bs cr cs ar Lsmcos( r ) (A.16) Ns 3
N r 2 LLLas,,, cr bs ar cs br L smcos( r ) (A.17) Ns 3
By substituting (A.15), (A.16) and (A.17) to matrix given in (A.7),
FV 22 GWcosrr cos( ) cos( r ) FV GW33FVi GWasr N 22 GWar r GW GWbsr Limscos( r ) cosr cos(r ) GWbr (A.18) N GW33 GWHX s GWGWHXi csr 22 cr GWcos( ) cos( ) cos HXGWrr33 r
j r By applying complex number relationship ejcosrr sin to (A.18),
FV IYFVjjjrrr22FV j r jj rrFV asr eaeaee aeaeiar GWN LLGW GW GW r 22jjrr j r j r j r j r GWbsrL ms JZGW a e e aeGW ae e a eGW ibr (A.19) N LL GW s GWjjjrrr22GW jjj rrrGW HXcsr K[HXae a e eHX a e ae eHX icr
By applying space-phasor definition to (A.19) and simplifying,
srN 3 r j r sr() Liesm rr (A.20) Ns 2
The total stator flux linkage is obtained as follows:
ss s ssssr() ()
CS33srN (A.21) r j r DTLLisl smsrr Lie sm EU22Ns
The total rotor flux linkage space-phasor can be obtained by manipulating (A.8) and (A.9) in a similar manner. The total rotor flux linkage space-phasor is given below.
201
Appendix A: Induction Machine Modelling
CSFV2 rr33NNs DTLLiLierr j r (A.22) rr DTrl GWsm rr sm s EU22HXNNss
A.3 Induction Machine Model in Arbitrary Reference Frame
Space-phasors for induction machine quantities can be represented using several reference frames such as stationary (or stator) reference frame, rotor reference frame, arbitrary reference frame and special reference frame. The stationary reference frame is fixed to the stator and is stationary, whereas the rotor reference frame is fixed to rotor and it rotates with the rotor at same angular velocity. The arbitrary reference frame is a general reference frame which rotates at arbitrary angular velocity. Special reference frames are defined to suit with control strategy and it rotates in a way that is required by the control strategy. Most common examples for special reference frames are the synchronous reference frames that are fixed to rotor flux or stator flux of an induction machine. Figure A.3 illustrates stationary, rotor and arbitrary reference frames that are used for space-phasor definition. q g is g r q
r q
d
d d
Figure A.3 Space-phasor representation using stationary, rotor and arbitrary reference frames 202
Appendix A: Induction Machine Modelling
As discussed before, a dynamic model for induction machine can be obtained using the space-phasor representation of physical quantities. The stator voltage of the induction machine in stationary reference frame can be written as follows:
d vRis ss (A.23) s s ssdt
Similarly, the rotor voltage can be given in rotor reference frame.
d vRirr r (A.24) rrr rrdt rr where,
s vs is the stator voltage space-phasor in stationary reference frame,
s is is the stator current space-phasor in stationary reference frame,
s s is the stator flux linkage space-phasor in stationary reference frame,
r vrr is the rotor voltage space-phasor in rotor reference frame,
r irr is the rotor current space-phasor in rotor reference frame,
r rr is the rotor flux linkage space-phasor in rotor reference frame,
Rs and Rr are stator and rotor resistance.
The stator flux linkage per phase of a three-phase induction machine consists of three components. They are flux linkage due to self coupling, flux linkage due to mutual coupling with other two phases of stator windings and flux linkage due to three rotor windings or cage.
The total stator flux linkage is derived in terms of stator and rotor current space phasors in the previous section by considering induction machine geometry and symmetrical construction. The total stator flux linkage space phasor is given by (A.21).
By rewriting (A.21),
203
Appendix A: Induction Machine Modelling
ssrCS33N r j r ssrDTLLisl sm Liesm r (A.25) EU22Ns where,
Lsm is magnetising component of self inductance of a phase. This corresponds to self flux linkage that penetrates the airgap.
Lsl is the stator leakage inductance and its value is same for phase a , b and c because of symmetry of three phase windings and
N r is the rotor turns to stator turns ratio for the induction machine. Ns
Similarly, the rotor flux linkage can be written as follows:
CSFV2 rr33NNs DTLLiLierr j r (A.26) rr DTrl GWsm rr sm s EU22HXNNss
FVN The turns-ratio term (i.e. GWs ) in above equations can be eliminated from the HXNr equations by referring the values of rotor circuit quantities and parameters to the stator circuit similar to transformer theory. Following relationships exists.
FVN s rr GWvvrr r (A.27) HXNr
FVN r rr GWiirr r (A.28) HXNs
FVN s rr GWrr r (A.29) HXNr
204
Appendix A: Induction Machine Modelling
2 FVN s ' GWRrrR (A.30) HXNr
2 FVN s ' GWLLrl rl (A.31) HXNr where,
r r r vr , ir and r are the rotor voltage, current and flux linkage values referred to the stator circuit.
By substituting terns-ratio relationships given in (A.27), (A.28), (A.29) and (A.30) into (A.24), the rotor voltage equation can be written referred to stator circuit as follows:
2 FVNN FVFVN FV N rrrrr' s rd GWvRirrr GWr GW GW HXNNNNdtssrs HXHX HX
d vRirr' r (A.32) rrr dt r
Similarly, by substituting (A.28) into(A.25),
ssr j r ssLLiLiesl m m r (A.33) where,
3 L is the magnetising inductance of the motor and it is given by LL m msm2
Similarly, by substituting (A.28) and (A.29) into (A.26),
FVCS FV22 FV FV NNrr33 NN Ns rr DTLLiLie' rs rj r GWrrDT GWrl GWsm GW sm s HXNNssEU HX22 HX NNN srsHX
rrs ' j r ( (( -*6+ ( rr LLiLierl m m s
205
Appendix A: Induction Machine Modelling
Now, the induction motor model can be completely described by (A.23), (A.32), (A.33) and (A.34). However, in theses equations, the stator and rotor quantities (i.e. voltages, currents and flux) are represented in their own reference frames. In other words, the stator quantities are defined in stationary reference frame whereas the rotor quantities in rotor reference frame. It is advantageous to transform all the induction machine quantities into a single reference frame not only for analysis purposes but also for control design. Hence, space-phasors given in stationary and rotor reference frames are transformed into common arbitrary reference frame shown in Figure A.3, which rotates at angular velocity of g , as follows:
The space phasor transformations for stator current, voltage and flux are given below,
gs j g gs j g gs j g iiess , vvess , sse (A.35)
Similarly, the transformations for rotor current, voltage and flux are given below,
gr j()g r gr j()g r gr j()g r iierr , vverr , rre (A.36) where,
Superscript g represents arbitrary reference frame.
By substituting (A.35) and (A.36) into (A.23), (A.32), (A.33) and (A.34), induction machine model in arbitrary reference frame can be obtained as follows:
d vRig gg j g (A.37) s sgssdt s
d vRig ' gg j() g (A.38) rrrgdt r rr
g gg s LsmiLisr (A.39)
g gg rrsLirm L i (A.40)
206
Appendix A: Induction Machine Modelling
where,
r is the electrical angular velocity of the rotor
Ls is the stator inductance and is given by LLLs sl m ,
' Lr is the stator inductance and is given by LLLrrlm.
The electromagnetic torque of the induction machine can be obtained by considering power flow. The derivation of electromagnetic torque can be found in Section A.5. The expressions for electromagnetic torque are given below.
3 P TLii Im&'gg* (A.41) em 22 m sr
3 P Ti Im&'gg * (A.42) em 22 ss
3 P Ti Im&' gg* (A.43) em 22 rr
L 3 P m gg * Tiem Im&'sr (A.44) 22Lr where,
Im& ' - represents the imaginary component of the complex expression
Equations(A.37), (A.38), (A.39), (A.40) and (A.41) describe the complete induction machine model in arbitrary reference frame.
The above equations that describe the motor model can be rearranged to obtain direct relationship between voltages and current as follows:
Substituting (A.39) into (A.37) and rearranging,
dd vRiLg gg iL () ii ggg j (A.45) s sslmssdt dt srsg
207
Appendix A: Induction Machine Modelling
Substituting (A.40) into (A.38) and rearranging,
dd vRiLiLg ''gg ()() ii gg j g (A.46) rrrrlmdt r dt sr gr r
Equations (A.45) and (A.46) can be used to obtain complex variable dynamic equivalent circuit model in arbitrary reference frame for induction machine as shown in Figure A.4.
g g j ' ' g s L j()grr Rs sl Lrl Rr g g is ir
g g g Lm g vr vs s r
Figure A.4 Complex variable equivalent circuit representation for dynamic model of induction machine in arbitrary reference frame
A.4 State-space Model
The induction machine model in arbitrary reference frame discussed in the previous section can be represented as state-space model by manipulating (A.37), (A.38), (A.39) and (A.40). There are number of state-space models available. Different state-space models can be obtained by selecting different set of state variables. This thesis discusses one state-space model based on stator current and rotor flux state variables.
By rewriting(A.37), (A.38), (A.39) and (A.40) that describes the induction machine model in arbitrary reference frame,
d vRig gg j g (A.47) s sgssdt s
d vRig ' gg j() g (A.48) rrrgdt r rr
208
Appendix A: Induction Machine Modelling
g gg s LsmiLisr (A.49)
g gg rrsLirm L i (A.50)
By (A.49) Lr - (A.50) Lm
g gg2 LLrms rs() LLLi rsm
L g ' ggm s Lis sr (A.51) Lr where,
L2 ' ' m Ls is the stator transient inductance given by LLss(1 ) LLs r
By rearranging (A.50),
g 1 gg iLirrs()m (A.52) Lr
By substituting (A.51) into (A.47),
LL g gggd '' mm gg vRis ssssr()() Li jLi gs sr dt LrrL
LL g ''gg dd mm g g vRjLiLis ()sgssss r jg r (A.53) dt Lrr dt L
By substituting (A.52) into (A.48)
209
Appendix A: Induction Machine Modelling
R' g r ggd g g vLijrrsr()mg ()r r Ldr t
RL' R' d g rmggr g rsijjv()grrr (A.54) dt Lrr L
By substituting (A.54) into (A.53),
LL2 R' L '''d g mm gg r g mg Lisss ()() RRrg2 jLissr j r vs vr dt LrrLLrL r
LLR' d g 11'' gg mmr g g iRjLijs ''()()sgssrr '' vvsr (A.55) dt LssLLLrrsLLLsr where
L2 '' m RRRssr2 Lr
Now, (A.54) and (A.55) can be represented as matrix state-space equation as follows:
FV1 L R' GW()()RjL''m r j FVL FVg ''sgs rFVgg1 m FV d iis GWLLssLrLrssGW''v GWGW LLLGW(A.56) g GW' ' ggGWssr dt GWRL R GW GWv HXr GWrm r HXrrGWHX ()jjgr HX01 HXLLrr where,
CSL2 ' ' m Ls is stator transient inductance and given by LLssDT1 EULLrs
2 CSL '' m RRRssrDT EULr
210
Appendix A: Induction Machine Modelling
A.5 Power Flow and Electromagnetic Torque Production of Induction Machine
The electromagnetic torque production of induction machine can be obtained by considering power flow as follows:
The active power flow to the induction machine from the stator side can be obtained using following equation.
3 Pvi Re * (A.57) es 2 s s where,
Pes is the active power into the induction machine from the stator side.
Proof:
By applying space phasor definition given in (3.3) for stator voltage and current in (A.57),
3 Pvi Re * es 2 ss
32CS 2 7 Re(DTv () t av () t a22 v ())(() t i t ai () t a i ()) t 23EUas bs cs 3as bs cs (A.58) 2 Re (vtavtavtitaitait ( ) ( )22 ( ))( ( ) ( ) ( )) 3 as bs cs as bs cs vtitas() as () vtit bs () bs () vtit cs () cs ()
RHS of (A.58) is the total instantaneous active power into the induction machine from the stator side.
Now, total instantaneous active power into the induction machine can be written as follows:
33 PPPRe viss** Re virr (A.59) eeser22ss rr
211
Appendix A: Induction Machine Modelling
where,
Pe is the total power flow into the induction machine
Per is the rotor side power contribution
By applying the relationships given in (3.17) and (3.18) to (A.59), the power equation in arbitrary reference frame can be obtained.
3 PviviRe ss** rr e 2 ss rr
3 jj j() j () Re vegggggg ie ve grgr ie (A.60) 2 ss r r 3 Re vigg vi gg 2 ss rr
By substituting (A.39)and (A.40) to (A.37) and (A.38) respectively,
dd vRiLiLg gg i g jLijLi g g (A.61) s ssssdt m dt r gsgm s r
dd vRiLiLijg ' gg g () Lijg () Lig (A.62) rrrrdt r m dt s grrgrmr s
By substituting (A.61) and (A.62) to (A.60),
3 dd PRe( Riigg** Li g i g Li g * i g j Liigg** j Liigg ess2 ss sdt s m s dt r gsgss mrs (A.63) dd R' iigg** Li g i g Li g * i g j() Liigg* j ()) Liigg* rrrr rdt r m r dt s grrgrmrr sr
By rearranging (A.63),
3 22 dddd PRe( RiRiLiiLiiLiiLiig' g gg**** gg gg gg esrslrlm2 s r ssdt rr dt rr dt m ss dt (A.64) dd22 Ligg** i j Li g Li gg i j() Li g j Lii gg* ) mgsmgrrrmsrdt s rs dt r sr
212
Appendix A: Induction Machine Modelling
By simplifying
22 2 2 2 311gg' ddd g g gg PesrslrlmRe( RiRiLisr s LiLii r sr 222dt dt dt (A.65) gggg22 * jLigs srsr j()g r Li r jLiirm )
By simplifying further
22 FV2 2 2 33FVgg' d 11g g gg PRiRiessr r GW LiLiLiisls rl r m sr 22HXGWdt HX22 (A.66) 3 22 Re( jLigg j() Li jLiigg* ) 2 gs srsg r r rm r where,
3 FVgg22 Term Ri Ri' represents the coper losses of the stator and rotor, 2 HXGWsrsr
FV22 2 11gggg Term GWLislsrsr Li rl Li m i represents the energy stored in stator HX22 leakage, rotor leakage and magnetizing inductances.
FV22 3 gg gg* Term GWRe( jLigs srs j()g r Li r jLiirm r) represents the mechanical HX2 power component of the motor.
Therefore, the power transform into shaft power is given by,
3 22 PjLijLijLiiRe( gg() gg* ) (A.67) em 2 g s srsg r r r m r
The first two terms of (A.67) are imaginary. Therefore,
33 PjLiiLiiRe(gg** ) Im(gg ) (A.68) em 22r m sr r m sr
Now the electromagnetic torque of induction machine can be written as follows:
213
Appendix A: Induction Machine Modelling
P T em em rm p P em (A.69) 2 r 3 p LiiIm(gg* ) 22 m sr
By substituting (A.39) to (A.69),
3 p TLii Im(gg* ) em22 m sr * CS gg Li 3 p g sss Lim Im( s DT) (A.70) 22 EULm 3 p Im(i gg * ) 22 ss
By substituting (A.40) to (A.69),
3 p CS gg Li TL Im(DTrrr ig* ) em22 m L r EUm (A.71) 3 p Im( ggi * ) 22 rr
By substituting (A.40) to (A.69),
* 3 p CS gg Li TLi Im( g DTrsm ) em22 m s L EUr (A.72) L 3 p m gg * Im(isr ) 22Lr
214
Appendix A: Induction Machine Modelling
A.6 Matrix Representation of Reference Frame Transformation
The complex number relationship for reference frame transformation is given by (A.35). This relationship for stator current can be rewritten as follows:
gs j g iiess (A.73)
By substituting definition of stator phasor given in (A.1) into (A.73),
2 j iitaitaiteg (() ()2 ()) g s 3 as bs cs
24 2 j jj()gg () ig (() i teg i () te33 i () te ) s 3 as bs cs
22CS 2 ijiitgg( ( ) cos( ) j sin( ) it ( )DT cos( ) j sin( ) qs ds33 as g g bs EUg g 3 (A.74) CS44 it()DT cos( )j sin( ) ) cs EUg 33g
By separating real and imaginary components from (A.74),
FV24FV cos( ) cos( ) cos( ) ias FVi g 2 GWgg g GW qs GW33 GWg GWibs (A.75) HXi 3 GW24 ds sin( ) sin( ) sin( ) HXGWi HXGWgg33 g cs
A.7 Induction Machine Loss Modelling
This section describes the loss modelling of induction machine including core loss of stator, Rcs and rotor, Rcr in field oriented synchronous reference frame.
215
Appendix A: Induction Machine Modelling
e ' e ' eds L ()erdr Rs sl Lrl Rr e e iqs iqr
e Rcs Lm Rcr vqs
e e ' ' eqs L ()erqr Rs sl Lrl Rr e e ids idr
e Rcs Lm Rcr vds
Figure A.5 Synchronous reference frame IM model with core loss resistances
By rearranging (3.17),
ee e e eee e qsLi s qs L m i qr L sl i qs L m() i qs i qr L sl i qs L m i qm (A.76)
e where, iqm is the q axis magnetising current in synchronous reference frame
Similarly from (3.18),
eee dsLi sl ds Li m dm (A.77)
e where, idm is the d axis magnetising current in synchronous reference frame
By assuming the leakage flux is small compared magnetising flux (i.e. neglecting Lsl ), equations (A.76) and (A.77) can be written as follows:
ee ee qsLi m qm and dsLi m dm (A.78)
216
Appendix A: Induction Machine Modelling
Similarly, the rotor flux in qd axes can be written approximately by neglecting Lrl as follows:
ee ee qrL mi qm and drLi m dm (A.79)
With the above assumptions, the model given in Figure A.5 can be simplified as shown in Figure A.6.
Lie e R emdm ()ermdmL i s _ + _ + e e iqs i qm ' RcrR r e v Rcs Lm ' qs RcrR r
Lie e emqm ()ermqmL i Rs _ + + _ ie e ds idm RR' e cr r Rcs Lm vds ' RRcr r
Figure A.6 Simplified synchronous reference frame IM model with core loss resistances
By applying conditions for rotor flux reference frame given in (5.1) to (A.79),
ee qrLi m qm 0 ie 0 qm (A.80)
In steady state, the rotor flux in rotor flux oriented reference frame is given by ee drLi m ds . By applying this condition to (A.79),
217
Appendix A: Induction Machine Modelling
eee drLi m ds Li m dm iiee ds dm (A.81)
Also, the rotor copper loss resistance Rcr is very large compare to the rotor
' resistance, Rr . Therefore,
RR' cr r R' RR ' r cr r (A.82)
In steady state, the current through the magnetising inductance is DC in rotor flux oriented reference frame.
By applying above conditions to model given in Figure A.6, the model can be further simplified as shown in Figure A.7.
Lie e R emds ()ermdsLi s _ + _ + e iqs e ' e iqm 0 Rcs Rr vqs
e e iqms iqmr
Rs e ids
e vds
Figure A.7 Simplified IM model with core loss resistances in rotor flux oriented reference frame 218
Appendix A: Induction Machine Modelling
For above simplified model, following equations can be written.
Lie eeemds iiqms qs (A.83) Rcs
() Lie e ermds iqmr ' (A.84) Rr
ee iiqmr qms (A.85)
By manipulating (A.83), (A.84) and (A.85), following relationship can be obtained.
RL iieecs m i e (A.86) qms ''qs r ds RRrcs RR rcs
The stator copper loss, Pcus can be written as follows:
3 PRii()ee22 (A.87) cus2 s qs ds
The core losses, Pfe can be written as follows:
3 PRii()ee2 (A.88) fe2 cs qs qms
By substituting (A.86) into (A.88),
CS2 3 RL PRiDTeecs i m i e fe csDT qs ''qs r ds 2 EURRrcs RR rcs
R 2 3 cs ' ee Rirqs L m rds i (A.89) 2 ' 2 RRrcs R 3 cs '2ee 2 2 2 2 ' ee RriLi qs m r ds2 LRii m r r ds qs 2 ' 2 RRrcs
The rotor copper loss Pcur can be written as follows:
3 PRi '2e (A.90) cur2 r qmr
By substituting (A.85) and (A.86) into (A.90) and rearranging,
219
Appendix A: Induction Machine Modelling
R' 3 r 22ee 222 ee PRcur csi qsL m ri ds2L mR cs ri dsi qs (A.91) 2 ' 2 RRrcs
The total losses of the induction machine can be written as follows:
PPPPtot cus cur fe CSCS RR'2 RR' 2 RL2 2 RL' 2 2 33DTcs r r csee22 DT cs m r r m r PRtot si qs R si ds 22DT ''22 DT '' 22 EURRrcsrcs RR EURRrcsrcs RR (A.92) CSCS 33RR'2L 2 DTDTRiRics r ee22 m r DTDTsq ''ssd s 22EUEURRrcs RRrcs e 2 e 2 RiQqs RDrds()i
Where,
CS CS 3 RR' 3 L22 RRDTcs r and RR() DTmr QsDT ' DrDT s ' 2 EURRrcs 2 EURRrcs
220
Appendix B: Model of Internal Combustion Engine
APPENDIX B
MODEL OF INTERNAL COMBUSTION ENGINE
The MATLAB/Simulink model used for computer modelling of internal combustion engine is discussed in Chapter 3 is given in this appendix. This appendix is consists of M-file and a Simulink block diagram. The M-file is used for initializing of the parameters of Simulink model. This initialization file includes geometrical dimensions of piston and crankshaft arrangement, varies empirical coefficient related to various friction components and parameter related to viscosity of lubrication oil.
The attached Simulink block diagram uses the parameters initialized in the M-file to simulate the engine torque during starting of the engine cylinder pressure torque and various frictional torque components.
221 E:\chathura\my_document_02_Oct_2006\Engine_starting_modelling\initialization_engine_model.m Page 1 August 16, 2008 5:01:08 PM % this m file initialize ICE engine parameters for 1753cm3 4-cylinder inline 8 valve engine %------parameters necessery to calculate the cylinder pressu re Vd = 1753e-6; Rcrank = 41e-3 ; % radius of cranckshaft(m) Lcon = 130e-3 ; % conecting rod length(m) B = 82.5e-3 ; % diameter of the piston(m) CR = 14 ; % compression ratio Patm = 101.295e3; % atmospheric pressure n = 1.2 ; %polytropic index of the gas mixture
Ap = pi*(B/2)^2; % area of the piston
V_ivc = Rcrank*Ap*((2/(CR-1)) + (1-cos(pi)) + (Lcon/Rcrank) - sqrt((Lcon/Rcra nk)^2 - (sin(pi))^2)); % this is the valume of the cylinder after the intake stroke(i.e at beginig of compression stroke
P_ivc = Patm ; % the intake valve is open therefore cylinder pressure should be equal to the manifod pressure. by assuming manifold pressure is equal to atmospheric pressure P1_ivc = 1 atm
%------Parameters for calculating the friction mean effectiv e pressure------B1 = 82.5 ;% diameter of the piston(mm) S = 82 ; % stroke (mm) Db = 54 ; %crankshaft bearing diameter (mm) Lb = 21.6 ; %crankshaft bearing length (mm) nb = 5 ; % number of crankshaft bearings nc = 4 ; %number of cylinders
Dbp = 49 ; %big-end bearing diameter (mm)of connecting rod Lbp = 21.4 ; %big-end bearing length (mm)of connecting rod nbp = 4 ; % number of big-end connecting rod bearings nbv = 5 ; %number of camshaft bearings nv = 8 ;% no of valves Lv = 5; % maximum valve lift theta1 = 1432.3; %vogel parameter for multigrade oils SAE 10W/30 theta2 =132.9; %vogel parameter for multigrade oils SAE 10W/30 k = 0.0209e-3; %vogel parameter for multigrade oils SAE 10W/30 Temp_ref = 90; Temp = -20; 222 E:\chathura\my_document_02_Oct_2006\Engine_starting_modelling\initialization_engine_model.m Page 2 August 16, 2008 5:01:08 PM visc_ref = k*exp(theta1/(theta2 + Temp_ref)) ; % dynamic viscocity of oil un der fully warm conditions (at 90c) visc = k*exp(theta1/(theta2 + Temp)); % dynamic viscocity of oil under test conditions
%------cranshaft------index_cs = 0.4 ; % index of viscocity ratio
Ccb = 0.0205 ;% coeficient for main bearing friction
Ccs = 93600 ; % coeficient for oil seal friction
%------piston assembly------index_cp = 0.3 ; % index for viscosity ratio
Cpb = 0.0202 ; % coeficient for big end baring
Cps = 12 ;% coeficient for piston skirting
Cpr = 2308 ; % coeficient for piston rings
% Ccb*(((u[1])^0.6)*(Db^3)*Lb*nb/(S*nc*B1^2))*(visc/visc_ref)^index_cs + Ccs* (Db/(S*nc*B1^2)) % % (Cpb*(((u[1])^0.6)*(Dbp^3)*Lbp*nbp/(S*nc*B1^2)) + Cps*(((u[2])^0.5)/B1) + C pr*(((u[2])^0.5)/B1^2))*(visc/visc_ref)^index_cp
%------valve assembly------
Cvb = 6720 ; % coeficient for camshaft bearing contribution Cvh = 0.5 ; % coeficient for ocilating hydrodynamic lubrication friction cont ribution Cvm = 10.7 ; % coeficient for oscilating mix Cvs = 1.2 ; % coeficient for contribution from front oil seal of camshaft Cvf = 200 ; % coeficient for flat cam follower index_val = 0.7 ;% viscocity coeficient
%------auxiliary components------
%------oil pump------
223 E:\chathura\my_document_02_Oct_2006\Engine_starting_modelling\initialization_engine_model.m Page 3 August 16, 2008 5:01:08 PM alpha_o = 1.28 ; % coeficient of polynomial beta_o = 0.0126 ;% coeficient of polynomial gama_o = -8.4e-7 ;% coeficient of polynomial index_o = 0.3 ; % viscosity coeficient
%------fuel injectors------
alpha_fi = 1.72 ; % coeficient of polynomial beta_fi = 0.00069 ;% coeficient of polynomial gama_fi = 1.2e-7 ;% coeficient of polynomial index_fi = 0.5 ; % viscosity coeficient
% ------engine moment of inertia------
Jf = 0.12 ; % fixed moment of inertia mrec = 1.359 ; % mass of piston and connecting rods
% 2*mrec*Rcrank^2 + ((mrec*Rcrank^4)/(2*Lcon^2)) - (2*(cos(2*u[1]))*mrec*Rc rank^2) - ((mrec*(cos(4*u[1]))*Rcrank^4)/(2*Lcon^2)) % % (4*mrec*(sin(2*u[1]))*Rcrank^2) + ((2*mrec*(sin(4*u[1]))*Rcrank^4)/(Lcon^2) )
% ------moment of inertia of the experimental setup------
Jabb = 0.049 ; % moment of inertia of ABB motor Jsi = 0.109 ; % moment of inertia of the siemens motor Jcoup = 0.004*2; % moment of inertia of the coupling Jts = 57.2e-6 ; % moment of inertia of the torque sensor
Jex = Jabb + Jsi + Jcoup + Jts;
224 Ramp
position Torque
T cylinder1 z-1 Discrete-Time Integrator position Torque
cylinder2
Scope2
position Torque Engine_torque 150 ISA_torque Tisa Drive_torque cylinder3 speed Scope3 theta
synthezising_J
position Torque
cylinder4 Scope1
Gain1 2*pi/60
0 10 speed total_friction_torque speed_rpm Display2
engine_friction
225 1 Rcrank*Ap*((2/(CR-1)) + (1-cos(u[1]) + (Lcon/Rcrank) - sqrt((Lcon/Rcrank)^2 - (sin(u[1]))^2))) P_ivc*(V_ivc/u[1])^n position volume_variation pressure
Scope2
0.5 sin(u[1] - pi/2)
Gain1 pressure1 Switch
Patm
Constant
cylinder_pressure
T 1 crank angle Torque
indicated_torque_sub
226 1 cylinder_pressure
Rcrank*Ap*(u[1] - Patm)*(sin(u[2]))*(1 + (cos(u[2]))/sqrt((Lcon/Rcrank)^2 - (sin(u[2]))^2)) 1 T indicated_torque
2 crank angle
227 1 Rcrank*Ap*((2/(CR-1)) + (1-cos(u[1]) + (Lcon/Rcrank) - sqrt((Lcon/Rcrank)^2 - (sin(u[1]))^2))) position volume_variation
pipi P_ivc*(V_ivc/u[1])^n
pressure_polytropic
0.5 sin(u[1] - pi/2)
Gain1 condition Switch
Patm
atmospheric_presure
cylinder_pressure
T 1 crank angle Torque
indicated_torque_sub
228 1 cylinder_pressure
Rcrank*Ap*(u[1] - Patm)*(sin(u[2]))*(1 + (cos(u[2]))/sqrt((Lcon/Rcrank)^2 - (sin(u[2]))^2)) 1 T indicated_torque
2 crank angle
229 1 Rcrank*Ap*((2/(CR-1)) + (1-cos(u[1]) + (Lcon/Rcrank) - sqrt((Lcon/Rcrank)^2 - (sin(u[1]))^2))) position volume_variation
2*pi2pi P_ivc*(V_ivc/u[1])^n
pressure_polytropic
0.5 sin(u[1] - pi/2)
Gain1 condition Switch
Patm
atmospheric_presure
cylinder_pressure
T 1 crank angle Torque
indicated_torque_sub
230 1 cylinder_pressure
Rcrank*Ap*(u[1] - Patm)*(sin(u[2]))*(1 + (cos(u[2]))/sqrt((Lcon/Rcrank)^2 - (sin(u[2]))^2)) 1 T indicated_torque
2 crank angle
231 1 Rcrank*Ap*((2/(CR-1)) + (1-cos(u[1]) + (Lcon/Rcrank) - sqrt((Lcon/Rcrank)^2 - (sin(u[1]))^2))) position volume_variation
3*pi3pi P_ivc*(V_ivc/u[1])^n
pressure_polytropic
0.5 sin(u[1] - pi/2)
Gain1 condition Switch
Patm
atmospheric_presure
cylinder_pressure
T 1 crank angle Torque
indicated_torque_sub
232 1 cylinder_pressure
Rcrank*Ap*(u[1] - Patm)*(sin(u[2]))*(1 + (cos(u[2]))/sqrt((Lcon/Rcrank)^2 - (sin(u[2]))^2)) 1 T indicated_torque
2 crank angle
233 300
speed_rpm 0
Display Vd*1000/(2*pi*2) 1 1 Ccb*(((u[1])^0.6)*(Db^3)*Lb*nb/(S*nc*B1^2))*(visc/visc_ref)^index_cs + Ccs*(Db/(S*nc*B1^2)) total_friction_torque speed fmep_crankshaft Gain1
0
0 Display5 (Cpb*(((u[1])^0.6)*(Dbp^3)*Lbp*nbp/(S*nc*B1^2)) + Cps*(((u[2])^0.5)/B1) + Cpr*(((u[2])^0.5)/B1^2))*(visc/visc_ref)^index_cp
82/30 fmep_pistonassemply Display1
Gain
(Cvb*(((u[1])^0.6)*nbv/(S*nc*B1^2)) + Cvh*(((u[1])^0.5)*(Lv^1.5)*nv/(S*nc*B1)))*(visc/visc_ref)^index_val + Cvm*(2 + (10/(5 + visc*u[1])))*Lv*nv/(S*nc) + Cvs + Cvf*(2 + (10/(5 + visc*u[1])))*nv/(S*nc) 0
Display2 fmep_valve_assemply
0 alpha_o + (beta_o*u[1] + gama_o*(u[1])^2)*(visc/visc_ref)^index_o Display3 fmep_oilpump
0 alpha_fi + (beta_fi*u[1] + gama_fi*(u[1])^2)*(visc/visc_ref)^index_fi
fmep_FIE Display4
234 1 Jex Engine_torque Jex
2 ISA_torque
Jex
Jex_c
Jf 1 Jf Product Drive_torque Product1 2*mrec*Rcrank^2 + ((mrec*Rcrank^4)/(2*Lcon^2)) - (2*(cos(2*u[1]))*mrec*Rcrank^2) - ((mrec*(cos(4*u[1]))*Rcrank^4)/(2*Lcon^2))
Jt
4 theta ((u[2]^2)/2)*(4*mrec*(sin(2*u[1]))*Rcrank^2) + ((2*mrec*(sin(4*u[1]))*Rcrank^4)/(Lcon^2)) Jex 3 speed reciprocating Jex1
235 Appendix C: Experimental setup
APPENDIX C
EXPERIMENTAL SETUP
C.1 Overview of Experimental Setup
Figure C.1 shows the experimental setup used for the experimental studies presented in this thesis. The prototype ISA consists of low voltage induction machine, three-phase PWM inverter and three 12V batteries connected in series to form 36V battery. The DC bus of the ISA system is connected to load that consists of PWM DC/DC converter to form a variable load. The prototype ISA is mechanically coupled to a drive motor which is supplied by an inverter. The DC bus of drive inverter is connected to a chopper with resistor to dissipate energy when DC bus voltage exceeds a set limit. The drive motor drive system is supplied with 415V 3phase. Control algorithms for ISA system and variable load on 42V DC are implemented in dSPACE DS1104 controller board. The control algorithms required for drive motor and DC chopper are implemented in dSPACE DS1102 controller board. A PC with Pentium III 800MHz is hosted both dSPACE controller boards. The presumed advantage of this system is the rapid development speed of programs through the use of the dSPACE Real-time Interface (RTI). This feature enabled modelled systems developed in Matlab/Simulink software to be automatically encoded into C-code for downloading to the DS1104 and DS1102 DSP systems.
236 Appendix C: Experimental setup
Figure C.1 Experimental setup with proposed ISA
237 Appendix C: Experimental Setup
C.2 Equipment Details
Prototype Induction Machine ISA: Stator core and rotor of an industrial induction machine with new low voltage stator winding was applied for the prototype experimental setup. Following are the ratings of the induction machine.
Power: 5.5kW at 50Hz motoring
Rated Voltage: 22V at 50Hz
Rated Current: 220A
Poles: 6
Prototype ISA Inverter:
IPM Module: SEMIKRON SKiiP 613GD123-3DUL
Rating: IGBT 600A, 900V
Current and voltage sensors external to the module were used
DC Bus capacitors: 10 x 10mF (the capacitance value of the DC bus was changed by changing number of capacitors during the experiments)
Battery
Manufacturer: YUASA
Model: NP65-12 (3 x 12V in series) dSPACE 1104 Controller Board
Manufactuer: dSPACE GmbH
Technologiepark 25
33100 Pederborn
Germany
Processor: Main processor operating at 250MHz, 64-bit floating point.
Memory: Global memory: 32 MB SDRAM and flash memory 8MB.
238 Appendix C: Experimental Setup
ADC: The board has 4 multiplexed channels with 16-bit resolution; 28second conversion time. It also has 4 A/D channel with 12-bit resolution and 800 9second conversion time.
Slave DSP subsystem: Texas Instruments TMS320F240 DSP @ 20MHx clock frequency. It has 1x 3 phase PWM output and 4x1 phase PWM outputs.
C.3 Photographs of the Experimental Setup
Figure C.2 Overview of experimental setup
239 Appendix C: Experimental Setup
Figure C.3 ISA induction machine and drive motor
Figure C.4 PC which hosts dSPACE1104 and 1102 control boards
240 Appendix C: Experimental Setup
Figure C.5 PC which hosts dSPACE1104 and 1102 control boards
Figure C.6 36V battery
241 Appendix C: Experimental Setup
Figure C.7 Prototype ISA inverter, load controller and drive inverter
Figure C.8 Some control and power cable connections
242 Appendix D: List of Publications
APPENDIX D
LIST OF PUBLICATIONS
Refereed journal and magazine publications
[1] C. P. Mudannayake and M. F. Rahman, “Parameter Determination of an Induction Machine for Integrated Starter Alternator”, Australian Journal of Electrical and Electronics Engineering, vol. 3, issue 2, pp. 91-100. 2007
[2] C. P. Mudannayake and M. F. Rahman, “Control Design of an Induction Machine Based 42V Integrated Starter Alternator”, IEEE Industry Applications Magazine, No. 4: July – August, 2009
Refereed conference publications
[3] C. P. Mudannayake and M.F. Rahman, “Control Design of an Induction Machine Based Integrated Starter Alternator for 42V PowerNet” Proceedings of the IEEE Industry Applications Society, IAS2006, 41st Annual Meeting, October 8 - 12, 2006, Tampa, Florida USA, pp.1585-1592
[4] C. P. Mudannayake, M.F. Rahman and B. Karanayil, “Sensorless Induction Machine Based Integrated Starter Alternator with Loss Minimization” Proceedings of the IEEE Industry Applications Society, IAS2005, 40th Annual Meeting, October 2 - 6, 2005, Kowloon, Hong Kong, pp.1058-1065
[5] C. P. Mudannayake and M. F. Rahman, “Parameter Determination of Induction Machine Integrated Starter Alternator”, Proceeding of Australasian Universities Power Engineering Conference (AUPEC 2005), Hobart, Australia, September 25 – 28, 2005.
[6] C. P. Mudannayake and M.F. Rahman, “Sensorless Induction Machine Based Integrated Starter Alternator for the 42V PowerNet for Automobiles” Proceedings of the IEEE Conference on Vehicle Power and Propulsion, VPP2005, September 7 - 9, 2005, Chicago, Illinois USA, pp. 479- 486
243 Appendix D: List of Publications
[7] C. P. Mudannayake and M.F. Rahman, “A Study of the Transient Behaviour of a 42V Prototype Integrated Starter Alternator for Sudden Load Change” Proceeding of the 30th Annual Conference of the IEEE Industrial Electronic Society (IECON 2004), Busan, Korea, November 2 – 6, 2004, pp.1694- 1699
[8] C. P. Mudannayake and M.F. Rahman, “The Dynamic Characteristics of a 42V Prototype Integrated Starter Alternator during Sudden Load Changes” Proceedings of the IEEE Symposium of Vehicular Power and Propulsion, VPP2004, October, 2004, Paris, France
[9] C. P Mudannayake and M. F. Rahman, “Load Dump Transient Behaviour of a 42V prototype Integrated Starter Alternator”, Proceeding of Australasian Universities Power Engineering Conference (AUPEC 2004), Brisbane, Australia, 26 Sept. - 29 Sept. 2004. ISBN 1-864-99775-3 (CD ROM).
[10] C. P. Mudannayake and M.F. Rahman, “A Matlab/Simulink Model for a Prototype Integrated Starter Alternator for Automobiles” Proceeding of the 4th International IEEE Power Electronics and Motion Control Conference (IPEMC2004), Xian, China, August 14 – 16, 2004, pp. 1679- 1684
[11] C. P. Mudannayake and M.F. Rahman, “An Integrated Starter Alternator for the 42V PowerNet” Proceedings of the 5th International IEEE Power Electronics and Drive Systems, PEDS2003, Singapore, November 17 - 20, 2003, pp. 648- 653
[12] C. P. Mudannayake and M. F. Rahman, “An Integrated Starter Alternator for the 42V PowerNet”, Proceeding of Australasian Universities Power Engineering Conference (AUPEC 2003), Christchurch, New Zealand, 28 Sept. - 1 Oct. 2003. ISBN 0-473-09867-9 (CD ROM).
[13] T. Manmek, C. P. Mudannayake, “Real-Time Implementation of Voltage Dip Mitigation using D-STATCOM with Fast Extraction of Instantaneous Symmetrical Components,” Proceeding of the 7th International IEEE Conference on Power electronics and Drive systems (PEDS07), Bangkok, Thailand, November 27-30, 2007, pp. 568-575.
[14] T. Manmek, C. P. Mudannayake, and C. Grantham, “Voltage Dip Detection Based on an Efficient Least Squares Algorithm for D-STATCOM Application,”
244 Appendix D: List of Publications
Proceeding of the 2006 CES/IEEE – PELS International Power Electronics and Motion Control Conference (IPEMC06), Shanghai, P.R. China on the 13th Aug – 16th Aug 2006.
[15] T. Manmek, C. P. Mudannayake, and C. Grantham, “Robust Signal Processing System for Identification of Harmonics in an Active Power Filter Application,” Proceeding of the 6th International IEEE Conference on Power electronics and Drive systems (PEDS05), Kuala Lumpur, Malaysia, 28th Nov - 1st Dec 2005, pp.313-318.
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