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'Transistor Driver for Valve Amplifiers 36 Wireless World, April 1976 'Transistor driver for valve amplifiers Design for Williamson and other output stages by Seth Berglund Lunds University, Sweden There are certainly a lot of valve audio driver. On the contrary, by starting with amplifiers still in use, and many of them a bandwidth of lMHz, the high-fre­ have an inherent quality of perfor­ quency response can be exactly formed mance that makes it reasonable to give up to this frequency, using simple them a thorough repair, with or without operational amplifier techniques, and so an accompanying modernization. The it can easily be changed to suit diffennt work needed for repair may tend to output transformers: The output vol­ grow, however, since it may not be tage of the driver is sufficient even for sufficient to replace valves and a few large output tubes such as the KT88; electrolytic capacitors. A general they are assumed to work in class A or degradation of components may have AB in ti1e design that follows. taken place, and in nearly all instances of modernization it should be advanta­ Output valve biasing Fig. constant voltage is in this geous to replace the rectifying valve by 1. A Vgo When direct coupling to the output circuit added to a normal cathode bias. silicon or maybe selenium rectifiers. So valves is used, the grids can still be kept there may be some doubt as to what is at zero potential for the quiescent pOint, really needed and what is worthwhile. with a normal cathode bias for class A For those who are interested in giving or AB operation. But this is not neces­ their valve amplifier a positive moder­ sary andin my opinion not at all the best nization that will result in obvious Cg way. Let us therefore look at other ways improvements, a description is here of biasing. For the sake of simplicity, 0-1....... ---- -+ given of a transistor driving amplifier single valve biasing is discussed first, that can replace the voltage amplifying and the valves shown as triodes with Rg stages of many existing power ampli­ the usual assumption of zero grid fiers. The Williamson amplifierl has current, i.e. anode and cathode currents been chosen as a typical example for the are identical. If thus la is the anode discussion that follows, because it is a current of a triode and Rh the outer well-known design. Other amplifier cathode resistance, the negative grid designs that have been used for compa­ voltage with a normal cathode bias is rison are those designs by MullarQ.2 and Vgk = laRk' by It is possible, although not often used GEC3• Fig. 2. Further development of the 'the original idea was to design an . in practise, to modify the influence of the circuit in Fig. 1 by means of a gain amplifier with a bandwidth sufficiently anode current on this bias voltage by function A. in excess of the output transformer the addition of a constant voltage to the bandwidth, so that the only phase shift circuit, either in series with the cathode to take account of should be that of the allowable without instability. This was or, normally with less effort, in the grid transformer. A d.c. amplifier with a found to be the case for several output circuit, shown as the voltage V.o in Fig. bandwidth of about IMHz was thought transformers, at least with a resistive 1. The grid bias voltage is now Vgh to be sufficient. Direct coupling from load. With the Williamson transformer = Vgo - laRk' the input stage to the signal grids of the and output stage according to the It is important to note that Vgo may be output valves leaves the output trans­ original design, an essentially flat positive as indicated in the figure, or former as the only cause of phase shift amplitude-frequency response was negative. In the first case a larger at the low frequency end, and the shift obtained up to IMHz, and there was resistance Rk is required than for simple tends to only 90°. So there are no some stabilitv margain. cathode bias, which makes the gria problems of instability from negative If a loudspeaker or some load of a voltage more dependent on the anode feedback' at the low frequency end, complex nature is inserted, the phase current, i.e. there is a better stabiliza­ provided that the usual precautions as shift tends to become too large, and the tion of the quiescent point. In the case to supply line filtering are taken. only way to stability is then to reduce of negative polarity for VgO' the grid At the high frequency end of the the closed-loop gain. So extended voltage becomes less dependent on the transformer passband there is usually bandwidth is no radical solution for anode current, as Rk must be dimin­ one main resonant frequency, often at stability at the high frequency end in ished. For growing negative values of about 100kHz, around which the phase the same way as direct coupling is for Vgo' it becomes in the limit equal to the shift passes 90° by a considerable low frequencies. And after all, the aim deSired grid voltage. Then Rh must go to amount but does not reach 180°. It was should not be amplification up to radio zero and the result is a constant grid thought therefore that with a band­ frequencies, but an l.f. amplifier with a Voltage. width of at least IMHz for the driver, defined upper frequency limit. This does A grid bias that has.exactly the same the normal amount of 20dB frequency­ not mean that it is a wasted effort to dependence on the combination of a independent feedback should be start with a large bandwidth for the constant voltage and the anode current WirelessWorld, April 1976 37 as that of Fig. I, but with improved Fig. 3. Circuit principles for a complete means for the choice, can be obtained push-pull amplifier, but with output by a circuit as shown in Fig. 2. With the transformer omitted. The inputs are for notations according to this figure, and signal input and feedback. provided that the operational amplifier of voltage gain A has zero offset, the, constant part of the grid bias is VgO A VI' and the total grid bias becomes Vgk = A VI - (1 + A)IaRk In this circuit Rk can be a small resistance, which is an advantage for large output tubes where the power dissipated in Rk for a normal bias may be considerable. Most important is, however, the ease of adjustment to a desired bias. The bypass capacitor Ck has retained its function, and the time constant RkCk is chosen as for normal cathode bias. oV However, if Rk is small, so that it causes only negligable feedback by itself, the bias time-constant may be introduced by a separate RC-link, either before or after theamplifier. In the foregoing figures the bias and signal voltages have been mixed in the usual way by a grid resistor R and a coupling capacitor C . If a fulf signal g feedback from Rk is wanted, corre­ sponding to an unbypassed cathode resistance in Fig. I, some other type of mixing circuit is needed. This also holds, if direct coupling of the signal to the valve grid is used. Arrangement of amplifier Resistance R2 is selectable for choice of differential stages reduce hum. Only the To explain the main features of the voltage gain. negative high tension voltage needs a complete push-pull amplifier, its layout As the amplified part of the bias is a certain stabilization. is first shown by the simplified circuit of common-mode one, it corresponds to a The input common-emitter long­ Fig. 3. The circuit comprises three common cathode resistance with the tailed pair of Fig. 3 is a dual n-p-n differential stages, namely a signal value RkAl2, and the time constant of transistor Trl' and it is completed by an input stage, a biasing stage for the the RC-link is RCI2. The constant part n-channel dual f.e.t. Tr2' the two sensing of the currents in the output of the grid bias is simply an offset transistors of which are used as input valves, and between these a mixing and voltage, effected by an adjustment of source followers. This makes the ampli­ amplifying stage that drives the valves. the potentiometer R2, which is therefore fier compatible with valve amplifiers It is a symmetrical circuit throughout found to have the double function of with regard to input impedance as well for the input signals, and the necessary determining the gain by its resistance as to independency of the characteris­ d.c. balance is obtained at the emitter value and the constant voltage by its tics of the driving source. So all that is side of the input stage, in the figure by adjustment. required of the preamplifier is that it means of the potentiometer RJ• Another The resistances of Rk may be so small shall give sufficient voltage. important feature is that the differential that their direct influence on the valve The d.c. balancing potentiometer R4J, stages are all supplied by a current bias becomes negligible. They cause a a I5-turn trimming potentiometer, has source at the emitter side, instead of small lowering of the effective valve been moved away from the main signal just by a common emItter resistor. A transconductances. path into the f.e.t. source circuit, where high common-mode rejection ratio is Because the collector resistances of it gives a smooth adjustment of the thereby obtained, which means that the the mixing stage become fairly large, differential balance.
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