A Polar Modulator Transmitter for GSM/EDGE Michael R

2190 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004 A Polar Modulator Transmitter for GSM/EDGE Michael R. Elliott, Tony Montalvo, Brad P. Jeffries, Frank Murden, Jon Strange, Member, IEEE, Allen Hill, Sanjay Nandipaku, Member, IEEE, and Johannes Harrebek

Abstract—This 0.5- m SiGe BiCMOS polar modulator IC adds where is the AM and is the PM. This is the input signal EDGE transmit capability to a GSM transceiver IC without any to both the PM and AM signal paths. RF filters. Envelope information is extracted from the transmit IF The PM path limits the IF signal IF to remove the AM. and applied to the phase-modulated carrier in an RF variable gain amplifier which follows the integrated transmit VCO. The dual- The limited signal is then used as the reference input to a trans- band IC supports all four GSM bands. In EDGE mode, the IC pro- lational loop PLL. The PLL locks the VCO to the transmit fre- duces more than 1 dBm of output power with more than 6 dB of quency, transfers the PM of the reference input onto the VCO, margin to the transmit spectrum mask and less than 3% rms phase and acts as a tunable high-Q filter for input noise contributors in error. In GSM mode, more than 7 dBm of output power is produced the transmitter. The resulting VCO output from the PLL is then with noise in the receive band less than 164 dBc/Hz. VCO (2) I. INTRODUCTION HE EDGE standard [1] is an enhancement to GSM The IF signal IF is also input to the AM path where the AM T designed to accommodate higher data rate communica- is extracted by mixing the IF signal with an amplitude-limited tions. In order to achieve this, the modulation ( -shifted version of itself, yielding [eight–phase–shift keying (8-PSK)]) uses both amplitude modulation (AM) and phase modulation (PM) versus GSM IF (3) (GMSK) which requires only PM. Most GSM transmitters use an on-channel voltage-controlled oscillator (VCO) that is phase which can also be represented as modulated with a translational loop phase-locked loop (PLL) [2], [3]. This architecture is very efficient for constant AM, since all RF filtering can be eliminated, but it is incompatible IF with nonconstant-AM such as EDGE. (4) The AM of EDGE requires a linear transmitter architecture. One conventional approach for linear transmitters is direct up- The undesired component is removed by the amplitude conversion with an I/Q modulator. Unfortunately, since linear path filter leaving only the desired AM signal . The PM amplifiers tend to be noisier than saturated amplifiers, the up- and AM signals are then recombined in the RF variable-gain converter output is likely to have a wideband noise floor that amplifier (VGA) at the output of the transmitter. would fail the noise requirements of the standard. As a result, In GSM mode, the AM path is disabled and the transmitter RF filtering would be required at the transmitter output to meet reverts to a conventional nonlinear offset PLL transmitter. the receive-band noise requirements. Further, a linear modulator This implementation includes two RF paths, each covering is likely to consume more power than a nonlinear modulator. two bands to form a quad-band solution. The low band covers the GSM-850 (824–849 MHz) and GSM-900 (880–915 MHz) II. SYSTEM ARCHITECTURE AND REQUIREMENTS bands. The high band covers the DCS (1710–1785 MHz) A polar modulator architecture [4]–[8] was chosen to avoid and PCS (1850–1910 MHz). A detailed block diagram of the the external RF filters that would likely be necessary with other complete transmitter is shown in Fig. 2 with both signal paths transmitter topologies and for compatibility with existing GSM and the required calibration circuits. Although the solution is transmitters for an efficient dual-mode solution. A simplified a complete GSM/EDGE transceiver, the focus of this paper is block diagram of the GSM/EDGE transmitter is shown in Fig. 1. the polar modulator IC. An I/Q modulator generates an EDGE (or GSM) modulated IF One of the key challenges in the polar modulator architecture output from the baseband inputs. The IF output contains both is the tradeoff between the RF spectrum and noise. A summary AM and PM in EDGE mode, which can be generically repre- of the requirements [1] is shown in Table I. A system-level anal- sented as ysis of a polar modulator reveals that inadequate bandwidths in the PM and AM paths result in spectral regrowth that can vio- IF (1) late the transmit mask requirement as imposed by the standard. This spectrum degradation can be understood by comparing the amplitude and phase components of the signal to the composite Manuscript received April 15, 2004; revised July 8, 2004. signal and the transmit mask, as seen in Fig. 3. Note that the The authors are with Analog Devices, Inc., Greensboro, NC 27409 USA (e-mail: [email protected]). phase component of the signal exceeds the mask requirement Digital Object Identifier 10.1109/JSSC.2004.836340 by over 25 dB at 400-kHz offset from the carrier.

Fig. 1. Simpliﬁed transmitter block diagram.

Fig. 2. Detailed block diagram.

Fig. 4 shows the relationship between AM and PM band- TABLE I widths and margin to the transmit mask at the critical offset SYSTEM REQUIREMENTS of 400 kHz from the carrier. At 400-kHz offset, the spectrum is the most sensitive to transmitter impairments that result in spectral regrowth. Clearly, wider modulation bandwidths result in more margin to the mask requirement. However, as the signal path bandwidths are increased, the out-of-band noise also in- creases. The most stringent noise requirement in EDGE mode is the noise in the receive band at 20-MHz offset from the carrier. The transmit mask requirements are the same for both the 2192 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004

The transmit output in the presence of AM-path dc offsets contains both the desired signal and an undesired PM component with an amplitude equal to the dc offset. The PM signal exceeds the 400-kHz offset spectral mask requirement by more than 25 dB, as shown in Fig. 3. This translates to a dc offset limit in the amplitude path of 0.2% relative to the peak of the envelope, as veriﬁed with system simulations. In addition to dc offsets, ﬁnite isolation from input to output in the RF VGA can provide a path for the PM signal to leak to the transmit output and degrade the spectral mask. This is discussed in greater de- tail in Section III-C.

III. IMPLEMENTATION DETAILS A. PM Calibration The conflict between the narrow bandwidth required for noise filtering and the wide bandwidth required to meet the RF spectrum mask dictates very tight control of the phase and enve- Fig. 3. Signal bandwidths. lope bandwidths as described in Section II. The bandwidth of the transmit PLL (i.e., the phase modulator) is proportional to where is the charge-pump current and is low and high bands, but the noise requirement is more difficult the VCO gain. The VCO gain could have 50% variation and, in the low band, so the focus will be on the low band. as a result, the bandwidth can also vary significantly. In order Fig. 5 shows the relationship between the PM bandwidth and to maintain the bandwidth with the required 6% accuracy, we margin to the transmit mask on one axis and receive-band noise must calibrate the charge pump current such that the , margin on the other axis for EDGE mode. The transmitter is product is roughly constant. designed to have 6-dB margin to the transmit mask requirement The standard allows 200 s for the frequency synthesizer to to minimize the back-off required in the power amplifier—thus lock prior to a transmit burst. Since the transmit PLL lock time maximizing its efficiency. The receive-band noise target in- is much less than 200 s, we can use that time to calibrate the cludes 3 dB of margin to ensure manufacturability. Thus, the bandwidth. A block diagram of the calibration scheme is shown phase modulator bandwidth must be between 3.5–4.0 MHz in Fig. 7. The first step to calibrating the bandwidth is measuring or about 6%. The phase modulator is a PLL which has . Before we measure , we must set the VCO fre- significant bandwidth variation due to the variation of quency to close to the desired carrier frequency since the integrated VCO. The VCO gain variation can result in depends on frequency. nearly 30% bandwidth variation. A calibration is utilized to The calibration begins with the calibration of the VCO’s minimize bandwidth variation, as described in Section III-A. 2 bits of tank tuning. The control voltage is disconnected from The polar transmitter has other impairments that can degrade the PLL and set to a reference voltage, and the frequency is spectral mask performance beyond just the bandwidth require- determined by counting VCO periods for “ ” times the period ments. These impairments include delay mismatch between the of the 13-MHz GSM reference clock. “ ” is set such that the AM and PM paths, dc offsets in the amplitude path, and finite resolution is on the order of 1 MHz which is a good tradeoff RF isolation between the PM signal and the transmit output. The between calibration time and the desired accuracy. The counter AM and PM signals must be correctly time aligned before re- output is combination at the RF VGA. The delay mismatch system simulations demonstrate the requirements, as shown in Fig. 6. Notice COUNT (7) that the EVM performance is relatively insensitive, but the spectral mask margin degrades significantly with delay mismatch. where is the GSM reference time base and is the As a result, the group delay of the AM path must match the VCO period. The VCO frequency is then group delay of the PLL PM path to meet the transmit mask requirements. COUNT The amplitude path is also sensitive to dc offsets which pro- (8) vide a leakage path for the PM signal to the transmit output. This can be observed mathematically as follows: A successive approximation (SAR) algorithm is used to minimize the error between the VCO frequency and the desired carrier frequency, which is represented by “TX channel” in Fig. 7. TX DC (5) After the coarse frequency adjustment is completed, another SAR algorithm is used to set the VCO frequency close to the which can also be expressed as desired transmit carrier frequency by driving the control voltage with a digital-to-analog converter (DAC). This is required to ac- TX DC (6) count for the variation with control voltage within each ELLIOTT et al.: A POLAR MODULATOR TRANSMITTER FOR GSM/EDGE 2193

Fig. 4. Modulation bandwidth requirements.

Fig. 7. Phase-bandwidth calibration block diagram.

control voltage to two different settings and measuring the frequency deviation . A digital algorithm uses the measured data to set the charge-pump current in- Fig. 5. Receive-band noise and transmit mask versus PM bandwidth (EDGE versely proportional to . mode). In GSM mode the output SNR requirement is 6 dB more stringent than in EDGE mode. However, the PM bandwidth is much narrower than in EDGE mode. Thus, in GSM mode, the translational-loop PLL bandwidth is automatically reduced to meet the more stringent receive-band noise speciﬁcations. The bandwidth is reduced by decreasing the charge pump current by a factor of two and doubling the loop ﬁlter pole capacitance. The slight increase in closed-loop peaking of the PLL response is tolerable due to the relaxed bandwidth requirements in GSM mode and the added stability of the bandwidth calibration that is not typically present in most PLLs.

B. RF Path: GMSK Modes A simpliﬁed schematic of the GSM-band PM path in GMSK mode is shown in Fig. 8. The description of the PM path is fo- cused on the GSM band which has a much more stringent noise Fig. 6. Transmit mask and EVM versus delay mismatch (EDGE). requirement. The DCS/PCS band implementation is simply frequency scaled from the GSM-band implementation. The most capacitor setting of the VCO. Once the VCO is operating at difﬁcult requirement is the noise in the receive band, as men- the transmit frequency, the is measured by forcing the tioned in Section II. Ideally, the noise should be dominated by 2194 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004

Fig. 8. GMSK signal path. the VCO. For this to be true, the RF driver’s noise contribution that noise will be dominated by the VCO which means that any should be very small. elements following the VCO must contribute very little noise. The VCO—shown on the left of Fig. 8—uses a PMOS core Consider a simple calculation of the RF driver’s base-current despite the availability of SiGe bipolar transistors. The SNR is noise contribution. The RF driver amplifier in GMSK mode is a optimized by maximizing the voltage swing on the tank which is simple differential pair with a roughly 16-mA tail current. The proportional to and by minimizing the noise current added SNR at the input to the RF driver considering only the driver’s by the cross-coupled pair, which is proportional to . Thus, base current noise with a noiseless tail current is transistors with a low are preferable for a low-noise VCO. The inductors are made with 4- m-thick aluminum. The in- (9) ductor Q is about 11 at 870 MHz. We use two single-ended inductors rather than a center-tapped structure—which could have where is the rms signal amplitude at the input to the higher Q in less area—because electromagnetic simulations pre- driver and is the effective parallel impedance of the dict that this structure reduces magnetic coupling from the VCO VCO’s tank. With a 500-mVp input amplitude, , to the transmitter’s output. As mentioned in Section II, any cou- and , the SNR is about 162 dB/Hz, which would pling from the phase-modulated carrier to the output results in clearly not be acceptable. Thus, the VCO’s relatively high degradation of margin to the transmit mask. The varactors are output impedance must be isolated from the RF driver with a made from collector-base pn diodes. The VCO’s frequency is buffer. The buffer also isolates the VCO’s tank from the input digitally tunable in four roughly 45-MHz steps in order to cover capacitance of the driver, which enhances the tuning range. the wide tuning range requirement with reasonably low . The VCO buffer is a simple emitter follower which is lightly The primary challenge here is low phase noise at 20-MHz capacitively coupled to the tank. As seen in (9), special care offset from the carrier. Aside from the obvious requirements must be taken to minimize the noise current being injected into of a high-Q tank and minimal in the core transistors, a the tank. In fact, the design of the VCO’s buffer should be an low-noise bias current is critical to achieving low phase noise. integral part of the VCO design itself. The bias current is derived from a low-noise (i.e., high current) The GMSK driver is a simple open-collector fully switched reference and low-pass filtered using an off-chip ca- differential amplifier. If the tail current has a low-pass noise pacitor. Further filtering of high-frequency noise that could be characteristic, that noise is upconverted to the carrier frequency downconverted to the carrier frequency is applied by the tail and results in a bandpass output noise characteristic. Simula- inductor/capacitor combination [9]. The tail current is propor- tions predict that, as long as the input amplitude is several hun- tional to absolute temperature (PTAT).A PTATtail current keeps dred millivolts or larger, the output noise will be dominated the voltage swing on the tank roughly temperature-independent by upconverted bias current noise. The ratio of dc current to since the temperature coefficient of the metal used to make the noise current of a resistively degenerated current source (ne- inductors results in a roughly inverse PTAT tank impedance. glecting noise contributions from base resistance and from the The VCO signal must be amplified in order to drive the base-voltage source) is 50- external load at 5 dBm. The ultimate noise requirement is 165 dBc/Hz at 20-MHz offset from the carrier. Practically, (10) ELLIOTT et al.: A POLAR MODULATOR TRANSMITTER FOR GSM/EDGE 2195

Fig. 9. EDGE mode RF driver. where is the voltage drop across the degeneration 6-dB steps—is provided by digitally deselecting fingers of Q5 resistor. Due to voltage headroom constraints, the maximum and Q6. Additional power control is provided in the envelope practical is 600 mV. If our target for this contributor detector circuits with 1-dB step resolution for a total range of to noise is 175 dB/Hz, the minimum bias current is 14 mA. 38 dB. The amplitude path also provides an additional 30 dB of We use 16 mA to ensure adequate SNR under all process analog-controlled attenuation for power ramping. conditions. Special care must be taken to lowpass filter the At first glance, it may appear that the EDGE-mode RF path is base voltage source and to ensure that the base voltage source similar to a linear modulator since Q5/6 are essentially a linear has sufficiently low impedance to make the current source’s transconductor. The difference is that a linear modulator re- base current noise negligible. We reuse the capacitors from quires a complex mixer and a quadrature local oscillator and so the envelope path filter to provide a sufficiently low-frequency is inherently noisier than this implementation of a polar modu- filter of the GMSK driver’s bias voltage. lator. Further, since Q5/Q6 are not, on average, in their balanced state, they are less noisy than a balanced transconductor con- C. RF Path: EDGE Modes suming the same current. A simplified schematic of the RF path used for EDGE Recall that the phase-modulated carrier has much wider band- modes—with the EDGE-specific components emphasized—is width than the desired output signal. Leakage from the EDGE shown in Fig. 9. The VCO and VCO buffer are the same as driver’s input to the output can result in a failure of the transmit those in Fig. 8. The RF driver consists of Q1–6. Q7–8 is the mask requirement. The input signal amplitude is 500 mV and GMSK-mode driver described in the previous section with the amplitude at the collectors is about 800 mVp (2 dBm in 200 the tail current source omitted since it is a high impedance ). Recall from Section II that the phase-modulated carrier fails in EDGE mode. Q9–10 is a dummy driver that is included the transmit mask by 25 dB and our transmit spectrum goal by to ensure balance. Q7–10 are shown here to emphasize the 31 dB. Thus, the isolation required at maximum gain is 35 dB. importance of balance. The output signal is The isolation requirement is increased as the output power is reduced. We use very careful layout to ensure that the coupling from the driver’s input to the output is perfectly balanced. Imperfect matching of the capacitance at the collectors of Q5 (11) and Q6 provides another path for the phase-modulated carrier where is the carrier frequency, is the PM, and is the to leak to the output. The rectification at those nodes creates a amplitude of the input signal. The envelope information is repre- frequency component at twice the input frequency, which mixes sented as . Note that, although the circuit appears to be with the input frequency to produce an input frequency compo- balanced, the envelope is always positive so is always greater nent at the output. Minimizing those capacitances and ensuring than . is 16 mA. The maximum instantaneous output adequate matching solves the problem. power is achieved when mA and . On average, Imperfect matching of and is yet another mechanism is about 0.707 16 mA since the peak-to-average ratio by which the phase-modulated carrier can leak to the output as of the EDGE signal is 3 dB. Coarse power control—in three shown in Section II. System simulations predict that the dc offset 2196 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004

Fig. 10. Envelope-ﬁlter simpliﬁed schematic.

Fig. 12. Transmitter output spectrum (EDGE mode). Fig. 11. Die photograph. of the amplitude path to the PM path. The PM path is a type-II in the envelope path must be less than 0.2% relative to the peak PLL which has the following generic transfer function: of the envelope signal. This corresponds to a dc offset referred to the bases of Q5 and Q6 of less than 1 mV. The desired matching (12) in and is achieved by using large interdigitated devices and as much degeneration as voltage headroom will allow. which yields the following group delay The envelope path circuits—described in the next section—cannot achieve the desired offset so a calibration is (13) required. A low-input-offset ( 0.25 mV) comparator measures the dc offset between the bases of Q5 and Q6. The offset is minimized by an 8-b DAC using a successive approximation It can be seen from this equation that the group-delay charac- algorithm. teristic of the PLL is near zero at low frequency offsets which is the region of interest. Thus, the amplitude path filter must main- D. Envelope Path Filter Implementation tain the same low group-delay characteristics. The filter implementation mimics the PLL with two A simplified single-ended representation of the amplitude integrators and a zero for compensation. The first integrator is path filter is shown in Fig. 10. The filter has two primary formed by the cell Q1/R2 and load capacitor C2. The second requirements, wideband noise suppression and delay matching integrator is formed by the cell Q2/R4 and load capacitor ELLIOTT et al.: A POLAR MODULATOR TRANSMITTER FOR GSM/EDGE 2197

(a)

(b)

Fig. 13. (a) Transmit mask versus frequency (low band; 400-kHz offset). (b) Transmit mask versus frequency (high band; 400-kHz offset).

C1. The zero is implemented at the filter input with R1/C1. of the RF VGA (Q3/R5). This results in voltage distortion In order to minimize the output noise of the filter, the voltage at the output of the filter to predistort the input voltage to the output is not buffered prior to the VGA input so as not to intro- VGA such that a linear relationship is maintained. duce any unfiltered wideband noise or additional current dissi- pation. As a result, the VGA input impedance must be accounted IV. MEASUREMENT RESULTS for in the design of the filter. As shown in Fig. 10, the output of the amplitude filter is a The polar modulator IC is implemented in 0.5- m SiGe voltage input to the RF VGA. The signal of interest, however, BiCMOS with a die area of 7.8 mm . A die photograph is is the collector current, , of the RF VGA, as described in shown in Fig. 11. A spectrum of the transmitter output in Section III-C. That is, the transmit output AM is controlled by EDGE mode along with the transmit mask is shown in Fig. 12. the VGA output current . This current output must Clearly, the transmitter has significant margin to the require- be a linear representation of the AM signal to prevent spectral ments. The spectral integrity is maintained across all transmit mask degradation. System simulations predict an HD3 of 40 bands in both GSM and EDGE modes, as shown in Fig. 13. dBc to prevent significant impacts on the 400-kHz mask. The Fig. 14 shows a plot of the receive band noise measurements final Gm cell of the amplitude filter (Q2/R4) is a scaled version for both GSM and EDGE modes against the requirements of the 2198 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 39, NO. 12, DECEMBER 2004

(a)

(b) Fig. 14. (a) Receive band noise (low band). (b) Receive band noise (high band; 1910 MHz).

TABLE II V. C ONCLUSION SUMMARY OF TRANSMITTER PERFORMANCE In this paper, a polar modulator IC for EDGE was described. The polar modulator architecture eliminates the requirement for external surface acoustic wave ﬁlters that would likely be required with a linear transmitter because the RF path is composed of fully saturated—and thus low-noise—circuits. Implementation challenges were analyzed and the primary challenge—phase-modulator bandwidth control—was ad- dressed by a calibration of the transmit PLL’s loop bandwidth. Measurement results prove that the solution meets the EDGE and GSM requirements with comfortable margin.

REFERENCES [1] ETSI EN 300 910. GSM 05.05, 1999. [2] J. Strange and S. Atkinson, “A direct conversion transceiver for multi- band application,” in RFIC Symp. Dig., 2000, pp. 25–28. [3] T. Yamawaki, M. Kokubo, K. Irie, H. Matsui, K. Hori, T. Endou, H. Hagisawa, T. Furuya, Y. Shimizu, M. Katagishi, and J. R. Hildersley, “A 2.7-V GSM RF transceiver IC,” IEEE J. Solid-State Circuits, vol. 32, pp. 2089–2096, Dec. 1997. standard. The stability of the noise and spectral performance is [4] L. R. Kahn, “Single-sideband transmission by envelope elimination and due to the calibration techniques employed, in particular,the PLL restoration,” in Proc. IRE, July 1952, pp. 803–806. [5] D. Su and W. J. McFarland, “An IC for linearizing RF power ampli- bandwidth control as described in Section III-A. A summary of ﬁers using envelope elimination and restoration,” IEEE J. Solid-State the transmitter measured performance is shown in Table II. Circuits, vol. 33, pp. 2252–2258, Dec. 1998. ELLIOTT et al.: A POLAR MODULATOR TRANSMITTER FOR GSM/EDGE 2199

[6] E. McCune and W. Sander, “EDGE transmitter alternative using non- Jon Strange (M’00) received the B.Sc. degree in linear polar modulation,” in Proc. Int. Symp. Circuits and Systems, vol. physics from the University of Durham, Durham, 3, May 25–28, 2003, pp. III-594–III-597. U.K., in 1984 and the M.Sc. degree in microelec- [7] W. B. Sander, S. V. Schell, and B. L. Sander, “Polar modulator for multi- tronics from the University of Edinburgh, Edinburgh, mode cell phones,” in Proc. Custom Integrated Circuits Conf., Sept. U.K., in 1985. 2003, pp. 439–445. From 1985 to 1987, he was a Researcher with [8] T. Sowlati, D. Rozenblit, E. MacCarthy, M. Darngaard, R. Pullela, D. the VLSI Department, Thorn-EMI Central Research Koh, and D. Ripley, “Quad-band GSM/GPRS/EDGE polar loop trans- Laboratories, and from 1988 to 1991 he held several mitter,” in Proc. Int. Solid State Circuits Conf., San Francisco, CA, 2004, design and engineering management positions for paper 10.3. LSI Logic. In 1991, he cofounded Mosaic Micro [9] E. Hegazi, H. Sjoland, and A. A. Abidi, “A ﬁltering technique to lower Systems, Ltd., an analog design consultancy special- LC oscillator phase noise,” IEEE J. Solid-State Circuits, vol. 36, pp. izing in RF and radio system IC design. Since 1996, he has been with Analog 1921–1930, Dec. 2001. Devices, Inc., Kent, U.K., were he is currently Engineering Director within the RF and Wireless Business Unit responsible for IC product design in the area of radio systems integration. He has authored eight technical papers and has been Michael R. Elliott received the B.S. degree from the granted two U.S. patents, with four pending. University of Missouri-Rolla in 1994 and the M.S. Mr. Strange receoved the the Chalmers Prize in physics in 1984. degree from Purdue University, West Lafayette, IN, in 1995, both in electrical engineering. He joined Analog Devices, Inc., Greensboro, NC, in 1996 as an Analog Integrated Circuit Designer. His areas of interest include high-speed data converters, Allen Hill received the B.S. degree from Guilford frequency synthesizers, and communications circuits College, Greensboro, NC. and systems. He is an Applications Engineer with 23 years ex- perience at Analog Devices, Inc., Greensboro, where he is presently with the Communications Group of the High Speed Converters Division.

Tony Montalvo received the B.S. degree in physics from Loyola University, New Orleans, LA, in 1985, the M.S. degree in electrical engineering from Columbia University, New York NY, in 1987 and the Ph.D. degree in electrical engineering from North Carolina State University, Raleigh, in 1995. From 1987 to 1991, he was a Flash Memory De- signer with AMD. From 1995 to 2000, he with was with Ericsson, Research Triangle Park, NC, where he Sanjay Nandipaku (M’00) received the B. Tech. and was the Manager of the RF and Analog IC group. M.S. degrees from the Indian Institute of Technology, Since 2000, he has been the Director of the Analog Madras, in 1988 and 1991, respectively, both in elec- Devices Raleigh Design Center, Greensboro, NC. He is also an Adjunct Pro- trical engineering. fessor at North Carolina State University. He holds 15 patents and has authored His interests lie in the area of communication or coauthored 10 papers. system design, specializing in RF wireless systems Dr. Montalvo has been on the ISSCC Program Committee since 2002. He was area. He is currently an RF System Design Engi- named Outstanding Teacher in 1995 at North Carolina State University. neer with Analog Devices, Inc., Wilmington, MA, working on the development of architectures and products for mobile wireless market space. Prior to this, he was a Research Engineer with the Center for Brad P. Jeffries received the B.S. degrees in elec- Development of Telematics, Bangalore, India, and the RF Group Lead with trical engineering and computer engineering from Midas Communication Technologies, Madras, India. North Carolina State University, Raleigh, in 2000. In 2000, he joined Analog Devices, Inc., Greens- boro, NC, where he has been involved in the design of communications integrated circuits. Johannes Harrebek was born in Aarhus, Denmark, on May 11, 1971. He received the M.Sc.E.E. degree from University of Aalborg, Aalborg, Denmark, in 1995. He was a Research and Development (R&D) RF Engineer with Cortech from 1995 to 1998 on DECT terminal development. In 1998, he joined Frank Murden received the B.S. degree from the Bosch Telecom, Denmark, as an R&D RF engi- University of Louisville, Louisville, KY,and the M.S. neer on HSCSD GSM mobile phone development degree from the University of Arizona, Tucson, both projects with a focus on frequency synthesis and in electrical engineering. antenna design. From 2000 to 2001, he was with Since graduation, he has been with Analog De- the Department of GSM/EDGE Systems, Siemens Mobile, Denmark. He vices, Inc., Greensboro, NC, doing a wide variety of joined Analog Devices, Inc., in 2001 on the Analog Devices EDGE radio designs. implementation project and is now RF Systems Development Manager with Analog Devices Wireless System Applications Center, Aalborg, Denmark, developing form factor reference designs on the latest Analog Devices wireless systems solutions.