CALIFORNIA STATE UNIVERSITY, NORTHRIDGE IMPATT DIODE POWER ACCUMULATOR A thesis submitted in partial satisfaction of the requirements for the degree of Master of Science in Engineering by Steven Eugene ~amilton / June, 1979 The Thesis of Steven Eugene Hamilton is approved: Steven D. Gavazza Edmond S. Gillespie~ California State University, Northridge ii TO MY LOVING WIFE AND CHILDREN iii TABLE OF CONTENTS Page ABSTRACT ix Section I INTRODUCTION 1 Section II ACTIVE CIRCUIT DESIGN 3 Section III SINGLE DIODE WAVEGUIDE OSCILLATOR 14 Section IV HI-PAC DESIGN 20 Section V POWER COMBINER PERFORMANCE 46 Section VI CONCLUSIONS 57 REFERENCES 59 APPENDIX A DIODE CHARACTERIZATION 62 LIST OF TABLES v LIST OF FIGURES vi iv f ' LIST OF TABLES . TABLE 1 Power and Efficiency Versus Rc p4~e 2 Power and Efficiency Results 52 Al Varian Diode VSX9251 AD 99 A2 Diode Comparison 100 v LIST OF FIGURES Figure Page 1 IMPATT Diode 5 (a) Structure {b) Field Profile (c) Electron Energy (d) Voltage Wave Form (e) Injected and External Current 2 Equivalent Circuit of Diode and Load 8 (a) Diode Chip and Load (b) Packaged Diode and Load ·3 Coaxial Oscillator 10 (a) Structure {b) Equivalent Circuit 4 Design Curves for a Single Step 12 Transformer 5 Single Diode Waveguide Oscillator 15 (a) Side View (b) End View 6 Equivalent Circuit of Single Diode 17 Waveguide Oscillator 7 Kurokawa Waveguide Oscillator End View 21 8 Kurokawa Waveguide Oscillator Top View 22 9 24 Diode HiPac 24 10 Module Configuration 26 (a) Kurokawa Module (b) HiPac Module 11 Cross-Sectional View of Diode 27 12 Module Spacing for HiPac 29 13 Waveguide TE011 Mode Cavity 31 14 Power Versus Current for Different 36 Circuit Load vi Figure pjge 15 Equivalent Circuit of HiPac 16 Power and Efficiency Versus Current 40 for Rc < Ropt 17 Power and Efficiency Versus Current 41 for Rc = Ropt 18 Power and Efficiency Versus Current 42 for Rc > Ropt 19 IMPATT Oscillator Test Set-up 45 20 Assembled End View of HiPac 47 21 Disassembled View of HiPac 48 22 Power and Efficiency Versus Current 49 8 Diodes in HiPac 23 Comparison of HiPac Performance to so Eight Times Single Diode Data 24 Power and Efficiency Versus Current 53 4 Diodes in HiPac 25 Power and Efficiency Versus Current 54 2 Diodes in HiPac 26 Mechanical Tuning Bandwidth of HiPac 56 Al Low-High-Low Diode 64 (a) Doping Profile (b) Electric Field A2 Hie;h-Low Diode 65 (a) Doping Profile (b) Electric Field A3 Cross Section of High-Low Diode 66 A4 High-Low Diode 67 (a) Layer Definition (b) Electric Field AS Mean Life to Failure 76 A6 Cross-Sectional View of Diode 78 A7 Doping Profile of High-Low Diode 82 vii Figure AB C-V Plot of High-Low Diode PB~e A9 Equivalent Circuit of Packaged Diode 90 AlO Equivalent Circuit of the Packaged 92 Diode and Load Reactance viii ABSTRACT IMPATT DIODE POWER ACCUMULATOR by Steven Eugene Hamilton Master of Science in Engineering This thesis presents the development of a new type of I~ATT diode power accumulator. The design has twice the capacity of similar accumulators of the same size. The circuit offers high combining efficiency, reduced thermal interaction and broad tuning bandwidth. The basic concepts of IMPATT oscillators are dis­ cussed. A single diode version of the power accumulator is presented in detail along with an equivalent circuit. A technique to optimize IMPATT oscillators is covered with experimental verification. The power combiner performance is demonstrated utilizing 2, 4 and 8 diodes. The broad tuning bandwidth of the combiner is presented for the eight diode configuration only. The optimum performance of a power combiner is attained when all the diodes, to be combined, have similar characteristics. To achieve the optimum condition, a ix technique is presented which determines if the diodes are suitable for combiner application. X SECTION I INTRODUCTION The development of solid state microwave sources has experienced tremendous growth over the past ten years. Their high reliability, low cost and small volume are characteristics which attract many designers. Solid state sources have been developed for communications, space and radar systems to replace low and medium power tube type transmitters. The principal devices responsible for solid state growth have been FETs and IMPATTs*. These devices have all served as the fundamental building blocks in the new solid state components. Within the past six years the IMPATT diode has clearly set the pace for solid state technology by replacing several tube type transmitters. This has been accomplished by the development of many new devices and power combining circuits. Recently, improvements in combiner power level have been attained only through the development of higher power diodes. This presents a problem to the solid state design­ er in that future requirements will exceed the capability * FET is an acronym for Field Effect Transistor. IMPATT is an acronym for Impact and Transit Time Device. 1 ~'· 2 of present combining techniques. To maintain their growth, solid state designers have resorted to utilizing circuits which can potentially meet future requirements by combining larger numbers of diodes; unfortunately, the new circuits operate at reduced efficiency, require increased fabrica­ tion cost and exhibit reduced reliability. The aforemen­ tioned characteristics reduce the advantages that solid state sources have over other transmitter technologies. It is the object of this thesis to reclaim some of the desirable characteristics of solid state sources and, in addition, to satisfy future power requirements. To accom­ plish this task, a new combining technique has been developed which is capable of summing more devices than was previously attainable. This technique offers high combin­ ing efficiency and greater reliability than other designs. In order to demonstrate the new combiner circuit, three oscillators were designed and fabricated. The design utilizes a waveguide cavity with a plurality of coaxial modules located along the cavity walls. IMPATT diodes are located at one end of a coaxial transmission line with a bias filter at the other end. The test results are presented both in tabular and graphical form to illustrate the circuits capability and to compare its performance against previously developed technology. A simple model will be presented to discuss the various characteristics of the new circuits, hereafter referred to as HiPac. ~·· SECTION II ACTIVE CIRCUIT DESIGN A. Introduction The design of an active circuit utilizing negative resistance devices is a complicated process. The param­ eters which must be considered involve areas of semi­ conductor physics and standard microwave circuit technol­ ogy. Unfortunately, the non-linearity of the device, which makes it useful, also prevents closed form solutions in large signal analysis. In addition, as one strives for high power sources, devices are typically combined in complex microwave circuits. These circuits usually have mutual coupling interactions which are as equally difficult to analyze as the active device. In this section, several aspects of the design will be presented, although only those subjects pertinent to the design process will be discussed. Approximations and limits will be presented as required, and more detailed studies will be referenced. B. IMPATT Diode The type of device used in this study is a gallium arsenide IMPATT [1] [2]. The GaAs IMPATT is the highest efficiency device in the IMPATT family [3] [4]. Silicon 3 4 and indium phosphide materials are also used in construct­ ing IMPATT diodes; but they operate at half the efficiency of GaAs. As previously stated, the IMPATT diode is a negative resistance device. The negative resistance occurs as a result of a 180° phase difference which is developed be­ tween the ac current and voltage within the device. The diode is essentially composed of two constituents: (1) the avalanche zone and (2) the drift zone (also known as depletion zone). Consider the simple P+N N+ diode shown in Figure la, with reverse bias applied as indicated. Within the diode, an electric field profile, as shown in Figure lb, is developed. The high field, which exists between x1 and x2 , establishes the avalanche zone. In this region avalanche breakdown occurs which generates hole electron pairs. The region between x2 and x3 is called the drift zone, where the field is high enough to maintain constant drift velocity but not high enough for avalanche to occur. Figure lc shows the energy band diagram under breakdown conditions. The holes generated in the avalanche zone go into the p+ region while the electrons are injected into the N region, drift zone. If an ac voltage, as shown in Figure ld, is applied to the diode, via noise or some other stimulus, the electric field will change periodically with time around some average value. The rate of impact ionization will follow the change in field almost 5 14--- DRIFTZONE ~ p+ N N+ -1 I i I ~+ ~~-- AVALANCHE ZONE (a) Structure ELECTRIC FIELD ~ p+ I N N+ I I X x 1 x 2 x3 DISTANCE (b) Field Profile ELECTRON ENERGY DISTANCE (c) Electron Energy AC VOLTAGE 8 = Wt (d) Voltage Wave Form CURRENT T 2T (e) Injected and External Currents Figure 1. lmpatt Diode 6 instantaneously. However, the carrier density does not follow the field change in unison because the generation of carriers depends on the number already generated. So when the field is maximum, carrier generation is still increas­ ing and does not peak until after the field has decreased by some amount. Thus we have the carrier density peak lagging the ac voltage by about 90° (see Figure ld and le). The injected electrons then enter the drift zone where, because of the field, they travel at a saturated or scattering-limited velocity.