Coexistence of G.Fast and Vdsl in Fttdp and Fttc Deployments

23rd European Signal Processing Conference (EUSIPCO)

COEXISTENCE OF G.FAST AND VDSL IN FTTDP AND FTTC DEPLOYMENTS

Rainer Strobel∗ †, Wolfgang Utschick∗

∗Fachgebiet Methoden der Signalverarbeitung †Lantiq Deutschland GmbH Technische Universität München, 80290 München, Germany 85579 Neubiberg, Germany {rainer.strobel,utschick}@tum.de [email protected]

ABSTRACT Hybrid copper/fibernetworks bridge the gap between the fiber link and the customer by using copper wires over the last me- CPE CPE CPE ters. This solution combines energy efficiency and low cost of CPE the copper network with higher fiber data rates. ITU recently VDSL finished the G.fast standard for high speed data transmission Cabinet on copper wires for this application. Fibers DP Coexistence with legacy VDSL2 systems is an important CPE topic for the introduction of the new technology, as the systems share a significant part of the frequency spectrum. Fig. 1. FTTdp coexisting with FTTC in one cable bundle This paper investigates the performance of G.fast coexisting with VDSL2. Methods for decentralized spectrum optimization and protection of legacy services are presented.

Index Terms— spectrum optimization, alien crosstalk, it- CPE CPE CPE erative water-ﬁlling, FTTdp, coexistence Fiber CPE VDSL G.Fast 1. INTRODUCTION Cabinet

Coexistence of G.fast [1] with legacy ADSL and VDSL2 [2] CPE is a key to success of the new technology. The VDSL2 fiber to the curb-architecture (FTTC) is extended [3] by the fiber Fig. 2. VDSL2/G.fast multi-mode cabinet to the distribution point (FTTdp) architecture using G.fast for the copper link between the distribution point (DP) and the customer premises equipment (CPE). Some of the subscribers 2. SYSTEM MODEL are still served with the legacy service, while others have been upgraded to G.fast. But they share the same cable binder, and 2.1. Transmission Technology Basics therefore, there is crosstalk between the services (see Fig. 1). G.fast as well as VDSL2 are based on DMT modulation [2] Another coexistence scenario is shown in Fig. 2. The [1]. We assume multi-carrier transmission with K carriers on FTTC locations can be upgraded to serve G.fast. Higher data L lines in upstream (us) and downstream (ds) direction. The rates are available for subscribers located close to the cabinet, (k) (k) transmit power p for downstream and p can be config- while subscribers with longer lines or with legacy equipment ds l us l ured for each line l = 1,...,L and carrier k = 1,...,K to are served with the legacy service. achieve the desired power spectral densities (PSD) ψds l(f) Both G.fast deployment strategies result in mutual cou- and ψus l(f) according to plings between G.fast and VDSL2 services. This paper investigates the performance impact of such couplings and shows fk +∆fl/2 methods for crosstalk management, which are primarily im- (k) p = ψ(ds/us) l(f)df (1) plemented on the G.fast side and do not require coordination (ds/us) l Z between G.fast and VDSL2. fk−∆fl/2

This work has been founded by the research project “FlexDP - Flexible with a subcarrier spacing ∆fl and a carrier frequency fk = Breitband Distribution Points”, funded by the Bayerische Forschungsstiftung k∆fl. The line indices are grouped into lines using VDSL2

I I CPE NEXT l ∈ V and G.fast l ∈ F. VDSL2 and G.fast use different DP NEXT subcarrier spacings, ∆fl =4.3125 kHz for l ∈ IV and ∆fl = 51.75 kHz for l ∈ IF. While A/VDSL use frequency division CPE duplexing (FDD), G.fast uses time division duplexing (TDD). VDSL Cabinet US/DS The spectral characteristics of both services are very dif- FEXT ferent. G.fast uses a low per-line sum transmit power of CPE psum l = 4 dBm for l ∈ IF, which is spread over a wide DP frequency band from 2.2 MHz to 106 MHz, while VDSL2 uses more than 10 times higher transmit power of psum l = Fig. 3. Crosstalk couplings between G.fast and VDSL2 14.5 dBm for l ∈ IV which is concentrated on a smaller frequency band from 25 kHz to 17.6 MHz. VDSL2 runs a DMT I symbol rate of 1/tsym l = 4 kHz for l ∈ V while G.fast uses The interference plus noise PSD ψin v(f) is given by a much higher symbol rate of 1/tsym l = 48 kHz for l ∈ IF. For each line l, subcarrier k and direction (us, ds), a cer- ψ in (f)= ψFEXT (f)+ ψ NEXT (f) + ψn (k) (ds/us)v (ds/us)vd (dp/cpe)vd tain signal-to-interference-noise ratio SINR is present. I (ds/us) l d∈ dist v (k) X (4) The number of bits b(ds/us) l per symbol is obtained by where Idist v are the indices of lines interfering v. The (k) background noise is assumed to be additive white Gaus- (k) SINR(ds/us) l sian (AWGN) zero-mean noise with a ﬂat noise PSD ψn. b(ds/us) l = min log2 1+ ,bmax l (2) $ Γ !% ! The NEXT and FEXT PSDs used in Eq. (4) are derived by multiplication of the disturber transmit PSD ψd(f) with the with Γ that accounts for the SNR gap [4] and a maximum squared crosstalk transfer function, e. g., HNEXT dp vd(f) to be I 2 supported bit loading bmax l = 12 for l ∈ F and b max l = ψNEXT dp vd(f) = ψds d(f)|HNEXT dp vd(f)| . The crosstalk I 15 for l ∈ V. The data rate of line l is given by from disturber d to victim v is modeled according to [5]. The interference plus noise PSD can be converted into a K η fk +∆f/2 (ds/us) l (k) (3) (k) R(ds/us) l = b(ds/us) l. noise power per tone pin v = ψin v(f)df, which gives tsym l k=1 fk−∆f/2 X the SINR to be R Transmission overhead is incorporated by an efﬁciency factor (k) 2 (k) η where the values ηds l =0.675 and ηus l =0.2625 for l ∈ IF |H | p SINR(k) = (ds/us) v (ds/us) v (5) are used for G.fast while for VDSL2, ηds l = ηus l = 0.925 (ds/us) v (k) pin (ds/us) v for l ∈ IV is assumed. Additional cyclic extension overhead is considered within tsym. (k) (k) with a direct channel gain Hds v in downstream and Hus v in upstream direction for line v. 2.2. Mutual Couplings between G.fast and VDSL2

Due to the different duplexing schemes and the overlapped 2.3. Spectral Masks frequency spectrum between 2.2 and 17.6 MHz, there is The used transmit spectrum ψ(f) is bounded by limit PSDs crosstalk between G.fast and VDSL2, called alien crosstalk. or masks ψlimit(f), which are defined in the standards for As shown in Fig. 3, there are four different coupling paths. G.fast [6] and for VDSL2 [2]. G.fast uses only one limit PSD Each of the four receiving points (VDSL2 Cabinet, due to TDD while DSL uses differentlimit PSDs and different VDSL2 CPE, G.fast DP, G.fast CPE) experiences noise from in-band frequencies f ∈ FVds and f ∈ FVus for downstream three sources. For the affected (victim) line v, it is caused by and upstream direction. The in-band PSD holds for frequen- NEXT (near end crosstalk) ψNEXT vd(f) and far-end crosstalk cies where data is transmitted. It is controlled in frequency (FEXT) ψFEXT vd(f) from the interfering (disturber) line d (k) domain with per-tone power values pl . To protect the legacy and receiver noise ψn. service, power back-off masks ψpbo ds(f) for downstream and Self-FEXT, the far-end crosstalk between lines of the ψpbo us(f) for upstream may be used, as described later. same service is reduced by crosstalk cancellation. There- The out-of-band transmit spectrum cannot be reduced to I fore, G.fast victim lines v ∈ F are disturbed by VDSL2 zero. It is reduced with time domain transmit filters accord- I lines d ∈ V and the vice versa. We distinguish between ing to the standard requirements. Fig. 4 shows the G.fast (k) (k) downstream HFEXT ds vd and upstream HFEXT us vd FEXT and limit PSD for the 106 MHz profile together with an exam- (k) (k) DP-side HNEXT dp vd and CPE-side HNEXT cpe vd NEXT cou- ple of a measured transmit PSD with the corresponding in- plings for subcarrier k. band and out-of-band characteristics. The actual out-of band

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−60 constraints. This is a similar approach as for cognitive ra- Limit ψlimit(f) dio with temperature-interference constraints in the wireless −80 Actual ψ(f) context [7]. The general optimization problem with spec- Backoff ψ (f) pbo ds tral mask and per-line sum-power constraints of each service Backoff ψ (f) −100 pbo us reads as

In-band L PSD/dBm/Hz −120 max Rds l + Rus l (6) (k) pl ∀k=1,...,K; l=1,...,L l=1 −140 X 0 50 100 150 200 (k) (k) s.t. 0 ≤ p(ds/us) l ≤ pmask (ds/us) l ∀k =1,...,K; l =1,...,L Frequency/MHz K (k) s.t. pds l ≤ psum l ∀l =1,...,L Fig. 4. In-band and out-of-band spectrum for 106 MHz G.fast Xk=1 K (k) PSD depends on the filters used in a specific implementa- s.t. pus l ≤ psum l ∀l =1,...,L k=1 tion. ψ(f) must also satisfy the per-line sum-power psum l X constraint. Therefore, ψ(f) in Fig. 4 is below the limit mask with a sum-power limit psum l and a spectral mask constraint (k) fk +∆f/2 in most cases. p = ψlimit (f)df. mask l fk −∆f/2 l No coordination between legacy lines and G.fast lines is R 2.4. Long Range G.fast and 35 MHz VDSL2 assumed. Each group of lines determines the optimized transmit spectrum independent of the others. The systems are only Some operators prefer an intermediate upgrade step be- coupled by the crosstalk which each of the receivers observes. tween vectored VDSL2 and G.fast FTTdp. There are two Spectrum Optimization: The optimal transmit spec- approaches to serve this demand. Besides the idea of an ex- trum is obtained using an iterative water-filling approach [8]. tended VDSL2 using 35 MHz bandwidth, which increases the G.fast as well as VDSL2 have spectral mask constraints and overlapping frequency band between G.fast and VDSL2, we sum-power constraints. Water-filling with support of spectral focus on long reach G.fast as an alternative solution. It allows mask constraints has been investigated in [9]. A further ex- higher transmit power for G.fast, e.g., 8 dBm or 14.5 dBm in tension of the algorithm is required to include the bit loading combination with the higher VDSL2 limit mask for frequen- upper bound b into the optimization. Otherwise, power is cies below 30 MHz and a longer cyclic extension to support max wasted on carriers with high SINR. long lines. This system is compatible to a future upgrade The maximum bit loading is transformed into an equiv- to G.fast FTTdp, but it increases the alien interference into alent spectral mask constraint. It must be noted that, in legacy VDSL2 lines when no further actions are taken. opposite to the spectral mask constraint [9], the maximum bit loading constraint may affect convergence. The maxi- 3. COEXISTENCE STRATEGIES mum bit loading is converted into a maximum required SINR bmax l SINRmax l = 2 +1 and incorporated into the spectral This analysis focuses on two coexistence strategies, crosstalk mask constraint avoidance and optimized overlapped spectrum. (k) (k) (k) p˜ = min p · SINRmax l · ∆SNR,p (7) mask l in l mask l 3.1. Crosstalk Avoidance Strategy (ds/us) (ds/us) (ds/us) The straight-forward strategy to guarantee spectral compati- with an additional margin ∆SNR ≥ 1 to relax the bit load- bility between G.fast and VDSL2 is crosstalk avoidance. In ing constraint and avoid rate reductions during iterative water- this case, the VDSL2 frequency bands are completely ex- filling convergence. cluded from the used G.fast frequency band and only the out- For water-filling with spectral mask and sum-power con- of-band spectrum overlaps. This approach results in a min- straints, the carriers k =1,...,K for each line l are grouped I imum disturbance of the legacy service, but it causes a sub- into three groups. Carriers k ∈ l,0 with zero power, carriers I (k) (k) stantial reductionof the G.fast data rates. It shall be noted that k ∈ l,mask where the spectral mask constraint pl ≤ p˜mask l is there is still some interference between both services due to active and the remaining carriers Il,fill. The power per carrier their out-of-band transmit power. and line for each of these groups is given by

0 for k ∈ I 3.2. Overlapped Spectrum Strategy l,0 (ds/us) (k) (k) for I p = p˜mask (ds/us) l k ∈ l,mask (ds/us) (8) A better approach is to allow a spectral overlap between the (ds/us) l (k)  pni (ds/us) l µ − otherwise services and optimize the transmit spectrum with spectral  (ds/us) l (k) 2 |H(ds/us) l|  

1110 23rd European Signal Processing Conference (EUSIPCO)

(k) and µl for line l and direction (us or ds) is given by the water- The power limit pˆmask (ds/us) l including the power back-off ﬁlling solution for G.fast lines for Eq. (6) is

(k) (k) (k) ∀k : fk ∈FVus ∪FVds 1 pin l (k) (k) min p ,p˜ pbo mask l I µl = psum l + (k) − p˜mask l . pˆ mask = (ds/us) (ds/us) ∀l ∈ F |I fill|  2  l  ! l, I |H | I (ds/us) (k) k∈Xl,fill l k∈Xl,mask p˜ otherwise (9) mask (ds/us) l   (14) Power Back-Off: Sometimes, sum-rate optimization is  The upstream and downstream power constraint values p(k) not sufficient, because it is required to protect the legacy ser- pbo ds (k) vice. This is done by limiting the crosstalk into VDSL2 to a and ppbo us are derived from upstream and downstream back- maximum value pnmax according to off PSD masks for G.fast ψpbo ds(f) and ψpbo us(f) using Eq. (1) and the masks are given by (k) pni (ds/us) v ≤ pnmax (10) ψnmax 2 2 for f ∈FVus |hNEXTsum dp(f)| |hdp cab(f)| ψpbo ds(f)= 2 (15) I I ψnmax|hdp cab(f)| for all G.fast lines d ∈ F disturbing legacy line v ∈ V. Solv- 2 for f ∈F ( |hFEXTsum(f)| Vds ing Eq. (6) including the constraint (10) is not feasible in ψnmax 2 2 for f ∈FVus practice, as it requires knowledge of the crosstalk between ψ (f)= |hFEXTsum(f)| |hdp cab(f)| (16) pbo us ψnmax 2 for f ∈FVds G.fast and VDSL2 lines as well as a central coordination for ( |hNEXTsum cpe(f)| both services. With the help of crosstalk statistics, more precisely, with where hdp cab(f) is the transfer function of the cable between (k) DP and street cabinet and ψ = p /∆f. The resulting worst case values for aggregate DP-side NEXT h , nmax nmax NEXTsum dp back-off PSDs for a multi-mode cabinet are shown in Fig. 4. (k) (k) CPE-side NEXT hNEXTsum cpe and FEXT hFEXTsum, a more strict limit PSD for the G.fast lines is defined, which guar- 4. SIMULATIONS antees that (10) is satisfied with a certain probability. Approximations for worst case cabinet NEXT and FEXT For both scenarios, Fig. 1 and 2, crosstalk avoidance is com- have been proposed in [10]. The channel model [5], which pared with overlapped spectrum with and without power is used in this paper, indicates that the worst case approxima- back-off for G.fast. In Fig. 6, 7 and 8, the dashed lines are tions presented there can be applied with small changes for data rates for overlapped spectrum while the solid lines are this application. Therefore, the following approximations are for crosstalk avoidance. Each figure shows downstream (DS) used for the worst case couplings and upstream (US) rates for DSL and G.fast. Simulation conditions are the 0.5mm PE quad cable with f 1.5 |h (f)|2 = N 0.6 10−44/10 (11) up to 30 pairs according to [11], a colored background noise NEXTsum dp dist f 0 of −140 dBm/Hz below 30 MHz and −150 dBm/Hz above f 0.75 30 MHz according to [12] and ∆SNR = 2 dB. Power back- |h (f)|2 = N 0.6 10−44/10 (12) NEXTsum cpe dist f off settings for G.fast use ψnmax = −120 dBm/Hz. 0 2 0.6 f davg −39/10 |hFEXTsum(f)| = N 10 (13) 4.1. FTTdp Scenario dist f 1km 0 with f0 =1 MHz and a numberof Ndist disturbers and the av- G.Fast DS 600 erage disturber line length davg. Fig. 5 shows the comparison G.Fast US between these approximations and the crosstalk model. VDSL2 DS 400 VDSL2 US

−20 Rate/Mbit/s 200 −30 DP NEXT model

TF/dB CPE NEXT model 0 −40 0 100 200 300 400 500 600 700 DP NEXT apx line length/m −50 CPE NEXT apx

0 5 10 15 20 25 30 Fig. 6. Rate vs. reach curves for G.fast from the DP Frequency/MHz For the FTTdp scenario, a cable length of 300 m between Fig. 5. Worst case NEXT and FEXT for 0.5mm PE line VDSL2 street cabinet and G.fast DP is assumed. The ﬁnal

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drop between DP and the CPEs is assumed to be random 5. CONCLUSION uniformly distributed between 10 and 400 m. The VDSL2 and G.fast lines are randomly placed within the binder. The For FTTdp, the overlapped spectrum approach gives a signif- scenarios of interest are crosstalk avoidance with G.fast start- icant gain, compared to crosstalk avoidance. The NEXT be- ing at 23 MHz, and overlapped spectrum applying iterative tween VDSL2 and G.fast is attenuated through the line length water-ﬁlling. Fig. 6 shows the resulting data rates. The max. between cabinet and DP. FEXT from G.fast is small, because downstream gain for G.fast using the overlapped spectrum is the transmit power of G.fast is small. about 120 Mbit/s with an mean gain of 75 Mbit/s compared to For co-located scenarios, crosstalk avoidance causes per- crosstalk avoidance. formance losses for VDSL2, especially in upstream direction. However, there is a substantial rate loss for the legacy This is maintained by the proposed power back-off, which al- lines due to alien crosstalk, as can be seen in the lower right lows to adjust the performanceloss. Fig. 8 indicates that very of Fig. 6. With power back-off, the rate vs. reach curves long lines may beneﬁt from switching to VDSL2 rather than as shown in Fig. 7 are achieved. The downstream rate gain using long reach G.fast with the selected power back-off. is still up to 80 Mbit/s and 50 Mbit/s on average, but with a Overlapped spectrum still outperforms crosstalk avoid- small loss for legacy lines. ance, and, in the worst case, converges to the crosstalk avoidance rates with conservative power back-off settings.

G.Fast DS 600 G.Fast US REFERENCES VDSL2 DS [1] ITU-T Rec. G.9701, “Fast Access to Subscriber Terminals - 400 VDSL2 US Physical layer specification,” 2013. [2] ITU-T Rec. G.993.2, “Very high speed digital subscriber line Rate/Mbit/s 200 transceivers 3 (VDSL2),” 2006. [3] F. Mazzenga, M. Petracca, F. Vatalaro, R. Giuliano, and G. Ciccarella, “Coexistence of fttc and fttdp network architec- 0 0 100 200 300 400 500 600 700 tures in different vdsl2 scenarios,” Transactions on Emerging line length/m Telecommunications Technologies, 2014. [4] J. Cioffi, “A multicarrier primer,” ANSI T1E1, vol. 4, pp. 91– Fig. 7. G.fast from the DP with power back-off 157, 1991. [5] R. Strobel, R. Stolle, and W. Utschick, “Wideband Model- ing of Twisted-Pair Cables for MIMO Applications,” in IEEE 4.2. Multi-Mode Cabinet Scenario Globecom 2013 - Symposium on Selected Areas in Communi- cations (GC13 SAC), 2013. In the multi-mode cabinet scenario, G.fast and VDSL2 lines [6] ITU-T Rec. G.9700, “Fast access to subscriber terminals start at the same point which is the worst case in terms of (FAST) - Power spectral density specification,” 2013. alien NEXT. The line length from the cabinet to the CPEs is [7] J.-S. Pang, G. Scutari, D. P. Palomar, and F. Facchinei, “De- uniformly distributed between 10 m and 700 m. Long range sign of cognitive radio systems under temperature-interference G.fast as described in Sec. 2.4 is used for this scenario, be- constraints: A variational inequality approach,” Signal Pro- cause standard G.fast doesn’t support lines longer than 400 m. cessing, IEEE Transactions on, vol. 58, no. 6, pp. 3251–3271, 2010. 600 G.Fast DS [8] W. Yu, W. Rhee, S. Boyd, and J. Cioffi, “Iterative water-filling G.Fast US for Gaussian vector multiple-access channels,” IEEE Trans- VDSL2 DS actions on Information Theory, vol. 50, no. 1, pp. 145–152, 400 VDSL2 US 2004. [9] G. Scutari, D. P. Palomar, and S. Barbarossa, “Asynchronous

Rate/Mbit/s 200 iterative water-ﬁlling for Gaussian frequency-selective interference channels,” IEEE Transactions on Information Theory, vol. 54, no. 7, pp. 2868–2878, 2008. 0 0 100 200 300 400 500 600 700 [10] ETSI Standard, “Ts 101 270-1,” Transmission and Multiplex- line length/m ing (TM); Access transmission systems on metallic access cables; Very high speed Digital Subscriber Line (VDSL); Part 1: Functional requirements, 2003. Fig. 8. G.fast and VDSL2 FTTC with power back-off [11] Broadband Forum, “TR-285 Broadband Coppera Cable Mod- Power back-off avoids disturbance of the legacy lines and els,” 2015. still gives some gain for G.fast lines, which is max. 50 Mbit/s [12] L. Humphrey, “On VHF bachground noise assumptions for in downstream, as Fig. 8 shows. G:FAST,” 2013.

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