Improved Active Power Filter Performance

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 2, DEC 2015 687 Improved Active Power Filter Performance for Renewable Power Generation Systems SHAIK GOUSEPASHA, M.Tech, JNTU HYDERABAD, Ph.No:9640275218 [email protected],

I. INTRODUCTION Abstract—An active power filter implemented with a four-leg voltage-source inverter using a predictive control scheme is presented. The use of a four-leg voltage-source inverter allows the com- ENEWABLE generation affects power quality due to its pensation of current harmonic components, as well as unbalanced current generated by single-phase nonlinear loads. A detailed yet Rnonlinearity, since solar generation plants and wind power generators simple mathematical model of the active power filter, including the must be connected to the grid through high-power static PWM converters [1]. effect of the equivalent power system impedance, is derived and used The nonuniform nature of power generation directly affects voltage regulation to design the predictive control algorithm. The compensation and creates volt-age distortion in power systems. This new scenario in power performance of the proposed active power filter and the associ-ated distribution systems will require more sophisticated compensa-tion control scheme under steady state and transient operating conditions techniques. is demonstrated through simulations and experimental results. Although active power filters implemented with three-phase four- Index Terms—Active power filter, current control, four-leg con-verters, leg voltage-source inverters (4L-VSI) have already been presented in predictive control. the technical literature [2]–[6], the primary contri-bution of this paper is a predictive control algorithm designed and implemented specifically for this application. Traditionally, active power filters have been controlled using pretuned con-trollers, such as PI-type or NOMENCLATURE adaptive, for the current as well as for the dc-voltage loops [7], [8]. PI controllers must be de-signed based on the equivalent linear AC Alternating current. model, while predictive controllers use the nonlinear model, which is closer to real op-erating conditions. An accurate model obtained dc Direct current. using predictive controllers improves the performance of the active PWM Pulse width modulation. power filter, especially during transient operating conditions, because PC Predictive controller. it can quickly follow the current-reference signal while maintaining a constant dc-voltage. PLL Phase-locked-loop. v So far, implementations of predictive control in power con-verters d c dc-voltage. have been used mainly in induction motor drives [9]–[16]. In the case v T s System voltage vector [vs u vs v vs w ] . of motor drive applications, predictive control represents a very T intuitive control scheme that han-dles multivariable characteristics, is System current vector [is u is v is w ] . i T simplifies the treatment of dead-time compensations, and permits L Load current vector [iL u iL v iL w ] . pulse-width modulator replacement. However, these kinds of T vo VSI output voltage vector [vo u vo v vo w ] . applications present dis-advantages related to oscillations and i T instability created from unknown load parameters [15]. One o VSI output current vector [io u io v io w ] . advantage of the proposed algorithm is that it fits well in active ∗ ∗ ∗ ∗ T io Reference current vector [io u io v io w ] . power filter applica-tions, since the power converter output i n Neutral current. parameters are well known [17]. These output parameters are obtained from the converter output ripple filter and the power system Lf Filter inductance. equivalent impedance. The converter output ripple filter is part of the R active power filter design and the power system impedance is f Filter resistance. obtained from well-known standard procedures [18], [19]. In the case of unknown system impedance parameters, an estimation method can be used to derive an accurate R–L equivalent impedance model of the system [20]. Manuscript received July 4, 2012; revised October 13, 2012 and December 27, 2012; This paper presents the mathematical model of the 4L-VSI and the accepted March 21, 2013. Date of current version August 20, 2013. This work was principles of operation of the proposed predictive control scheme, supported in part by the Chilean Fund for Scientific and Tech-nological Development including the design procedure. The complete descrip-tion of the (FONDECYT) through project 1110592, in part by the Basal Project FB 0821, and in part by the CONICYT Initiation into Research 2012 11121492 Project. Recommended selected current reference generator implemented in the active power for publication by Associate Editor filter is also presented. Finally, the pro-posed active power filter and the effectiveness of the associated M. Malinowski. P. Acuna˜ and L. Moran´ are with the Department of Electrical Engineering, Universidad de Concepcion,´ Concepcion´ 4030000, Chile (e-mail: pabloacuna@ udec.cl; [email protected]). M. Rivera is with the Department of Industrial Technologies, Universidad de Talca, Curico´ 685, Chile (e-mail: [email protected]). J. Dixon is with the Department of Electrical Engineering, Pontificia Univer-sidad Catolica´ de Chile, Santiago 340, Chile (e-mail: [email protected]). J. Rodriguez is with the Department of Electronics Engineering, Universidad Tecnica´ Federico Santa Mar´ı a, Valpara´ ı so 1680, Chile (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.2013.2257854

Fig. 1. Stand-alone hybrid power generation system with a shunt active power filter.

Fig. 3. Two-level four-leg PWM-VSI topology. Fig. 2. Three-phase equivalent circuit of the proposed shunt active power filter. The voltage in any leg x of the converter, measured from the neutral point (n), can be expressed in terms of switching states, as follows: control scheme compensation are demonstrated through simula-tion and validated with experimental results obtained in a 2 kVA laboratory prototype. vx n = Sx − Sn vd c , x = u, v, w, n. (1) The mathematical model of the filter derived from the equiv-alent circuit shown in Fig. 2 is II. FOUR-LEG CONVERTER MODEL d i Fig. 1 shows the configuration of a typical power distribution o system with renewable power generation. It consists of various types vo = vx n − Re q io − Le q dt (2) of power generation units and different types of loads. Renewable sources, such as wind and sunlight, are typically used to generate where Re q and Le q are the 4L-VSI output parameters expressed as electricity for residential users and small industries. Both types of Thevenin impedances at the converter output terminals Ze q . Therefore, power generation use ac/ac and dc/ac static PWM converters for the Thevenin equivalent impedance is determined by a series connection voltage conversion and battery banks for long-term energy storage. of the ripple filter impedance Z and a parallel arrangement between the These converters perform maximum power point tracking to extract f the maximum energy possible from wind and sun. The electrical system equivalent impedance Zs and the load impedance ZL energy consumption behavior is random and unpredictable, and therefore, it may be single- or three-phase, balanced or unbalanced, Zs ZL and linear or nonlinear. An active power filter is connected in parallel at the point of common coupling to compensate current harmonics, Ze q = Zs + ZL + Zf ≈ Zs + Zf . (3) current unbalance, and reactive power. It is composed by an For this model, it is assumed that ZL Zs , that the resistive part of the electrolytic capacitor, a four-leg PWM converter, and a first-order system’s equivalent impedance is neglected, and that the series reactance output ripple filter, as shown in Fig. 2. This circuit considers the is in the range of 3–7% p.u., which is an acceptable approximation of the power system equivalent impedance Zs , the converter output ripple real system. Finally, in (2) filter impedance Zf , and the load impedance ZL . Re q = Rf and Le q = Ls + Lf . The four-leg PWM converter topology is shown in Fig. 3. This converter topology is similar to the conventional three-phase converter with the fourth leg connected to the neutral bus of the system. The fourth III. DIGITAL PREDICTIVE CURRENT CONTROL leg increases switching states from 8 (23 ) to 16 (24 ), improving control flexibility and output voltage quality [21], and is suitable for current The block diagram of the proposed digital predictive current unbalanced compensation. control scheme is shown in Fig. 4. This control scheme is basi-cally an optimization algorithm and, therefore, it has to be im-plemented in a microprocessor. Consequently, the analysis has to be developed using discrete mathematics in order to consider additional restrictions such as time delays and approximations 689

2 [k + 1] − io w [k + 1]) Fig. 4. Proposed predictive digital current control block diagram. 2 + (i∗ [k + 1] i [k + 1]) . (6) − o n o n ∗ The output current (io ) is equal to the reference (i o ) when g = 0. Therefore, the optimization goal of the cost function is to achieve a g value close to zero. The voltage vector vx N that minimizes the cost function is chosen and then applied at the next sampling state. During each sampling state, the switching state that generates the minimum value of g is selected from the 16 possible function values. The algorithm selects the switching state that produces this minimal value and applies it to the converter during the k + 1 state. IV. CURRENT dx x[k + 1] − x REFERENCE GENERATION dt ≈ Ts A dq-based current reference generator scheme is used to ob-tain the active power filter current reference signals. This scheme presents a fast and accurate signal tracking capability. This characteristic avoids voltage fluctuations that deteriorate the current reference signal affecting compensation performance [28]. The current reference signals are obtained from the corresponding load currents as shown in Fig. 5. This module calculates the reference signal currents required by the converter to compensate reactive power, current harmonic, and current imbalance. The displacement power factor (sin φ(L ) ) and the maximum total harmonic distortion of the load (THD(L ) ) defines the relation- (4) ships between the apparent power required by the active power filter, with respect to the load, as shown The 16 possible output 3) Cost Function Optimization: current predicted values can be In order to select the optimal ob-tained from (2) and (4) as switching state that must be applied to the power converter, the io [k + 1] = Leq (vx n [k] − vo [k]) + T 16 predicted values obtained for io s [k + 1] are compared with the reference using a cost function g, as follows: (5) As shown in (5), in order to predict the output current io at ∗ the instant (k + 1), the input g[k + 1] = (i o u [k voltage value vo and the converter output voltage vx N , + 1] − are required. The algorithm calculates all 16 values i [k associated with the possible o u combinations that the 2 state variables can achieve. + 1]) ∗ S + (i o v A P F sin φ(L ) + T H D

[k + 1] SL = 1 + THD( − io v where the value of THD(L ) includes the maximum [k + compensable harmonic current, defined as double the sampling 2 frequency fs . The frequency of the 1]) maximum current harmonic component that can be compensated is equal to one half of the converter switching frequency. The dq-based scheme operates in a rotating reference frame; therefore, the measured currents must be multiplied by the sin(wt) and cos(wt) signals. By using dq- transformation, the d current component is synchronized with the corresponding phase-to-neutral system voltage, and the q current component is phase-shifted by 90◦. The sin(wt) and cos(wt) synchronized reference signals are obtained from a synchronous reference frame (SRF) PLL [29]. The SRF-PLL generates a pure sinusoidal waveform even when the system voltage is severely 690 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 2, , DEC 2015

Fig. 5. dq-based current reference generator block diagram. distorted. Tracking errors are eliminated, since SRF-PLLs are designed to avoid phase voltage unbalancing, harmonics (i.e., less than 5% and 3% in fifth and seventh, respectively), and off-set caused by the nonlinear load conditions and measurement errors [30]. Equation (8) shows the relationship between the real currents iL x (t) (x = u, v, w) and the associated dq components (id and iq )

1 1 1

− − i d 2 sin ωt cos ωt 2 2 = q − √3 √3 . i 3 cos ωt sin ωt (8 2 )

A low-pass filter (LFP) extracts the dc component of the phase responding third-order harmonic content, and system current imbalance (with

currents id to generate the harmonic reference components −id . respect to positive sequence of the system current, is , 1 ). are The reactive reference components of the phase-currents obtained by phase-shifting the corresponding ac and dc compo- ◦ nents of iq by 180 . In order to keep the dc-voltage constant, ◦ the amplitude of the converter reference current must be mod- from the phase-currents, phase-shifted by 180 , as shown next

ified by adding an active power reference signal ie with the d-component, as will be explained in Section IV-A. The re- i∗ = (i + i + i ) . (10) ∗ ∗ sulting signals id and iq are transformed back to a three-phase o n − L u L v L w system by applying the inverse Park and Clark transformation, as shown in (9). The cutoff frequency of the LPF used in this One of the major advantages of the dq-based current refer- paper is 20 Hz ence generator scheme is that it allows the implementation of 1 1 a linear controller in the dc-voltage control loop. However, one 0 √ 2 i m p o r t a n t d i s a d v a n t a g e o f t h e dq- b a s e d c u r r e n t r e f e r e n c e f r a m e ∗ algorithm used to generate the current reference is that a second- io

u 2 1 1 √ 3 order harmonic component is generated in id and iq under un-

∗ = i o v √ − 2 2 balanced operating conditions. The amplitude of this harmonic 3 2

io∗w √ depends on the percent of unbalanced load current (expressed as 1 1 3 the relationship between the negative sequence current iL ,2 and

√ 2 − 2 − 2 the positive sequence current iL ,1 ). The second-order harmonic

cannot be removed from id and iq , and therefore generates a 1 0 0 i0 third harmonic in the reference current when it is converted sin 0 ωt cos ωt i∗ . (9) back to abc frame [31]. Fig. 6 shows the percent of system cur- − d × 0 cos ωt sin ωt i rent imbalance and the percent of third harmonic system current,

in function of the percent of load current imbalance. Since the The current that flows through the neutral of the load is com- load current does not have a third harmonic, the one generated pensated by injecting the same instantaneous value obtained by the active power filter flows to the power system. ˜ ACUNA et al.: IMPROVED ACTIVE POWER FILTER PERFORMANCE FOR RENEWABLE POWER GENERATION SYSTEMS 691

TABLE I SPECIFICATION PARAMETERS

Fig. 7. DC-voltage control block diagram.

A. DC-Voltage Control The dc-voltage converter is controlled with a traditional PI controller. This is an important issue in the evaluation, since

a the cost function (6) is designed using only current references, Note: Vbase = 55 V and Sbase = 1 kVA. in order to avoid the use of weighting factors. Generally, these weighting factors are obtained experimentally, and they are not well defined when different operating conditions are required. V. SIMULATED RESULTS Additionally, the slow dynamic response of the voltage across the electrolytic capacitor does not affect the current transient A simulation model for the three-phase four-leg PWM con- response. For this reason, the PI controller represents a simple verter with the parameters shown in Table I has been developed and effective alternative for the dc-voltage control. using MATLAB-Simulink. The objective is to verify the current The dc-voltage remains constant (with a minimum value of harmonic compensation effectiveness of the proposed control √ 6 vs(rm s) ) until the active power absorbed by the converter scheme under different operating conditions. A six-pulse rec- decreases to a level where it is unable to compensate for its tifier was used as a nonlinear load. The proposed predictive losses. The active power absorbed by the converter is controlled control algorithm was programmed using an S-function block by adjusting the amplitude of the active power reference signal that allows simulation of a discrete model that can be easily implemented in a real-time interface (RTI) on the dSPACE DS1103 ie , which is in phase with each phase voltage. In the block R&D control board. Simulations were performed considering a diagram shown in Fig. 5, the dc-voltage vd c is measured and ∗ then compared with a constant reference value vd c . The error 20 [μs] of sample time. In the simulated results shown in Fig. 8, the active filter starts (e) is processed by a PI controller, with two gains, Kp and Ti . Both gains are calculated according to the dynamic response to compensate at t = t1 . At this time, the active power filter in- requirement. Fig. 7 shows that the output of the PI controller is jects an output current io u to compensate current harmonic components, current unbalanced, and neutral current simultaneously. fed to the dc-voltage transfer function Gs , which is represented During compensation, the system currents is show sinusoidal by a first-order system (11) wave form, with low total harm onic disto rtion (TH D = 3.93 %).

√ At t = t2 , a three-phase balanced load step change is generated v d c 3 Kp vs 2 . from 0.6 to 1.0 p.u. The compensated system currents remain ∗ G (s) = ie = 2 Cd c v (11) sinusoidal despite the change in the load current magnitude. Fi- d c nally, at t = t3 , a single-phase load step change is introduced in The equivalent closed-loop transfer function of the given sys- phase u from 1.0 to 1.3 p.u., which is equivalent to an 11% cur- tem with a PI controller (12) is shown in (13) rent imbalance. As expected on the load side, a neutral current flows throu gh the neutr al cond uctor

(iL

n ), but on the sourc e side,

no neutral current is observed (is n ). Simulated results show that C(s) = Kp 1 + 1 (12) ·T s i currents. Additionally,the proposed Fig. 8 controlshows scheme that effectively the dc-voltage eliminates unbalancedremains sta ble thr ou gh out the wh ole act ive po we r filt er op era tio

2 n. v ω n d c = a · (s + a) . (13) 2 2 ie s + 2ζ ωn · s + ωn VI. EXPERIMENTAL RESULTS Since the time response of the dc-voltage control loop does The compensation effectiveness of the active power filter is not need to be fast, a damping factor ζ = 1 and a natural angular corroborated in a 2 kVA experimental setup. A six-pulse rec- speed ωn = 2π · 100 rad/s are used to obtain a critically damped tifier was selected as a nonlinear load in order to verify the ef- response with minimal voltage oscillation. The corresponding fectiveness of the current harmonic compensation. A step load integral time Ti = 1/a (13) and proportional gain Kp can be change was applied to evaluate the transient response of the dc- calculated as voltage loop. Finally, an unbalanced load was used to validate the perf orm ance of the neut ral curr ent com pens atio n. Bec ause √ the ζ = 3 Kp vs 2Ti (14) I/O board, all I/O Simulink blocks used in the simulations are ∗ 8 Cd c v d c 100% compatible with the dSPACE system capabilities. The com plete contr ol loop is exec uted by the contr oller ever y 20 μs, √ . K v ωn = 3 p s 2 (15) while the selected switching state is available at 16 μs. An aver- ∗ 2 Cd c vd c Ti age switching frequency of 4.64 kHz is obtained. Fig. 9 shows 692 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 2, , DEC 2015

Fig. 8. Simulation results: (a) Grid voltages, (b) Grid Currents (c) Unbalanced load currents, (d) Inverter Currents.

the transient response of the compensation scheme. Fig. 9(a) shows that the line current becomes sinusoidal when the active power filter starts compensation, and the dc-voltage behaves as expected. Experimental results shown in Fig. 9(b) indicate that the total harmonic distortion of the line current (THDi ) is reduced from 27.09% to 4.54%. This is a consequence of the good tracking characteristic of the current references, as shown in Fig. 9(d). In Fig. 10, the transient response of the active power filter under a step load change is shown. The line currents remain sinusoidal and the dc- voltage returns to its reference with a typ-ical transient response of an underdamped second-order system (maximum overshoot of 5% and two cycles of settling time). In this case, a step load change is applied from 0.6 to 1.0 p.u. Finally, the load connected to phase u was increased from 1.0 to 1.3 p.u. The corresponding waveforms are shown in Fig. 11. Fig. 11(a) shows that the active filter is able to Fig. 9. Experimental transient response after APF connection. (a) Load Cur-rent i , active compensate the current in the neutral conductor with fast transient L u response. power filter current io u , dc-voltage converter vd c , and system current is u . Associated frequency spectrum. (c) Voltage and system waveforms, vs u and i s u , is v , is w . (d) Current ∗ reference signals i o u , and active power filter current io u (tracking characteristic).

Moreover, Fig. 11(b) shows that the system neutral current io n is effectively compensated and eliminated, and system currents remain balanced even if an 11% current imbalance is applied. ˜ ACUNA et al.: IMPROVED ACTIVE POWER FILTER PERFORMANCE FOR RENEWABLE POWER GENERATION SYSTEMS 693 capability, and transient response. Simulated and experimental results have proved that the proposed predictive control algo-rithm is a good alternative to classical linear control methods. The predictive current control algorithm is a stable and robust solution. Simulated and experimental results have shown the compensation effectiveness of the proposed active power filter.

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Advantages of the proposed scheme are related to its simplicity, modeling, and implementation. The use of a predictive control algorithm for the converter current loop proved to be an effective solution for active power filter applications, improving current tracking 694 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 2, , DEC 2015

controlled voltage disturbance,” IEEE Trans. Power Electron., vol. 17, no. 2, pp. Marco Rivera (S’09–M’11) received the B.Sc. de-gree in 207–215, Mar. 2002. electronics engineering and M.Sc. degree in electrical engineering from the Universidad de Con-cepcion,´ Chile, [21] S. Ali, M. Kazmierkowski, “PWM voltage and current control of four-leg VSI,” in 2007 and 2008, respectively and the Ph.D. degree from presented at the ISIE, Pretoria, South Africa, vol. 1, pp. 196–201, Jul. 1998 the Department of Electron-ics Engineering, Universidad Tecnica´ Federico Santa Mar´ı a, Valpara´ ı so, Chile, in 2011, with a scholarship from the Chilean Research Fund [22] S. Kouro, P. Cortes, R. Vargas, U. Ammann, and J. Rodriguez, “Model predictive CONICYT. control—A simple and powerful method to control power con-verters,” IEEE During 2011 and 2012, he was at a Post Doctoral Trans. Ind. Electron., vol. 56, no. 6, pp. 1826–1838, Jun. 2009. position and as a part-time Professor of Digital Signal Processors and Industrial Electronics at Universidad [23] D. Quevedo, R. Aguilera, M. Perez, P. Cortes, and R. Lizana, “Model predictive Tecnica´ Federico Santa Mar´ı a, and currently he is a Professor in Universidad de Talca, control of an AFE rectifier with dynamic references,” IEEE Trans. Power Chile. His research interests include matrix converters, predictive and digital controls for Electron., vol. 27, no. 7, pp. 3128–3136, Jul. 2012. high-power drives, four-leg converters, renewable en-ergies, and development of high performance control platforms based on field-programmable gate arrays. [24] Z. Shen, X. Chang, W. Wang, X. Tan, N. Yan, and H. Min, “Predictive digital current control of single-inductor multiple-output converters in CCM with low cross regulation,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 1917–1925, Apr. 2012. [25] M. Rivera, C. Rojas, J. Rodriidguez, P. Wheeler, B. Wu, and J. Espinoza, “Predictive current control with input filter resonance mitigation for a direct matrix converter,” IEEE Trans. Power Electron., vol. 26, no. 10, pp. 2794–2803, Oct. 2011. Juan Dixon (M’90–SM’95) received the B.S. de-gree in [26] M. Preindl and S. Bolognani, “Model predictive direct speed control with finite electrical engineering from the Universidad de Chile, Santiago, Chile, in 1977, and the M.S.Eng. and Ph.D. control set of PMSM drive systems,” IEEE Trans. Power Electron., 2012. degrees from McGill University, Montreal, QC, Canada, in 1986 and 1988, respectively. [27] T. Geyer, “Computationally efficient model predictive direct torque con-trol,” In 1976, he was with the State Transportation IEEE Trans. Power Electron., vol. 26, no. 10, pp. 2804–2816, Oct. 2011. Company in charge of trolleybuses operation. In 1977 and 1978, he was with the Chilean Railways Company. Since 1979, he has been with the Depart-ment of Electrical [28] M. I. M. Montero, E. R. Cadaval, and F. B. Gonzalez, “Comparison of control Engineering, Pontificia Universi-dad Catolica´ de Chile, strategies for shunt active power filters in three-phase four-wire systems,” IEEE Santiago, where he is currently Trans. Power Electron., vol. 22, no. 1, pp. 229–236, Jan. 2007. a Professor. He has presented more than 70 works in international conferences and has published more than 30 papers related with power electronics in IEEE Transactions and [29] S.-K. Chung, “A phase tracking system for three phase utility interface inverters,” IEE proceedings. His research interests include electric trac-tion, power converters, IEEE Trans. Power Electron., vol. 15, no. 3, pp. 431–438, May 2000. PWM rectifiers, active power filters, power-factor com-pensators, multilevel, and multistage converters. He has consulting work related with trolleybuses, traction [30] M. Karimi-Ghartemani, S. Khajehoddin, P. Jain, A. Bakhshai, and M. Mojiri, substations, machine drives, hybrid electric vehicles, and electric railways. He has created an electric vehicle laboratory where he has built state-of-the-art vehicles using “Addressing DC component in PLL and notch filter algo-rithms,” IEEE brushless dc machines with ultracapacitors and high specific-energy batteries. Trans. Power Electron., vol. 27, no. 1, pp. 78–86, Jan. 2012. [31] L. Czarnecki, “On some misinterpretations of the instantaneous reactive power p-q theory,” IEEE Trans. Power Electron., vol. 19, no. 3, pp. 828– 836, May 2004.

Jose´ Rodriguez (M’81–SM’94–F’10) received the Engineering degree in electrical engineering from the Pablo Acuna˜ (M’12) received the B.S. degree and the Universidad Tecnica´ Federico Santa Mar´ı a, in Valpara Graduate degree in electronics engineering in 2004 and ´ı so, Chile, in 1977, and the Dr.-Ing. de-gree in electrical 2007, respectively, from the University of Con-cepcion,´ engineering from the University of Erlangen, Erlangen, Concepcion,´ Chile, where he is currently working toward Germany, in 1985. the Ph.D. degree. He has been with the Department of Electron-ics His current research interests include the areas of three- Engineering, Universidad Tecnica´ Federico Santa Mar´ı a, phase ac/dc static-power converters, and active power since 1977, where he is currently a Full Pro-fessor and filters applications using field programmable gate arrays Rector. He has coauthored more than 350 journal and and microcontroller systems-on-a-chip. conference papers. His main research interests include multilevel inverters, new converter topologies, control of power converters, and adjustable-speed drives. Dr. Rodriguez is member of the Chilean Academy of Engineering.

Luis Moran´ (F’05) received the Ph.D. degree from Concordia University, Montreal, QC, Canada, in 1990. Since 1990, he has been with the Department of Electrical Engineering University of Concepcion,´ Concepcion,´ where he is a Professor. He has writ-ten and published more than 30 papers in active power filters and static Var compensators in IEEE Transactions. His main areas of interests are in ac drives, power quality, active power filters, FACTS, and power protection systems. Dr. Moran´ is the principal author of the paper that got the IEEE Outstanding Paper Award from the Industrial Electronics Society for the best paper pub-lished in the IEEE TRANSACTION ON INDUSTRIAL ELECTRONICS during 1995, and the coauthor of the paper that was awarded in 2002 by the IAS Static Power Converter Committee.