An Effective Switching Algorithm for Single Phase Matrix Converter in Induction Heating Applications

Article An Effective Switching Algorithm for Single Phase Matrix Converter in Induction Heating Applications

Anand Kumar 1,* ID , Pradip Kumar Sadhu 1, Dusmanta Kumar Mohanta 2 and Maddikara Jaya Bharata Reddy 3

1 Department of Electrical Engineering, Indian Institute of Technology, Indian School of Mines, Dhanbad 826004, India; [email protected] 2 Department of Electrical and Electronics Engineering, Birla Institute of Technology, Mesra, Ranchi 835215, India; [email protected] 3 Department of Electrical and Electronics Engineering, National Institute of Technology, Trichy 620015, India; [email protected] * Correspondence: [email protected]; Tel.: +91-977-193-0334

Received: 26 June 2018; Accepted: 16 August 2018; Published: 18 August 2018

Abstract: Prevalent converters for induction heating (IH) applications employ two-stage conversion for generating high-frequency magnetic field, namely, AC to DC and then DC to high-frequency AC (HFAC). This research embarks upon a direct conversion of utility AC to high frequency AC with the design of a single-phase matrix converter (SPMC) as a resonant converter using a modified switching technique for IH application. The efficacy of the proposed approach is validated through different attributes such as unity power factor, sinusoidal input current and low total harmonic distortion (THD). The developed prototype-embedded system has high pragmatic deployment potential owing to its cost effectiveness using Arduino mega 2560 and high voltage/current as well as low switching time IXRH40N120 insulated-gate bipolar transistor (IGBT). Different results of the prototype-embedded system for IH application have been verified using Matlab Simulink environment to corroborate its efficacy.

Keywords: Arduino 2560; bi-directional switches; induction heating; resonant converter; resonant frequency and single-phase matrix converter

1. Introduction The recent trend shows that the industrial as well as domestic induction heating (IH) has become extremely popular because of its unique advantages such as higher efficiency, reduced heating time and environmental friendliness. To implement the IH for different appliances, a high-frequency (HF) alternating electromotive force (e.m.f.) is required that typically lies between 20 KHz to 100 KHz depending on the type of applications such as brazing, melting processes and for domestic cooking [1–4]. In the last few decades, various new topologies of HF resonant inverter have been proposed to generate H.F alternating e.m.f. [5]. However, ongoing research and development are entering into a new phase ensuring cost-effectiveness [6,7], increased cooling capabilities [8] and high efficiency [9,10] within the field of electrical power conversion and process. The conventional IH system follows two stages: (a) rectification [11,12] and (b) HF resonant inverter operation [13–15]. In the first stage, DC power is obtained using a full bridge diode rectifier. After rectification, a small value of inductor and capacitor is connected to obtain DC with ripple content ensuring unity input power factor [12]. This high ripple DC link voltage acts as a power supply for the HF resonant inverter (Figure1a). This is also an indirect method for the conversion of supply/grid frequency to HFAC.

Electronics 2018, 7, 149; doi:10.3390/electronics7080149 www.mdpi.com/journal/electronics Electronics 2018, 7, 149 2 of 17 Electronics 2018, 7, x FOR PEER REVIEW 2 of 17

VariousVarious topologies topologies of of HF HF resonant resonant inverters inverter haves have been been developed developed such such as as the the half-bridge half-bridge series series resonantresonant inverter inverter (HB-SRI) (HB-SRI) [ 16[16]], full, full bridge bridge series series resonant resonant inverter inverter (FB-SRI) (FB-SRI) [17 [17]], single, single switch switch topologytopology [ 18[18]] etc. etc. Although, Although, these these converters converters have have been been designed designed using using insulated-gate insulated-gate bipolar bipolar transistortransistor (IGBT) (IGBT) because because of its of higher its current/voltagehigher current/voltage handling handling capability, capabil reducedity, control reduced complexity control andcomplexity less cost. Inand addition, less cost. IGBTs In shouldaddition, have IGBTs low reverseshould recoveryhave low time reverse and high recovery switching time frequency. and high Itswitching can be seen frequency. that owing It to can different be seen stages that inowing the conventional to different stages IH system, in the its conventional efﬁciency is automatically IH system, its reduced.efficiency Moreover, is automatically indirect reduced. method Moreover, of conversion indirect uses method reactive of energy conversion storage uses elements. reactive Thus,energy thisstorage method elements. makes theThus converter, this method bulky makes and unnecessary the converter losses bulky occur and acrossunnecessary the diode. losses Additionally, occur across two-stagethe diode. conversion Additionally of AC, two to HFAC-stage conversionconversion increases of AC to the HF numberAC conversion of components increases used the along number with of complexcomponents control used algorithms along (suchwith ascomplex Phase Lockedcontrol Loop algorithms (PLL), Proportional(such as Phase Integral Locked Derivative Loop (PID)(PLL), andProport Fuzzyional logic Integral controller) Derivative for obtaining (PID) high and powerFuzzy factor,logic controller) low total harmonic for obtaining distortion high (THD)powerat factor, the inputlow sidetotal andharmonic good power distortion quality (THD) [19, 20at ].the Overall, input side these and existing good topologiespower quality increase [19,20 the]. Overall, cost as well these asexisting complexity topolog of theies controller.increase the cost as well as complexity of the controller.

POT

IH coil

Iac Idc Idc Iac R0,L0 1 − φ AC Cr 50HZ

V0 Vdc Vm t t t

(a)

Iac Iac

Cr L0

r.m.s 230V R0 50Hz

BIDIRECTIONAL SWITCH (S1,S2,S3,S4)

(b)

FigureFigure 1. 1.(a ()a Conventional) Conventional IH IH Topology Topology and and (b ()b Block) Block diagram diagram of of proposed proposed topology. topology.

Because of some preceding demerits, the popularity ofI0 an indirect method started decreasing and therefore, researchers started focussing on a direct method of AC conversion [21]. However, 230V Power Circuit of R0 L0 a lot of work has been done in this ﬁeldSPMC and has been widely used in several applications [22,23]. 50Hz Cr By employing direct AC–AC converters, reductions in both component count, as well as intermediate DC link reactive element, have been accomplished [24,25]. Matrix converters, Cycloconverter and AC voltage controllers are examples of direct AC–AC conversion. Recent research shows that with the evolution of these direct230V AC–AC converters and their various advanced control techniques, it is Proposed Controller possible to avail it in many applications such as traction system, industrial as well as domestic IH. Vg1 The matrix converters are12V newly advanced converters (AC–AC) that enables highS1a power,S4a,S3a, densityS2a and uC Isolation ZCD INT0 eliminates DC link components with improvedVg operational lifeCircuit [26]. Vg2 INT1 TLP250 S2b,S3b,S4b,S1b Currently, direct AC–AC conversion is being applied for IH applications [27–29]. All these converters use HB–SRI, which relies on four-quadrant equivalent switching devices that are combination of two anti-series IGBTs.Diode In [30], a SiC based AC to AC converter for domestic IH 12Vdc Rectifier

Electronics 2018, 7, 149 3 of 17 has been proposed, which uses solely four switches and only single stage energy conversion has been achieved. Moreover, single-phase matrix converter (SPMC) (which is the part of Matrix Converter) may also be used for IH applications [31]. Numerous studies have been done related to SPMC and its switching algorithm but because of its complicacy, less use of SPMC has been seen for IH application till date [21,32]. However, some authors have proposed a suitable switching algorithm ensuring unity power factor, commutation strategy and low THD for SPMC in IH applications [33,34]. But these switching algorithms seem too complicated. Therefore, in this article, to enhance potency and eliminate DC link components, a SPMC (direct AC–AC converter) is proposed using modified technique to make it appropriate for IH applications (Figure1b). This modified switching algorithm/technique requires only two pulse width modulation (PWM) signal to operate SPMC as a frequency changer or as resonant converter for IH applications and these pulses has been generated through embedded system. Due to requirement of less number of PWM signals, the overall cost and complexity of the controller reduces. The major contribution of this research is to develop an effective switching pattern such that SPMC works as a resonant converter and hence, can be applied for IH applications. In pursuance of this goal, switching frequency is kept higher than the resonant frequency so as to ensure zero voltage switching (ZVS). The high power factor, sinusoidal input current and low THD have been major achievements for improving its efficacy considerably. The proposed prototype system is in full agreement with recent developments in embedded technology Arduino mega 2560 and high voltage/current as well as low switching time IXRH40N120 IGBT (AppendixA, IXYS Corporation, Santa Clara, CA, USA). Another unique feature of the proposed technique based on embedded system is that it utilizes a single stage conversion of 50–60 Hz AC power to HFAC directly (i.e., as a frequency changer) without any intermediate DC link element using proposed switching algorithm and it has been congruous with simulated results. Using this technique, a higher frequency can be generated to meet the criteria of IH system. The rest of the article is divided into five sections. In Section2, the modified switching algorithm for SPMC as a resonant converter for IH application is extensively explained. Simulation results using modified switching algorithm for SPMC as a resonant converter for IH application has been done in MATLAB Simulink environment and is presented in Section3 for validation. Finally, the hardware/experimental results have been shown in Section4 which validates the simulation results and the main conclusion of the proposed article is given in Section5.

2. Single-Phase Matrix Converter and Its Modiﬁed Switching Technique for Induction Heating (IH) Applications

2.1. Single-Phase Matrix Converter (SPMC) In this presented work, modified switching technique has been developed for SPMC topology to generate high frequency directly from grid/supply frequency. SPMC topology was first invented by Zuckerberger in 1997. It consists of a matrix of input and output lines with four bidirectional switches which connect the 1-ø input to 1-ø output (Figure2a,b). Each bidirectional switch has the capability of conducting current as well as blocking voltage of both polarities simultaneously depending on the control signal. Generally, common emitter configuration is used for making bi-directional switches for SPMC. It is also referred to as 2 × 2 order matrix converter. Actually, SPMC can be operated in many types of converters such as controlled rectifier (AC–DC), inverter (DC–AC), a boost converter (DC–DC) and as a cycloconverter (AC–AC) [35]. However, in this article, only SPMC as a frequency changer or as resonant converter is discussed, which is appropriate for the IH applications. This section discusses how input frequency can be synthesized with the help of this SPMC configuration using modified switching technique and makes it essential for IH applications. Electronics 2018, 7, x FOR PEER REVIEW 2 of 17

Various topologies of HF resonant inverters have been developed such as the half-bridge series resonant inverter (HB-SRI) [16], full bridge series resonant inverter (FB-SRI) [17], single switch topologyElectronics 2018[18], 7 ,etc 149. Although, these converters have been designed using insulated-gate bipolar4 of 17 transistorElectronics (IGBT) 2018, 7, x becauseFOR PEER REVIEWof its higher current/voltage handling capability, reduced control4 of 19 complexity and less cost. In addition, IGBTs should have low reverse recovery time and high switching frequency. It can be seen that owing to different stagesSia in the conventionalSib IH system, its 1 Load

efficiency is automatically- reduced. Moreover, indirect method of conversion uses reactive energy ø AC Supply storage elements. Thus, thisS 1method Smakes2 the converter bulkyIGBT and unnecessary losses occur across (

R Diode , the diode. Additionally, two-stage conversionRL of AC to HFAC conversion increases the number of , components used along with complex controlRLC algorithms (such as Phase Locked Loop (PLL), , 50 Hz

S3 ) Proportional Integral Derivative (PID)S4 and Fuzzy logic controller)Diode for obtainingIGBT high power factor, low total harmonic distortion (THD) at the input side and good power quality [19,20]. Overall, these existing topologies increase the cost as well as complexityi=1,2,3 ,of4 the controller.

(a) (b)

POT FigureFigure 2. 2. (a()a) 1-ø1-ø matrixmatrix converterconverter topology topology and and ( b()b) bi-directional bi-directional switch switch (common (common emitter emitter configurationconfiguration mode). mode). IH coil Iac Idc dc ac 2.2. Proposed Switching Technique for SPMC Iin IH Applications I 2.2. Proposed Switching Technique for SPMC in IH Applications R0,L0 1 − φ AC Cr In the50 proposedHZ switching technique, only two PWM signal (i.e., Vg1 and Vg2) is required to In the proposed switching technique, only two PWM signal (i.e., Vg1 and Vg2) is required to operateoperate SPMC SPMC as as a resonanta resonant converter converter for for IH IH applications. applications Regarding. Regarding the the generation generation of of these these pulses, pulses, a controller has been designed which is based on embeddedV0 system. The IH system based on SPMC a controller has been designedVdc which is based on embedded system. The IH system based on SPMC Vm topology using proposed controller which generates pulses Vg1 and Vg2 is shown in Figure 3. In this topology using proposedt controller whicht generates pulses Vg1 and Vg2 is shown in Figure3. In this figure, the proposed controller comprises of zero crossing detector (ZCD), microcontrollert unit and ﬁgure, the proposed controller comprises of zero crossing detector (ZCD), microcontroller unit and isolation circuit. For generation of pulses, the AC voltage (230 Vr.m.s) is stepped down to 12 Vr.m.s. isolation circuit. For generation of pulses, the AC(a) voltage (230 Vr.m.s) is stepped down to 12 Vr.m.s. Subsequently, this stepped down AC is given to ZCD block that is used to synchronize the pulses Subsequently, this stepped down AC is given to ZCD block that is used to synchronize the pulses (Vg1 (Vg1 and Vg2) with the input AC supply.Iac Now this output ofIac ZCD is fed to the microcontroller Atmega and Vg2) with the input AC supply. Now this output of ZCD is fed to the microcontroller Atmega 2560, 2560, which detects the rising and falling edge of the pulse, i.e., Vg (generated from the ZCD). which detects the rising and falling edge of the pulse, i.e., Vg (generatedCr L0 from the ZCD). According to According to the program fed to the microcontroller, when it detects the rising edge of the pulse (Vg), the program fed to the microcontroller,r.m.s when it detects the rising edge of the pulse (Vg), it generates 230V R0 it generates the pulse (Vg1) of 10 ms (if the 100 Hz output of the converter is required). When it detects the pulse (Vg1) of 10 ms (if50Hz the 100 Hz output of the converter is required). When it detects the falling the falling edge, again it generates a pulse (Vg2) of 10 ms, but this pulse is in complete phase opposition edge, again it generates a pulse (Vg2) of 10BIDIRECTIONAL ms, but this pulse is in complete phase opposition from from previous pulses, shown in Figure 4b. For generating a higher frequency of the pulse, the time previous pulses, shown in Figure4b. For generatingSWITCH a higher frequency of the pulse, the time period period of the pulses should be considered(S 1,lessS2,S3,S 4dur) ing programming. The detailed explanation of of the pulses should be considered less during programming. The detailed explanation of prototype prototype implementation and the experimental results of this proposed controller regarding implementation and the experimental results of this(b) proposed controller regarding generation of pulses generation of pulses (Vg1 and Vg2) have been well explained in section 4. (Vg1 and Vg2Figure) have 1. been(a) Conventional well explained IH Topology in Section and4. (b) Block diagram of proposed topology.

I0 I0

230V Power Circuit of R0 L0 0 L0 230V Power CircuitSPMC of R 50Hz Cr SPMC 50Hz Cr

230V 230V Proposed Controller Proposed Controller Vg1 12V S1a,S4a,S3a,S2a 12V uC Isolation Vg1 ZCD uC INT0 S1a,S4a,S3a,S2a Vg IsolationCircuit Vg2 ZCD INT0INT1 Vg CircuitTLP 250 Vg2 S2b,S3b,S4b,S1b INT1 TLP250 S2b,S3b,S4b,S1b

Diode Diode 12Vdc Rectifier 12Vdc Rectifier

Figure 3. Proposed IH system using SPMC topology incorporated with proposed controller. Figure 3. Proposed IH system using SPMC topology incorporated with proposed controller. Now, with the help of these two pulses (Vg1 and Vg2), frequency synthesization has been well presented. The proposed configuration (power circuit of SPMC as in Figure 3) of SPMC as a frequency changer/resonant converter for IH applications is shown in Figure 4a–c which shows the waveform

Electronics 2018, 7, 149 5 of 17

Now, with the help of these two pulses (Vg1 and Vg2), frequency synthesization has been well presented. The proposed conﬁguration (power circuit of SPMC as in Figure3) of SPMC as a frequency changer/resonantElectronics 2018, 7, x FOR converter PEER REVIEW for IH applications is shown in Figure4a–c which shows the waveform5 of 19 of SPMC as a frequency changer (i.e., at 100 Hz output) and as a resonant converter (i.e., at high frequency of SPMC as a frequency changer (i.e., at 100 Hz output) and as a resonant converter (i.e., at high of output) using proposed control technique. In Figure4a, Vi is the input supply voltage, Ls and Cf frequency of output) using proposed control technique. In Figure 4a, Vi is the input supply voltage, are the input inductor and ﬁlter capacitor respectively, that are used to reduce the electromagnetic Ls and Cf are the input inductor and filter capacitor respectively, that are used to reduce the interference (EMI) effect and also prevent the HF component of voltage/current (generated from the electromagnetic interference (EMI) effect and also prevent the HF component of voltage/current load side). Four bidirectional switches are used and each of them is a combination of two IGBTs and (generated from the load side). Four bidirectional switches are used and each of them is a combination two diodes. It is already known that the resonant inverter works at a resonant frequency (Equation (1)). of two IGBTs and two diodes. It is already known that the resonant inverter works at a resonant However,frequency for (Equation the IH application, (1)). However, the switchingfor the IH frequencyapplication, of thethe resonantswitching inverter frequency should of the be resonant kept higher orinverter lower thanshould the be resonant kept higher frequency or lower to than ensure the zero resonant voltage frequency switching to ensure (ZVS) orzero zero voltage current switching switching (ZCS)(ZVS) conditions or zero current for reducing switching the ( switchingZCS) condition lossess acrossfor reducing switches the [36 switching]. In this work,losses ZVSacross condition switches has been[36]. achievedIn this work by, maintaining ZVS condition the has switching been achieved frequency by highermaintaining than the the resonant switching frequency. frequency To higher analyze thethan system the resonant behaviour, frequency. IH coil To and analyze its load the system can be behaviour, modelled asIH thecoil seriesand its equivalent load can be of modelledR0 and L0, whichas the isseries already equivalent shown of in R Figure0 and L40a., which In addition, is already the shown resonating in Figure capacitor 4a. In (addition,Cr) is connected the resonating in series withcapacitorR0 and (CrL) 0isto connected create the in series series resonance with R0 and condition. L0 to create the series resonance condition.

S1a D2 S2b D4 Vin

D1 D3 t Ls S1b S2a

Vout Sin ω t

m IH LOAD Cf t = V i V R0 L0 Cr Vg1 S3b S4a D6 D8 (S1a,S4a,S3a,S2a) t Vg2 D5 D7 S4b (S2b,S3b,S4b,S1b) S3a t 0 t1 t2 t3 t4 (a) (b)

Vin

Vout t

Vg1 (S1a,S4a,S3a,S2a) t Vg2 (S2b,S3b,S4b,S1b) t 0 (c)

Power circuit of SPMC and its output voltage waveform ( ) SPMC as a resonant converter FigureFigure 4.4. Power circuit of SPMC and its output voltage waveforma (a) SPMC as a resonant converter for (IH) application; (b) Output voltage waveform of SPMC as frequency changer operation (at 100 for (IH) application; (b) Output voltage waveform of SPMC as frequency changer operation (at 100 Hz Hz output) and (c) Output voltage waveform of SPMC as resonant converter (at HF output) using output) and (c) Output voltage waveform of SPMC as resonant converter (at HF output) using modiﬁed modified switching technique. switching technique. The operation of the SPMC using modified switching technique for IH application can be understood by using 4 modes of operation according to the polarity of the input voltage. Modes 1

Electronics 2018, 7, x FOR PEER REVIEW 6 of 19 and 2 are explained for the positive half cycle and Modes 3 and 4 are explained for the negative half cycle. ElectronicsMode2018 1 ,(70, 149< t < t1): In the positive half cycle, switches S1a, S4a, S2b, S3b are forward biased and6 of S 173a, S2a, S4b, S1b are reverse biased. Forward biased switches can be turned ON at any time between 0 to t1 by applying the pulses (Vg1 and Vg2). During this time 0 < t < t1, among the forward biased switches The operation of the SPMC using modiﬁed switching technique for IH application can be only two switches (i.e., S1a and S4a) are receiving PWM signal (Vg1) shown in Figure 4b. Owing to this, understood by using 4 modes of operation according to the polarity of the input voltage. Modes 1 and only S1a and S4a will be turned ON to create a path for the load current that is, 2 are explained for the positive half cycle and Modes 3 and 4 are explained for the negative half cycle. S1a → D1 → load → S4a → D7 shown in Figure 5a. Mode 1 (0 < t < t1): In the positive half cycle, switches S1a,S4a,S2b,S3b are forward biased and Mode 2 (t1 < t < t2): This mode is also for the positive half cycle. In this mode, since switches S1a S3a, S2a, S4b, S1b are reverse biased. Forward biased switches can be turned ON at any time between 4a g1 and0 to St1 areby applyingnot receiving thepulses PWM signal (Vg1 and (V V),g2 these). During switches this get time turned0 < t < OFF t1, among. Now in the this forward mode, biasedamong theswitches forward only biased two switches, switches only (i.e., twoS1a and switchesS4a) are (S2b receiving and S3b) are PWM receiving signal PWM (Vg1) shownsignal (V ing2 Figure) shown4b. in FigureOwing 4b to. Owing this, only to this,S1a Sand2b andS4a Swill3b will be be turned turned ON ON. to Now, create the a path for the the load load current current becomes that is, reverseS1a → i.e.,D1 → S23loadbb→→→ DS4a → loadD7 →shown S 35 → in DFigureshown5a. in Figure 5b.

S2b S1a S2b S2a S1a S1b S1b S2a

Io Io Vi(t) IH LOAD Vi(t) IH LOAD

S3b S3a S3b S3a S4a S4b S4a S4b

(a) (b)

FigureFigure 5. 5. PositivePositive mode mode of of operation (a) Mode 11 andand ((bb)) ModeMode 2.2.

Mode 3 (t2 < t < t3): In the negative half cycle, switches S1a, S4a, S2b, S3b are reverse biased and S3a, Mode 2 (t1 < t < t2): This mode is also for the positive half cycle. In this mode, since switches S2a, S4b, S1b are forward biased. Forward biased switches can be turned ON at any time between t2 to S1a and S4a are not receiving PWM signal (Vg1), these switches get turned OFF. Now in this mode, t3 by applying the pulses (Vg1 and Vg2). During this time t2 < t < t3, among the forward biased switches among the forward biased switches, only two switches (S2b and S3b) are receiving PWM signal (Vg2) only two switches (i.e., S3a and S2a) are receiving PWM signal (Vg1) shown in Figure 4b. Owing to this, shown in Figure4b. Owing to this, S2b and S3b will be turned ON. Now, the path for the load current only S3a and S2a will be turned ON to create a path for the load current that is, becomes reverse i.e., S2b → D3 → load → S3b → D5 shown in Figure5b. S3a → D6 → load → S2a → D4 shown in Figure 6a. Mode 3 (t2 < t < t3): In the negative half cycle, switches S1a,S4a,S2b,S3b are reverse biased and Mode 4 (t3 < t < t4): This mode is also for the negative half cycle. In this mode, since switches S3a S3a, S2a, S4b, S1b are forward biased. Forward biased switches can be turned ON at any time between and S2a are not receiving PWM signal (Vg1), these switches gets turned OFF. Now in this mode, among t2 to t3 by applying the pulses (Vg1 and Vg2). During this time t2 < t < t3, among the forward biased the forward biased switches, only two switches (S4b and S1b) are receiving PWM signal (Vg2) shown in switches only two switches (i.e., S3a and S2a) are receiving PWM signal (Vg1) shown in Figure4b. Figure 4b. Owing to this, S4b and S1b will be turned ON. Now, the path for the load current becomes Owing to this, only S3a and S2a will be turned ON to create a path for the load current that is, reverse i.e., S → D → load → S → D shown in Figure 6b. ElectronicsS3a → D 20186 →, 7,4load xb FOR→ PEER8S2a REVIEW→ D4 shown 1b in2 Figure6a. 7 of 19 The aforementioned four modes of operations have been applied and are illustrated in Figures 5 and 6.

S1a S2b S2a S1a S1b S2a S1b S2b

Io Io Vi(t) IH LOAD Vi(t) IH LOAD

S3b S3a S3a S4a S4b S4a S4b S3b

(a) (b)

FigureFigure 6. 6. NegativeNegative mode mode of of operation operation ( a) Mode 3 and ((bb)) ModeMode 4.4.

From the above modes of operation it can be concluded that SPMC can work as a frequency changer device as well as resonant converter. From the circuit diagram shown in Figure 4a, it can be seen that there are two paths for load current in each half cycle. In the positive half cycle, the two paths for the load current are S1a, D1, load, S4a, D7 and S2b, D3, load, S3b, D5, respectively. In the negative half cycle, the two paths for the load current are S4b, D8, load, S1b, D2 and S3a, D6, load, S2a, D4, respectively. Therefore, in this proposed switching algorithm, in each half cycle, the direction of load current could be changed depending on the time period of the conduction of switches. In other words, the desired output frequency of the load voltage/current depends on the switching frequency of the switches. Due to its frequency changer operation, it can be applied in IH application which has been shown in Figure 4c. By using the above modes of operation, switches’ operation status is given in Table 1.

Table 1. Switches operation status of SPMC.

Input Voltage Switches Time Output Voltage Mode (Vin) status Interval (Vout) (S1a/S4a) ON Mode 1 0 to t1 Vout > 0 (S2b/S3b) OFF Vin > 0 (S1a/S4a) OFF Mode 2 t1 to t2 Vout < 0 (S2b/S3b) ON (S3a/S2a) ON Mode 3 t2 to t3 Vout > 0 (S4b/S1b) OFF Vin < 0 (S3a/S2a) OFF Mode 4 t3 to t4 Vout < 0 (S4b/S1b) ON

There are several merits of the proposed switching strategy over conventional techniques [21]. Some of them are: • Compared to a previous switching strategy, the modified switching strategy has a simple but unique generation capability of resonant frequency or switching frequency which is the basic need of SPMC as a resonant converter for IH application. • Using this modified/proposed switching technique, SPMC can achieve a high frequency current very easily but using traditional/conventional switching technique, SPMC can generate only integral multiple of input supply frequency i.e., 50 Hz,100 Hz, 150 Hz and so on. That is why previous switching topology cannot be applied in the field of IH applications. • Also, the design of the controller for the proposed technique is quite simple because it needs to generate only two pulses as compared to previously developed switching techniques in which four pulses are needed for synthesization of frequency. Owing to this, the proposed technique reduces the design complexity of the controller.

Electronics 2018, 7, 149 7 of 17

Mode 4 (t3 < t < t4): This mode is also for the negative half cycle. In this mode, since switches S3a and S2a are not receiving PWM signal (Vg1), these switches gets turned OFF. Now in this mode, among the forward biased switches, only two switches (S4b and S1b) are receiving PWM signal (Vg2) shown in Figure4b. Owing to this, S4b and S1b will be turned ON. Now, the path for the load current becomes reverse i.e., S4b → D8 → load → S1b → D2 shown in Figure6b. The aforementioned four modes of operations have been applied and are illustrated in Figures5 and6. From the above modes of operation it can be concluded that SPMC can work as a frequency changer device as well as resonant converter. From the circuit diagram shown in Figure4a, it can be seen that there are two paths for load current in each half cycle. In the positive half cycle, the two paths for the load current are S1a, D1, load, S4a, D7 and S2b, D3, load, S3b, D5, respectively. In the negative half cycle, the two paths for the load current are S4b, D8, load, S1b, D2 and S3a, D6, load, S2a, D4, respectively. Therefore, in this proposed switching algorithm, in each half cycle, the direction of load current could be changed depending on the time period of the conduction of switches. In other words, the desired output frequency of the load voltage/current depends on the switching frequency of the switches. Due to its frequency changer operation, it can be applied in IH application which has been shown in Figure4c. By using the above modes of operation, switches’ operation status is given in Table1.

Table 1. Switches operation status of SPMC.

Input Voltage (Vin) Mode Switches Status Time Interval Output Voltage (Vout) (S /S ) ON Mode 1 1a 4a 0 to t V > 0 (S /S ) OFF 1 out Vin > 0 2b 3b (S1a/S4a) OFF Mode 2 t1 to t2 Vout < 0 (S2b/S3b) ON (S /S ) ON Mode 3 3a 2a t to t V > 0 (S /S ) OFF 2 3 out Vin < 0 4b 1b (S3a/S2a) OFF Mode 4 t3 to t4 Vout < 0 (S4b/S1b) ON

There are several merits of the proposed switching strategy over conventional techniques [21]. Some of them are: • Compared to a previous switching strategy, the modified switching strategy has a simple but unique generation capability of resonant frequency or switching frequency which is the basic need of SPMC as a resonant converter for IH application. • Using this modified/proposed switching technique, SPMC can achieve a high frequency current very easily but using traditional/conventional switching technique, SPMC can generate only integral multiple of input supply frequency i.e., 50 Hz, 100 Hz, 150 Hz and so on. That is why previous switching topology cannot be applied in the field of IH applications. • Also, the design of the controller for the proposed technique is quite simple because it needs to generate only two pulses as compared to previously developed switching techniques in which four pulses are needed for synthesization of frequency. Owing to this, the proposed technique reduces the design complexity of the controller. • The proposed switching technique can be applied for both operation of SPMC i.e., as a frequency changer or as resonant converter. Consequently, the proposed configuration of SPMC for IH applications shown in Figure4a has been modelled as an RLC circuit, which consists of a resistor (Ro), an inductor (Lo) and a capacitor (Cr) can be used to analyze the system behavior. It should also be noted that, at the resonant frequency, maximum output power is transferred to the load. Owing to this, a practical converter for IH applications always works at equal to or greater than the resonant frequency. For analyzing the circuit of SPMC as a resonant converter for IH applications, the following equations have been applied: Electronics 2018, 7, 149 8 of 17

2.2.1. Resonant Frequency

1 fr = √ (1) 2π L0Cr In terms of angular resonant frequency:

1 ωr = √ (2) L0Cr

2.2.2. Characteristics Impedances

s L0 1 Zeq = = = 2π fr L0 (3) Cr 2π frCr In terms of angular frequency: s L0 1 Zeq = = = ωr L0 (4) Cr ωrCr

2.2.3. Load Quality Factor

Zeq 2π f L 1 Q = = r 0 = (5) R0 R0 2π fr R0Cr In terms of angular frequency:

Zeq ω L 1 Q = = r 0 = (6) R0 R0 ωr R0Cr

2.2.4. Output Impedance of Equivalent Circuit (Figure4a)

1 1 Zeq = R0 + j 2π fr − = R0 1 + jQ 2π fn − (7) 2π frCr 2π fn s 2 2 1 Zeq = R0 1 + Q 2π fn − (8) 2π fn n o where 2π f = 2π f , ϕ = arg{Z(2π f )} = arctan Q 2π f − 1 ≥ 0. n 2π fr n 2π fn

2.2.5. Fundamental Output Voltage

( ) Vd, 0 < 2π fst < π V0 = (9) −Vd, π < 2π fst < 2π 2V V = d m π

2.2.6. Ieq, That Is, Load Current Flowing Through Tank

iLo = Im sin(ωt − ϕ) (10)

Vm 2Vd 2Vd cos ϕ 2Vd where Im = = = = r . |Zeq| π|Zeq| πRo πR 1+Q2 2π f − 1 o n 2π fn Electronics 2018, 7, 149 9 of 17

2.2.7. The Output Power

R 2V2 P = I2 0 = d (11) out m 2 2 πR 1 + Q2 2π f − 1 o n 2π fn

In Equation (11), at ωn = 2π fn = 1, the circuit becomes resonant, therefore, maximum power is transferred to the load. Moreover, output power could be varied with the help of a different value of quality factor (Q). Electronics 2018, 7, x FOR PEER REVIEW 10 of 19 3. Simulation Results and Its Discussion Table 2. Parameters used for the simulation. To validate the modified proposed technique, simulation has been done in MATLAB/SIMULINK (R2012a, Dhanbad,Symbol Jharkhand, India)Parameters environment by using parametersValue given in Table2. Firstly, simulation has beenV donein for the generalInput Voltage SPMC under R (100 Ω230) loadVr.m.s to create different output frequencies, that is, 100Ls Hz, 150 HzFilter and inductance 200 Hz, by using the modified20 mH proposed technique with experimental validation.Cf Subsequently,Filter simulationCapacitance has been done for3 the uF SPMC as a resonant converter, f Fundamental Frequency 50 Hz which works at a switching frequency of 25 kHz (Table2) in order to validate the modified technique. Cr Resonant Capacitor 0.8 uF Various results and waveforms of voltage and current have been taken along with THD of the input L0 Coil Inductance 52.7 uH current through the simulation. Figure7a shows the PWM controller that generates pulses ( Vg1 R0 Coil Equivalent Resistance 5 Ω and Vg2) of different frequencies. Figure7b depicts the simulation results of pulses waveform ( Vg1 P0 Output Power for heating 1100 W and Vg2), generated by the PWM controller are given to the switches of SPMC converter to generate fs Resonance Frequency output voltage/current of a different step up frequency. The frequency25 kHz and duty cycle of this pulse (Switching Frequency) is maintained at 25 kHz and 50% respectively. By changing the time period of this PWM controller, Figure 8a,b shows the waveforms of the input voltage (230 Vr.m.s) and input current. Figure 8c theshows desired the outputvalue of frequency THD in input can be current achieved. which Basically, was found in thisto be study, 2.60%. SPMC To achieve as a resonant this low converter THD forvalue, IH application a passive filter is focused has been upon. used which This direct is an essential AC–AC part converter of SPMC has when been it designed needs to operate to operate as at aresonant resonant converter frequency for (switching IH system.frequency) The HF switching of 25 kHz, results which in the is generation greater than of HF the harmonics frequency that as per −6 −6 calculationhas an inherent from equationtendency (1),to back with flow the parameterstowards the ofsupplyL0 = 52.7side ×and10 deteriorateand Cr = the 0.8 power× 10 qualityto satisfy, theresulting criteria in of a ZVS. wide variety of problem like distortion in the grid voltage/current. Thus, the low value of THD of 2.60% in input current ensures the attenuation of HF harmonics and makes the power supply of IH system practically viable. TableAs aforementioned, 2. Parameters usedfirst simulation for the simulation. has been done for generating 100 Hz output using proposed switching technique which is shown in Figure 9a,b. Figure 10 shows the typical Symbol Parameters Value simulated results of output voltage and load current for SPMC as a resonant converter in IH applications. The root mean Vsquarein (RMS) value of outputInput voltage Voltage and load current are 225.2 230V and Vr.m.s 5.162 A, which have been calculatedLs from the continuousFilter RMS inductance block. Therefore, average output 20 mH power can be C Filter Capacitance 3 uF calculated by usingf the product of these two RMS values. Here, for the calculation of maximum f Fundamental Frequency 50 Hz output average power, ideally φ (power factor) is taken as unity because it is known that at Cr cos Resonant Capacitor 0.8 uF resonant frequency,L0 capacitive reactance andCoil inductive Inductance reactance becomes equal, 52.7 but uHit is not the case for practical purposeR0 s. Coil Equivalent Resistance 5 Ω P0 OutputP Power= VI cos for φ; heating cos φ ≈ 1 1100 W fs Resonance Frequency (Switching Frequency) 25 kHz PW=225.. 2 × 5 162 ×= 1 1162

PWM Generator

S1a,S4a,S3a,S2a

NOT Gate

S2b,S3b,S4b,S1b

(a) (b)

Figure 7. (a) PWM Controller; (b) Simulation results of pulses i.e., Vg1 and Vg2 (180° out◦ of phase). Figure 7. (a) PWM Controller; (b) Simulation results of pulses i.e., Vg1 and Vg2 (180 out of phase).

Electronics 2018, 7, 149 10 of 17

Figure8a,b shows the waveforms of the input voltage (230 V r.m.s) and input current. Figure8c shows the value of THD in input current which was found to be 2.60%. To achieve this low THD value, a passive filter has been used which is an essential part of SPMC when it needs to operate as resonant converter for IH system. The HF switching results in the generation of HF harmonics that has an inherent tendency to back flow towards the supply side and deteriorate the power quality, resulting in a wide variety of problem like distortion in the grid voltage/current. Thus, the low value of THD of 2.60% in input current ensures the attenuation of HF harmonics and makes the power supply of IH system practically viable. As aforementioned, first simulation has been done for generating 100 Hz output using proposed switching technique which is shown in Figure9a,b. Figure 10 shows the typical simulated results of output voltage and load current for SPMC as a resonant converter in IH applications. The root mean square (RMS) value of output voltage and load current are 225.2 V and 5.162 A, which have been calculated from the continuous RMS block. Therefore, average output power can be calculated by using the product of these two RMS values. Here, for the calculation of maximum output average power, ideally cos φ (power factor) is taken as unity because it is known that at resonant frequency, capacitive reactance and inductive reactance becomes equal, but it is not the case for practical purposes. P = VI cos φ; cos φ ≈ 1

P = 225.2 × 5.162 × 1 = 1162 W Electronics 2018, 7, x FOR PEER REVIEW 11 of 19

(a)

(b)

(c)

Figure 8. Simulated waveforms of (a) input voltage (Vin) (b) input current (Iin) and (c) THD analysis Figure 8. Simulated waveforms of (a) input voltage (Vin)(b) input current (Iin) and (c) THD analysis of of input current (Iin). input current (Iin).

(a) (b)

Figure 9. Simulated waveforms of (a) Output voltage (Vout) and (b) Output current (Iout) at 100 Hz.

(a) (b)

Electronics 2018, 7, x FOR PEER REVIEW 11 of 19 Electronics 2018, 7, x FOR PEER REVIEW 11 of 19

(a) (a)

(b) (b)

(c) (c) Electronics 2018Figure, 7, 8 149. Simulated waveforms of (a) input voltage (Vin) (b) input current (Iin) and (c) THD analysis 11 of 17 Figure 8. Simulated waveforms of (a) input voltage (Vin) (b) input current (Iin) and (c) THD analysis of input current (Iin). of input current (Iin).

(a) (b) (a) (b) Figure 9. Simulated waveforms of (a) Output voltage (Vout) and (b) Output current (Iout) at 100 Hz. FigureFigure 9. Simulated 9. Simulated waveforms waveforms of of ( a(a)) Output Output voltage ( (VVoutout) and) and (b) ( Ob)utput Output current current (Iout) (atIout 100) atHz 100. Hz.

(a) (b) (a) (b)

Electronics 2018, 7, x FOR PEER REVIEW 12 of 19 Electronics Figure 10.2018(,a 7), x Simulated FOR PEER REVIEW waveform of output voltage (Vout) and (b) simulated waveform of12 output of 19 I currentFigure ( out 10) as. ( a) resonantSimulated converter waveform for of IHoutput application. voltage (Vout) and (b) simulated waveform of output Figurecurrent 10 (I.out (a) )as Simul a resonantated waveform converter offor output IH application voltage .( Vout) and (b) simulated waveform of output Figurecurrent 11 depicts (Iout) as a theresonant simulated converter result for IH of application the output. average power. In this study, the passive filter has beenFigure designed 11 depicts to protect the simulated from the result high-frequency of the output component average power. at the In input this study, side. the The p equivalentassive filterFigure has been 11 depicts designed the simulatedto protect resultfrom ofthe the high output-frequency average component power. In thisat the study, input the side. passive The circuit of the passive filter is shown in Figure 12a. From this figure, it can be observed that Zeq is too filterequivalent has been circuit designed of the passive to protect filter fromis shown the inhigh Figure-frequency 12a. From component this figure, at it the can inputbe observed side. Thethat high at a higher resonant frequency (switching frequency), which prevents the flow of HF component equivalentZeq is too high circuit at aof higher the passive resonant filter frequency is shown in(switching Figure 1 2frequency),a. From this which figure, prevents it can be theobserved flow of that HF currentZcomponenteq is at too the high gridcurrent at a side. higher at the The resonant grid simulated side. frequency The voltagesimulated (switching waveform voltage frequency), waveform across which the across filterprevents the capacitor filter the flowcapacitor is of shown HF is in Figurecomponentshown 12b. in This Figure current figure 12 bat shows. Thisthe gridfigure how side. shows the The HF how simulated component the HF voltagecomponent has beenwaveform has blocked been across blocked at the the inputatfilter the c side.inputapacitor side. is shown in Figure 12b. This figure shows how the HF component has been blocked at the input side.

Figure 11. Simulation results of output average power (P). FigureFigure 11. 11.Simulation Simulation results results of of output output average average power power (P). (P). ) in

) Ls in Ls V IH Load ( IH Load Zeq Cf Zeq Cf Input Voltage ( V

Input Voltage Passive Filter Passive Filter (a) (b) (a) (b) Figure 12. (a) Passive filter and (b) voltage across capacitive filter. Figure 12. (a) Passive filter and (b) voltage across capacitive filter. Figure 12. (a) Passive ﬁlter and (b) voltage across capacitive ﬁlter. In this section, various simulation results of the proposed SPMC as a resonant converter for IH In this section, various simulation results of the proposed SPMC as a resonant converter for IH application using the modified switching technique and its performance analysis have been application using the modified switching technique and its performance analysis have been provided. As aforementioned, the RLC circuit, which is modeled as an IH load, has been used for provided. As aforementioned, the RLC circuit, which is modeled as an IH load, has been used for analysis. In the next section, prototype implementation of SPMC as a resonant inverter and its results analysis. In the next section, prototype implementation of SPMC as a resonant inverter and its results has been discussed. has been discussed.

Electronics 2018, 7, 149 12 of 17

In this section, various simulation results of the proposed SPMC as a resonant converter for IH application using the modiﬁed switching technique and its performance analysis have been provided. As aforementioned, the RLC circuit, which is modeled as an IH load, has been used for analysis. In the next section, prototype implementation of SPMC as a resonant inverter and its results has beenElectronics discussed. 2018, 7, x FOR PEER REVIEW 13 of 19

4.4. Prototype Implementation and ItsIts Results To verify the converter performance followed by simulation result of SPMC,SPMC, based on proposed switching techniquetechnique for for IH IH application, application, a prototype a prototype laboratory laboratory set up set has up been has developed been developed with resistive with loadresistive which load is shownwhich inis Figureshown 13 in. Figure When this13. converterWhen this needs converter to be operatedneeds to be as aoperated resonant as converter a resonant for IHconver application,ter for IH IH application, coil could be IH connected coil could instead be connected of resistive instead load. of An resistive embedded load technology-based. An embedded Arduinotechnology mega-based 2560 Arduino (Mouser mega electronics, 2560 (M Banglore,ouser electronics India) is,used Banglore for generating, India) is PWMused for for generating the gate of thePWM switches. for the Ingate this of study, the switches. firstly, a In prototype this study, of the firstly, SPMC a prototype has been testedof the forSPMC 100 Hzhas outputbeen tested (i.e., as for a frequency100 Hz output changer (i.e., operation) as a frequency using thechanger modified operation) technique. using For the this, modified only the technique. resistive load For is this, assumed. only Subsequently,the resistive load this is converter assumed. has Subsequently, been tested asthis a resonantconverter inverter has been for tested IH application as a resonant using inverter the same for modifiedIH application technique using to the show same the modified uniqueness technique of developed to show technique. the uniqueness of developed technique.

Bi-directional Switch

(a) (b)

Figure 13 13.. ExperimExperimentalental setup of ( a) SPMC and its (b) bi-directionalbi-directional switch.

The implementation of the hardware circuit is divided into four parts: (a) designing of Power The implementation of the hardware circuit is divided into four parts: (a) designing of Power circuit; (b) ZCD circuit; (c) controller; (d) isolation and protection circuit. For designing of the power circuit; (b) ZCD circuit; (c) controller; (d) isolation and protection circuit. For designing of the power circuit, eight IGBTs and eight diodes have been used. Two IGBTs and two diodes have been used for circuit, eight IGBTs and eight diodes have been used. Two IGBTs and two diodes have been used making one bidirectional switch. In this study, four bidirectional switches have been used for a for making one bidirectional switch. In this study, four bidirectional switches have been used for a prototype implementation of the power circuit of SPMC, which has been shown in the Figure 13b. It prototype implementation of the power circuit of SPMC, which has been shown in the Figure 13b. is known that ZCD is used to detect every zero crossing of input AC voltage for synchronization of It is known that ZCD is used to detect every zero crossing of input AC voltage for synchronization of pulses. ZCD has been implemented with the help of Opamp IC741 (Fairchild semiconductor, San pulses. ZCD has been implemented with the help of Opamp IC741 (Fairchild semiconductor, San Jose, Jose, CA, United States), which works as a voltage amplifier. A synchronized pulse (Vg) generated CA, United States), which works as a voltage ampliﬁer. A synchronized pulse (V ) generated from from the ZCD is given to interrupt pin of Atmega 2560 (Mouser electronics, Banglore,g India) (i.e., the ZCD is given to interrupt pin of Atmega 2560 (Mouser electronics, Banglore, India) (i.e., digital digital pin 2 and digital pin 3) which is assigned as INT0 and INT1, detects the rising and falling edge pin 2 and digital pin 3) which is assigned as INT0 and INT1, detects the rising and falling edge of the of the pulse i.e., Vg (generated from the ZCD). As explained in section 2.2, that according to program pulse i.e., Vg (generated from the ZCD). As explained in Section 2.2, that according to program fed fed to the microcontroller, when it detects the rising edge of the pulse (Vg), it generates the pulse (Vg1) to the microcontroller, when it detects the rising edge of the pulse (V ), it generates the pulse (V ) of 10 ms (if the 100 Hz output of the converter is needed). Similarly, wheng it detects the falling edge,g1 of 10 ms (if the 100 Hz output of the converter is needed). Similarly, when it detects the falling edge, again microcontroller generates a pulse (Vg2) of 10 ms, but this pulse is in complete phase opposition again microcontroller generates a pulse (V ) of 10 ms, but this pulse is in complete phase opposition from previous pulse. The experimental validationg2 of synchronized pulse is shown in Figure 14a,b from previous pulse. The experimental validation of synchronized pulse is shown in Figure 14a,b shows the validity of synchronization of output voltage with respect to pulses (Vg1 and Vg2). shows the validity of synchronization of output voltage with respect to pulses (V and V ). g1 g2

Electronics 2018, 7, 149 13 of 17 Electronics 2018, 7, x FOR PEER REVIEW 14 of 19

Electronics 2018, 7, x FOR PEER REVIEW 14 of 19 ZCD Pulses generated from µC Output(Vg) ZCD Pulses generated from µC Output(Vg) s . m . r V s . 8

µC . m . Output(Vg1 & Vg2) r V 225 8 µC Output Voltage . Output(Vg1 & Vg2)

225 Output Voltage (a) (b)

(a) (b) FigureFigure 14.14. ((a)) ExperimentalExperimental resultsresults ofof ZCDZCD andand microcontrollermicrocontroller output.output. (Scale:(Scale: 55 V/div)V/div) and (b) experimentalexperimentalFigure 14. validation(validationa) Experimental ofof synchronizedsynchronized results of ZCD outputoutput and w.r.t.w.r.t microcontroller. pulses.pulses. (Scale:(Scale: output outputoutput. (Scale: voltage,voltage, 5 V/div) 7575V/div V/divand ( andband) Time,Timeexperimental, 1010 ms/div).ms/div) validation. of synchronized output w.r.t. pulses. (Scale: output voltage, 75 V/div and Time, 10 ms/div). After the synchronized pulses (Vg1 and Vg2) generation, it is given to the isolation circuit which After the synchronized pulses (Vg1 and Vg2) generation, it is given to the isolation circuit which isolatesAfter the converter the synchronized (higher power pulses level)(Vg1 and and V g2controller) generation, part it (lower is given power to the level). isolation The circuit supply which for the isolates the converter (higher power level) and controller part (lower power level). The supply for the microcontrollerisolates the converter and isolation (higher circuit power islevel) given and through controller a diode part (lower rectifier, power which level). is Theshown supply in the for blockthe microcontroller and isolation circuit is given through a diode rectiﬁer, which is shown in the block diagrammicrocontroller of Figure and3. Subsequently, isolation circuit an isisolation given through circuit hasa diode been rectifier, prepared which which is isshown also called in the theblock gate diagram of Figure3. Subsequently, an isolation circuit has been prepared which is also called the driverdiagram circuit of .Figure For this, 3. Subsequently, TLP250 optocoupler an isolation has circuit been has used. been preparedThe circuit which diagram is also calledand prototypethe gate gate driver circuit. For this, TLP250 optocoupler has been used. The circuit diagram and prototype implementationdriver circuit. ofFor the this, isolation TLP250 or optocouplerdriver circuit has are been shown used. in FigureThe circuit 15a,b ,diagram respectively and . prototypeThe output implementation of the isolation or driver circuit are shown in Figure 15a,b, respectively. The output pulsesimplementation (Vg1 and Vg2 )of from the isolationthe isolation or driver circuit circuit are givenare shown to the in switches Figure 1 5ofa,b SPMC, respectively power .circuit. The output pulsespulses (V g1(Vandg1 andV g2Vg2)) from from the the isolationisolation circuitcircuit are given given to to the the switches switches of of SPMC SPMC power power circuit. circuit. Vcc(9V) Vcc(9V)

1 8 I/P from 1 8 1K I/PuC from O/P to Gate uC 2 7 1K AC TLP250 Oterminals/P to Gate of 470uF 100ohm 2 7 12VAC TLP250 terminalsMOSFETs of 12V 470uF 100ohm 3 6 MOSFETs 3 6 44K 44K

4 5 4 5

G1 G2 G1 G2 ((aa))

ZeroZero Crossing DetectorDetector Circuit

IsolationIsolation or or Driver Driver CircuitCircuit

((bb)) Figure 15. (a) Circuit diagram and its (b) experimental setup of isolation or driver circuit. Figure 15. (a) Circuit diagramdiagram andand itsits ((b)) experimentalexperimental setupsetup ofof isolationisolation oror driverdriver circuit.circuit. As discussed in section 2.2, proposed controller comprises of three main units i.e., ZCD unit, As discussed in section 2.2, proposed controller comprises of three main units i.e., ZCD unit, microcontrollerAs discussed unit in Section and isolation 2.2, proposed circuit unit. controller On combining comprises these of three three units, main the units detailed i.e., hardware ZCD unit, microcontroller unit and isolation circuit unit. On combining these three units, the detailed hardware microcontrollercircuit diagram unit of andthe controller isolation circuitis shown unit. in the On Figure combining 16. these three units, the detailed hardware circuitcircuit diagramdiagram ofof thethe controllercontroller isis shownshown inin thethe FigureFigure 16 16..

Electronics 2018, 7, 149 14 of 17 Electronics 2018, 7, x FOR PEER REVIEW 15 of 19 Electronics 2018, 7, x FOR PEER REVIEW 15 of 19 Vcc(9V) 12V Vcc(9V) 1K 12V 1K 12V 1 8 uC 100ohm S1a,S4a,S3a,S2a 12V 1 8 1K ATMEGAuC 8 S1a,S4a,S3a,S2a 100ohm 2 7 1K O/P to Gate 1K ATMEGAINT0 8 TLP250 AC 230V 12V 2 7 terminalsO/P to Gate of 1K INT0 TLP250 AC 230V 12V 3 6 44K terminalsIGBTs of 12V INT1 3 6 44K IGBTs 5 1212V K INT1 4 12K 4 5 GND G1 GND G2 G1 Vcc(G92V) Vcc(9V)

1 8 100ohm 1 8 1K S2b,S3b,S4b,S1b 100ohm 2TLP250 7 1K SO2b/,PS to3b ,GateS4b,S 1b 2TLP250 7 terminalsO/P to Gate of 3 6 44K terminalsIGBTs of 3 6 44K IGBTs 4 5 4 5 G1 G2 G1 G2 Figure 16. CircuitCircuit implementation implementation of of proposed proposed controller controller for for SPMC SPMC as asa resonant a resonant conv convertererter for forIH Figure 16. Circuit implementation of proposed controller for SPMC as a resonant converter for IH IHapplic applications.ations. applications. As aforementioned, using the modified technique, SPMC is operated first as a frequency changer AsAs aforementioned,aforementioned, usingusing thethe modifiedmodified technique,technique, SPMCSPMC isis operatedoperated firstfirst asas a frequency changer (which converts 50 Hz grid frequency to 100 Hz with the input of 230 Vr.m.s) with a resistive load and (which(which convertsconverts 50 Hz Hz grid grid frequency frequency to to 100 100 Hz Hz with with the the input input of of230 230 Vr.m.s Vr.m.s) with) with a resistive a resistive load load and later, it is tested as a resonant converter for IH application. Figure 17a,b shows the experimental andlater, later, it is it tested is tested as asa resonant a resonant converte converterr for for IH IH application. application. Figure Figure 1177a,ba,b shows the experimentalexperimental results of SPMC as a frequency changer (at 100 Hz output voltage/current) and SPMC as a resonant resultsresults ofof SPMCSPMC asas aa frequencyfrequency changerchanger (at(at 100100 HzHz outputoutput voltage/current)voltage/current) and SPMC asas aa resonantresonant converter for IH applications at a frequency of 25 kHz which validate simulation results. As seen converterconverter forfor IH IH applications applications at at a frequencya frequency of 25of kHz25 kHz which which validate validate simulation simulation results. results. As seen As fromseen from the Figure 17a,b, current and voltage are almost in the same phase. So experimentally, the load thefrom Figure the Fi 17gurea,b, 17 currenta,b, current and voltage and voltage are almost are almost in the samein the phase. same phase. So experimentally, So experimentally, the load the power load power factor in case of when SPMC has operated as a frequency changer (i.e., at 100 Hz output on factorpower in factor case ofin whencase of SPMC when has SPMC operated has operated as a frequency as a frequency changer changer (i.e., at 100 (i.e., Hz at output 100 Hz on output resistive on resistive load) was found to be 0.98 which is quite close to unity. In the case of when SPMC has operated load)resistive was load) found was to found be 0.98 to whichbe 0.98 is which quite is close quite to close unity. to Inunity. the I casen the of case when of when SPMC SPMC has has operated operated as as resonant converter for IH applications was found to be 0.91. Figure 18 shows the experimental result of resonantas resonant converter converter for for IH IH applications applications was wa founds found to to be be 0.91. 0.91. Figure Figure 18 18 shows shows the the experimental experimental resultresult ofof output voltage with respect to input current which validate that output is perfectly synchronized with outputoutput voltagevoltage withwith respectrespect toto inputinput currentcurrent whichwhich validatevalidate thatthat outputoutput isis perfectlyperfectly synchronizedsynchronized withwith input supply. The value of THD for the input current is experimentally found to be 3.91% which is quite inputinput sup supply.ply. The The value value of of THD THD for for the the input input current current is experimentally is experimentally found found to be to 3.91 be% 3.91% which which is quite is low. Various experimental results of voltage and current waveform under R load or IH coil have been quitelow. Various low. Various experimental experimental results results of voltage of voltage and current and current waveform waveform under under R load R loador IH or coil IH have coil havebeen taken in a 200 MHz digital signal oscilloscope (DSO) using a current sensor probe to verify the beentaken taken in a in200 a 200MHz MHz digital digital signal signal oscilloscope oscilloscope (DSO) (DSO) using using a acurrent current sensor sensor probe probe to to verify thethe validity of proposed/modified switching algorithm. It has been found that, the proposed technique validityvalidity ofof proposed/modifiedproposed/modified switching algorithm. I Itt has been found that, thethe proposedproposed techniquetechnique can be used in the field of IH applications. cancan bebe usedused inin thethe fieldfield ofof IHIH applications.applications.

Output Voltage Output Current(100Hz) Output(220V Voltager.m.s) Output Current(100Hz) (220Vr.m.s) 2 A 2 A r . m s r . m s 225 . 8 V

225 . 8 V Output Current (6A) Output Voltage(100Hz) Output Current (6A) Output Voltage(100Hz) (a) (b) (a) (b) Figure 17. (a) Experimental verification of simulated output voltage and current at 100 Hz i.e., as a Figure 17. (a) Experimental verification of simulated output voltage and current at 100 Hz i.e., as a Figurefrequency 17. changer.(a) Experimental (Scale: output veriﬁcation voltage, of 75 simulated V/div; output output current, voltage 1 A/div and current and time at, 100 10 ms/div) Hz i.e., asand a frequency changer. (Scale: output voltage, 75 V/div; output current, 1 A/div and time, 10 ms/div) and frequency(b) experimental changer. validation (Scale: output of output voltage, voltage 75 V/div;and current output of current, SPMC as 1 A/diva resonant and time,converter 10 ms/div) for IH (b) experimental validation of output voltage and current of SPMC as a resonant converter for IH andapplications (b) experimental at 25 KHz. validation (Scale: Output of output voltage, voltage 35 andV/div; current output of current, SPMC as 2 aA/div). resonant converter for IH applicationsapplications atat 2525 KHz.KHz. (Scale:(Scale: Output voltage,voltage, 3535 V/div;V/div; output output current, current, 2 2 A/div).

Electronics 2018, 7, 149 15 of 17 Electronics 2018, 7, x FOR PEER REVIEW 16 of 19

THD=3.91% Input Current (50Hz) 2 . 1 A r . m s 225 . 8 V Output Voltage (100Hz)

FigureFigure 18. Experimental 18. Experimental results results of of output output voltage voltage w.r.t.w.r.t. input current current (Scale: (Scale: Output Output voltage, voltage, 75 V/div; 75 V/div; input inputcurrent, current, 1 A/div; 1 A/div; and and Time, Time, 10 ms/div) 10 ms/div)..

5. Conclusion 5. Conclusions In this proposed work, a cogent switching algorithm has been employed for direct conversion of In this proposed work, a cogent switching algorithm has been employed for direct conversion utility frequency to HFAC through SPMC topology for IH applications. The algorithm requires less of utility frequency to HFAC through SPMC topology for IH applications. The algorithm requires number of components along with fewer PWM signals as compared to the conventional IH system, thus less number of components along with fewer PWM signals as compared to the conventional IH leading to reduction in the cost and complexity of the controller. This direct AC to high frequency AC system,conversion thus leading based toon reduction proposed inswitching the cost algorithm and complexity enhances of thethe controller.overall efficiency This direct of the AC IH tosystem. high frequencyVarious AC simulation conversion and based experimental on proposed results switchingcorroborate algorithm the potential enhances pragmatic the overallapplications efficiency of the of theSPMC IH system. as a resonant Various converter simulation using the and proposed experimental switching results technique corroborate to generate the 25 potential kHz current/voltage pragmatic applicationsdirectly from of the the SPMC 50 Hz as agrid resonant frequency. converter It has usingthe additional the proposed ability switching of reducing technique switching to generatelosses by 25 kHzincorporating current/voltage a ZVS directlycondition from, high the power 50 Hz factor grid and frequency. low input It hasTHD the which additional improve ability the power of reducing quality switchingat the input losses side by. incorporating a ZVS condition, high power factor and low input THD which improve the power quality at the input side. Author Contributions: A.K. and P.K.S. developed the concept and also wrote this research article. A.K. designed Authorand Contributions: performed all A.K.the experiment. and P.K.S. developed D.K.M. and the M.J concept.B. thoroughly and also wroteanalyze thisd the research data, article.simulation A.K. results designed and and performedexperimental all results. the experiment. All authors D.K.M. in this paper and M.J.B.R. contributed thoroughly equally.analyzed the data, simulation results and experimental results. All authors in this paper contributed equally. Funding: This research received no external funding. Funding: This research received no external funding. Acknowledgments:Acknowledgments:Authors Authors acknowledges acknowledges the the “Indian “Indian Institute Institute of of Technology Technology (Indian (IndianSchool School ofof Mines),Mines), Dhanbad,Dhanbad, India” India for” providing for providing infrastructure infrastructure support. support. ConflictsConflicts of Interest: of Interest:The authorsThe authors declare declare no conflictno conflict of interest. of interest.

Appendix A

Table A1. List of components used for experimental design and their speciﬁcation.

Components Speciﬁcation/Ratings GBT (IXRH40N120) (1200 V, 55 A) diode (10A7) (700 V, 10 A) microcontroller Atmega 2560 op-amp IC741 diode (1N4007) (1000 V, 1 A) centre taped transformer (12–0–12) V, 2 A TLP250 25 kHz IC Socket base 8 pin DIP heat sink for IGBT resistance 1 K, 12 K, 100 Ω, 44 K capacitor 470 µF IH coil Litz wire based

Electronics 2018, 7, 149 16 of 17

References

1. Acero, J.; Burdio, J.M.; Barragan, L.A.; Navarro, D.; Alonso, R.; Ramon, J.; Monterde, F.; Hernandez, P.; Llorente, S.; Garde, I. Domestic induction appliances. IEEE Ind. Appl. Mag. 2010, 16, 39–47. [CrossRef] 2. Sarnago, H.; Lucia, O.; Mediano, A.; Burdio, J.M. Modulation scheme for improved operation of an RB-IGBT-based resonant inverter applied to domestic induction heating. IEEE Trans. Ind. Electron. 2013, 60, 2066–2073. [CrossRef] 3. Sarnago, H.; Lucía, O.; Mediano, A.; Burdío, J.M. Class-D/DE dual-mode-operation resonant converter for improved-efﬁciency domestic induction heating system. IEEE Trans. Power Electron. 2013, 28, 1274–1285. [CrossRef] 4. Lucia, O.; Burdio, J.M.; Millan, I.; Acero, J.; Puyal, D. Load-adaptive control algorithm of half-bridge series resonant inverter for domestic induction heating. IEEE Trans. Ind. Electron. 2009, 56, 3106–3116. [CrossRef] 5. Lucía, O.; Maussion, P.; Dede, E.; Burdío, J.M. Induction heating technology and its applications: Past Developments, current Technology, and future challenges. IEEE Trans. Ind. Electron. 2011, 61, 2509–2520. [CrossRef] 6. Mishima, T.; Takami, C.; Nakaoka, M. A new current phasor-controlled ZVS twin half-bridge high-frequency resonant inverter for induction heating. IEEE Trans. Ind. Electron. 2014, 61, 2531–2545. [CrossRef] 7. Saha, B.; Kim, R.Y. High Power Density Series Resonant Inverter Using an Auxiliary Switched Capacitor Cell for Induction Heating Applications. IEEE Trans. Power Electron. 2014, 29, 1909–1918. [CrossRef] 8. Lucía, O.; Burdío, J.M.; Millán, I.; Acero, J.; Barragán, L.A. Efﬁciency-oriented design of ZVS half-bridge series resonant inverter with variable frequency duty cycle control. IEEE Trans. Power Electron. 2010, 25, 1671–1674. [CrossRef] 9. Sarnago, H.; Lucía, O.; Pérez-Tarragona, M.; Burdío, J.M. Dual-Output Boost Resonant Full-Bridge Topology and its Modulation Strategies for High-Performance Induction Heating Applications. IEEE Trans. Ind. Electron. 2016, 63, 3554–3561. [CrossRef] 10. Singh, B.; Pandey, R. Improved Power Quality Buck-Boost Converter fed LLC Resonant Converter for Induction Heater. In Proceedings of the 2016 IEEE 6th International Conference on Power Systems (ICPS), New Delhi, India, 4–6 March 2016; pp. 1–6. 11. Pham, H.; Fujita, H.; Ozaki, K.; Uchida, N. Dynamic analysis and control of a zone-control induction heating system. In Proceedings of the 2011 IEEE conference of Energy Conversion Congress and Exposition(ECCE), Phoenix, AZ, USA, 17–22 September 2011; pp. 4093–4100. 12. Fujita, H.; Akagi, H. Pulse-Density-Modulated Power Control of a 4 kW, 450 kHz Voltage-Source Inverter for Induction Melting Applications. IEEE Trans. Ind. Appl. 1996, 32, 279–286. [CrossRef] 13. Grajales, L.; Sabatk, J.A.; Wang, K.R.; Tabisz, W.A.; Lee, F.C. Design of a 10 kW, 500 kHz Phase-Shift Controlled Series-Resonant Inverter for Induction Heating. In Proceedings of the 1993 IEEE Industry Applications Conference Twenty-Eighth IAS Annual Meeting, Toronto, ON, Canada, 2–8 October 1993; pp. 843–849. 14. Yilmaz, I.; Ermi¸s,M.; Çadirci, I. Medium-frequency induction melting furnace as a load on the power system. IEEE Trans. Ind. Appl. 2012, 48, 1203–1214. [CrossRef] 15. Millan, I.; Burdio, J.M.; Acero, J.; Lucia, O.; Liorente, S. Series resonant inverter with selective harmonic operation applied to all-metal domestic induction heating. IET Power Electron. 2011, 4, 587–592. [CrossRef] 16. Koertzen, H.W.; Van Wyk, J.D.; Ferreira, J.A. Design of the half-bridge series resonant converters for induction cooking. In Proceedings of the Power Electronics Specialist Conference (PESC), Atlanta, GA, USA, 18–22 June 1995; pp. 729–735. 17. Dawson, F.P.; Jain, P. A Comparison of Load Commutated Inverter Systems for Induction Heating and Melting Applications. IEEE Trans. Power Electron. 1991, 6, 430–441. [CrossRef] 18. Koertzen, H.W.; Ferreira, J.A.; Van Wyk, J.D. A comparative study of single switch induction heating converters using novel component effectivity concepts. In Proceedings of the 23rd Annual IEEE Power Electronics Specialists Conference (PESC), Toledo, Spain, 29 June–3 July 1992; pp. 298–305. 19. Jimenez, O.; Lucia, O.; Urriza, I.; Barragan, L.A.; Mattavelli, P.; Boroyevich, D. An FPGA-based gain-scheduled controller for resonant converters applied to induction cooktops. IEEE Trans. Power Electron. 2014, 29, 2143–2152. [CrossRef] 20. Nagarajan, B.; Sathi, R.R. Phase locked loop based pulse density modulation scheme for the power control of induction heating applications. J. Power Electron. 2015, 15, 65–77. [CrossRef] Electronics 2018, 7, 149 17 of 17

21. Idris, Z.; Hamzah, M.K.; Saidon, M.F. Implementation of single-phase matrix converter as a direct AC–AC converter with commutation strategies. In Proceedings of the 2006 37th IEEE Power Electronics Specialists Conference, Jeju, South Korea, 18–22 June 2006; pp. 1–7. 22. Li, H.L.; Hu, A.P.; Covic, G.A. A direct AC–AC converter for inductive power-transfer systems. IEEE Trans. Power Electron. 2012, 27, 661–668. [CrossRef] 23. Arevalo, S.L.; Zanchetta, P.; Wheeler, P.W.; Trentin, A.; Empringham, L. Control and implementation of a matrix-converter-based AC ground power-supply unit for aircraft servicing. IEEE Trans. Ind. Electron. 2010, 57, 2076–2084. [CrossRef] 24. Kolar, J.W.; Schafmeister, F.; Round, S.D.; Ertl, H. Novel three-Phase AC–AC sparse matrix converters. IEEE Trans. Power Electron. 2007, 22, 1649–1661. [CrossRef] 25. Trentin, A.; Zanchetta, P.; Clare, J.; Wheeler, P. Automated optimal design of input ﬁlters for direct ac/ac matrix converters. IEEE Trans. Ind. Electron. 2012, 59, 2811–2823. [CrossRef] 26. Yamamoto, E.; Hara, H.; Uchino, T.; Kawaji, M.; Kume, T.J.; Kang, J.K.; Krug, H.P. Development of MCs and its applications in industry. IEEE Ind. Electron. Mag. 2011, 5, 4–12. [CrossRef] 27. Hosseini, S.H.; Shariﬁan, M.B.; Sabahi, M.; Yazdanpanah, A.; Gharehpetian, G.H. Bi-directional power electronic transformer for induction heating systems. In Proceedings of the 2008 Canadian Conference on Electrical and Computer Engineering, Niagara Falls, ON, Canada, 4–7 May 2008; pp. 347–350. 28. Kim, Y.; Okuma, S.; Iwata, K. Characteristics and Starting Method of a Cycloconverter With a Tank Circuit for Induction Heating. IEEE Trans. Power Electron. 1988, 3, 236–244. [CrossRef] 29. Sugimura, H.; Mun, S.P.; Kwon, S.K.; Mishima, T.; Nakaoka, M. Direct AC–AC resonant converter using one-chip reverse blocking IGBT-Based bidirectional switches for HF induction heaters. In Proceedings of the 2008 IEEE International Symposium on Industrial Electronics, Cambridge, UK, 30 June–2 July 2008; pp. 406–412. 30. Salehifar, M.; Moreno-eguilaz, M.; Sala, V.; Romeral, L. A Novel AC–AC Converter Based SiC for Domestic Induction Cooking Applications. In Proceedings of the 2013 28th Twenty-Eighth Annual IEEE Applied Power Electronics Conference and Exposition (APEC), Long Beach, CA, USA, 17–21 March 2013; pp. 3216–3223. 31. Nguyen-Quang, N.; Stone, D.A.; Bingham, C.M.; Foster, M.P. Single phase matrix converter for radio frequency induction heating. In Proceedings of the International Symposium on Power Electronics, Electrical Drives, Automation and Motion (SPEEDAM), Taormina, Italy, 23–26 May 2006; pp. 614–618. 32. Mohite, S.B.; Gujarathi, P.K. A Design and Implementation of a Novel Multimode Single Phase Matrix Converter. In Proceedings of the 2010 IEEE Power Electronics Electrical Drives Automation and Motion SPEEDAM, Pisa, Italy, 14–16 June 2010; pp. 1227–1230. 33. Nguyen-Quang, N.; Stone, D.A.; Bingham, C.M.; Foster, M.P. A three-phase to single-phase matrix converter for high-frequency induction heating. In Proceedings of the 2009 13th European Conference on Power Electronics and Applications, Barcelona, Spain, 8–10 September 2009; pp. 1–10. 34. Nguyen-Quang, N.; Stone, D.A.; Bingham, C.M.; Foster, M.P. Comparison of single-phase matrix converter and H-bridge converter for radio frequency induction heating. In Proceedings of the 2007 European Conference on Power Electronics and Applications, Aalborg, Denmark, 2–5 September 2007; pp. 1–7. 35. Jose, P.S.; Deepika, N.C.; Nisha, S.N. Applications of single phase matrix converter. In Proceedings of the 2011 International Conference on Emerging Trends in Electrical and Computer Technology, Nagercoil, India, 23–24 March 2011; pp. 386–391. 36. Kumar, A.; Sadhu, M.; Das, N.; Sadhu, P.K.; Roy, D.; Ankur, G. A Survey on High-Frequency Inverter and Their Power Control Techniques for Induction Heating Applications. J. Power Technol. 2011, 97, 201–213.

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